CA1050114A - Admittance measuring system for monitoring the condition of materials - Google Patents
Admittance measuring system for monitoring the condition of materialsInfo
- Publication number
- CA1050114A CA1050114A CA235,144A CA235144A CA1050114A CA 1050114 A CA1050114 A CA 1050114A CA 235144 A CA235144 A CA 235144A CA 1050114 A CA1050114 A CA 1050114A
- Authority
- CA
- Canada
- Prior art keywords
- admittance
- bridge network
- wire transmitter
- materials
- voltage
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
Classifications
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
- G01R27/00—Arrangements for measuring resistance, reactance, impedance, or electric characteristics derived therefrom
- G01R27/02—Measuring real or complex resistance, reactance, impedance, or other two-pole characteristics derived therefrom, e.g. time constant
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01N—INVESTIGATING OR ANALYSING MATERIALS BY DETERMINING THEIR CHEMICAL OR PHYSICAL PROPERTIES
- G01N27/00—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means
- G01N27/02—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating impedance
- G01N27/028—Circuits therefor
Abstract
ABSTRACT
An intrinsically safe system for monitoring the condition of mater-ials includes a low power, stable frequency RF oscillator of the class C type comprising a resonant circuit which is coupled to a bridge network including the admittance of materials between a probe electrode and a grounded support member juxtaposed to the materials. The output of the network generates an AC
error signal which is applied to a phase sensitive detector including a chopper and a low power chopper drive for generating a DC signal representing the mag-nitude of the AC signal at a predetermined phase angle. The bridge network which may be linearly calibrated is isolated from the oscillator and the output error signal circuitry so as to allow the oscillator and the output error sig-nal circuitry to float with respect to the grounded support member and the power supply associated therewith. The rms voltage across the admittance re-presenting the condition of materials is limited so as to permit the system to comprise a two-wire transmitter wherein the sole source of power for the trans-mitter is derived from a 4-20 milliamp current drawn by the transmitter.
An intrinsically safe system for monitoring the condition of mater-ials includes a low power, stable frequency RF oscillator of the class C type comprising a resonant circuit which is coupled to a bridge network including the admittance of materials between a probe electrode and a grounded support member juxtaposed to the materials. The output of the network generates an AC
error signal which is applied to a phase sensitive detector including a chopper and a low power chopper drive for generating a DC signal representing the mag-nitude of the AC signal at a predetermined phase angle. The bridge network which may be linearly calibrated is isolated from the oscillator and the output error signal circuitry so as to allow the oscillator and the output error sig-nal circuitry to float with respect to the grounded support member and the power supply associated therewith. The rms voltage across the admittance re-presenting the condition of materials is limited so as to permit the system to comprise a two-wire transmitter wherein the sole source of power for the trans-mitter is derived from a 4-20 milliamp current drawn by the transmitter.
Description
~05q~14 This invention relates to RF admittance measuring systems for moni-toring the condition of materials, and more particularly, to systems of this type which are adapted for use at remote locations.
Heretofore, two-wira transmitters have been utilized to monitor va-rious conditions at a remote location. Typically, a two-wire transmitter at a remote location is connected in series with a power supply and load at ano-ther location through two transmission wires. As the candition being monito-red at the transmitter varies, the effective series resistance across the t^
transmitter varies so as to produce a change in the current drawn by the trans-mitter which represents (e.g., is generally proportional to) the condition being monitored. A two-wire transmitter of this type is designed for low power consumption since the amount of power available to the transmitter from the remotely located power supply may be limited, Furthermore, certain appli-cations may require that the two-wire transmitter be "intrinsically safe'l so as to permit its use in the monitoring of conditions in an explosive environ-ment~ Under these ~ircumstances, low energy usually associated with low power consumption becomes important so as to preclude the possibility of ignition and e~plosion.
~lthough the state of the art in two-wire transmitters is adequate for monitoring various types of conditions, the prior art technology with res-pect to the RF admittance measurement is deficient for two-wire transmitters for the following reasons.
When measuring the RF admittance between a probe eflectrode and a reference surface such as a ~rounded vessel, the resistance in parallel with the capacitance between the probe electrode and the grounded vessel becomes very important from a power consumption standpoint. Heretofore, it has gene-rally been assumed that shunt resistance is sufficiently small in a suffi~ient-ly large number of applications so as to render the power provided by the 4 milliamp current in a 4-20 milliamp two-wire transmitter system insufficient to power the two-wire transmitter. In other words, the shunt resistance alone -1- ~
~ight consulrle more power than is available at the 4 milliamp condition l~avinglittle or no power to operate the circuitry of tho transmitter ~oreover, in order for an admittallce mcasurcmcnt to be accurate, rc-liable phase-s~nsitive detection must be utilized. However, such roliability usu~lly requires a substanti~l source of power which is inconsistent with the low power rcquirements o~ a two-wire transmitter as discussed above and the available power because o~ the shunt resistance. This combination of factors imposes severe restrictions on the power which is generally considcred neces-sary to provide a reliable RF signal from a suitable oscillator. Similar res-; In trictions are placed on the power generally considered necessary to assure ~hat the phase detector operates with a high degree of reliability AnQther problem which is somewhat unique to admittance measuremonts is the isolation of the bridge network in which the unkonwn admittance being measured is oonnected. Typically, the unknown admittance being measured is from a probe electrode to ground as disclosed in Maltby et al U.S. patent ; 3,781,672 issued December 25, 1973, and Maltby U.S. patent 3,7~6,980 issued ' - Dece~ber 19, 1972, both of-which are assigned bo the assignee of this invention. However, a power supply at a location remote from the bridge net-work as i~ the case of a two-wire transmitter~ may not be connected to grou~d j in a manner compatible with the bridge network. It is therefore neccssary to isolate the bridge network from the bridge power supply so as to permit the bridge network to be connected to ground regardless of the power supply cir-cuit. Moreover, if the voltaoe across the unknown admittance were reduced to minimize power consumption, the signal representing the unbalance of the bridge network would require amplification. Accordingly, the problem exists of pro-viding an isolated source of power for such amplification.
Other problems exis~ in assuring linear and stable calibration of the admittance measuring system. It is also important to provide a system ~hich ~ill wor~ with various types of probes and various lengths of cables a~-- sociated with the probes without adversely affecting tlle admittance measurell~nt.
rO a very large de~rec, the above-m0n~ioned problems are enco~ltered
Heretofore, two-wira transmitters have been utilized to monitor va-rious conditions at a remote location. Typically, a two-wire transmitter at a remote location is connected in series with a power supply and load at ano-ther location through two transmission wires. As the candition being monito-red at the transmitter varies, the effective series resistance across the t^
transmitter varies so as to produce a change in the current drawn by the trans-mitter which represents (e.g., is generally proportional to) the condition being monitored. A two-wire transmitter of this type is designed for low power consumption since the amount of power available to the transmitter from the remotely located power supply may be limited, Furthermore, certain appli-cations may require that the two-wire transmitter be "intrinsically safe'l so as to permit its use in the monitoring of conditions in an explosive environ-ment~ Under these ~ircumstances, low energy usually associated with low power consumption becomes important so as to preclude the possibility of ignition and e~plosion.
~lthough the state of the art in two-wire transmitters is adequate for monitoring various types of conditions, the prior art technology with res-pect to the RF admittance measurement is deficient for two-wire transmitters for the following reasons.
When measuring the RF admittance between a probe eflectrode and a reference surface such as a ~rounded vessel, the resistance in parallel with the capacitance between the probe electrode and the grounded vessel becomes very important from a power consumption standpoint. Heretofore, it has gene-rally been assumed that shunt resistance is sufficiently small in a suffi~ient-ly large number of applications so as to render the power provided by the 4 milliamp current in a 4-20 milliamp two-wire transmitter system insufficient to power the two-wire transmitter. In other words, the shunt resistance alone -1- ~
~ight consulrle more power than is available at the 4 milliamp condition l~avinglittle or no power to operate the circuitry of tho transmitter ~oreover, in order for an admittallce mcasurcmcnt to be accurate, rc-liable phase-s~nsitive detection must be utilized. However, such roliability usu~lly requires a substanti~l source of power which is inconsistent with the low power rcquirements o~ a two-wire transmitter as discussed above and the available power because o~ the shunt resistance. This combination of factors imposes severe restrictions on the power which is generally considcred neces-sary to provide a reliable RF signal from a suitable oscillator. Similar res-; In trictions are placed on the power generally considered necessary to assure ~hat the phase detector operates with a high degree of reliability AnQther problem which is somewhat unique to admittance measuremonts is the isolation of the bridge network in which the unkonwn admittance being measured is oonnected. Typically, the unknown admittance being measured is from a probe electrode to ground as disclosed in Maltby et al U.S. patent ; 3,781,672 issued December 25, 1973, and Maltby U.S. patent 3,7~6,980 issued ' - Dece~ber 19, 1972, both of-which are assigned bo the assignee of this invention. However, a power supply at a location remote from the bridge net-work as i~ the case of a two-wire transmitter~ may not be connected to grou~d j in a manner compatible with the bridge network. It is therefore neccssary to isolate the bridge network from the bridge power supply so as to permit the bridge network to be connected to ground regardless of the power supply cir-cuit. Moreover, if the voltaoe across the unknown admittance were reduced to minimize power consumption, the signal representing the unbalance of the bridge network would require amplification. Accordingly, the problem exists of pro-viding an isolated source of power for such amplification.
Other problems exis~ in assuring linear and stable calibration of the admittance measuring system. It is also important to provide a system ~hich ~ill wor~ with various types of probes and various lengths of cables a~-- sociated with the probes without adversely affecting tlle admittance measurell~nt.
rO a very large de~rec, the above-m0n~ioned problems are enco~ltered
-2-10501~4 when the system for monitoring the condition of materials comprises a battery-operated unit rather than a two-wire transmitter. Under these circumstances, the available power is again l;mited.
It is an overa~l objec~ of this invention to monitor ~he condition of materials at a remote location.utilizing RF admittance measurements.
It is a more specific object of this invention to minimize the power consumption necessary in making the RF admittance measurements.
It is also a more specific object of the invention to provide an in-trinsically safe system for measuring RF admittance measurements.
It is a still more specific object of this invention to provide a two-wire transm~tter which is capable of operating from the power supplied by 4-20 milliamps of current which flows through the two transmission wires con-necting the two-wire transmitter to a remotely located power supply.
In accordance with these objects, a particularly preferred embodi-ment of the invention comprises an admittance sensing probe including a sensing electrode adapted to detect the admittance of materials for monitoring the con-dition of the materials, an RF signal generator and a bri~dge network coupled to the RF signal generator. The bridge network includes the admitbance detec-ted by the probe such that the unbalance of the network corresponds to the con-dition of the materials being monitored while the RF signal generator applies a voltage of less than the ~r~ rms across the admittance detected, where V is the voltage across the two-wire transmitter. Output means are coupled to `the bridge network for changing the current flow drawn by the two-wire transmitter from ~ milliamps to 20 milliamps in response to the unbalænce of the bridge net-work so as to represent the condition of the materials.
It is also a specific object of this invention to provide isolation between a floating power supply and ~he probe so as to permit the admittance of the materials to be measured between the sensing electrodes and a grounded mem-ber.
It is a further object of this invention to provide DC isolation 10S01~4 which is not subject to high voltage breakdown.
In accordance with these objects, the preferred embodiment of the in-vention comprises means for DC isolating the bridge network from the RF signal generator and the output means. The DC isolating means may comprise a first transformer having a primary connected to the RF signal generator and a secon-dary forming part of the bridge network. The DC isolating means may further comprise a second transformer having a primary connected to the bridge network and a secondary connected to the output means.
In order to increase the output vol~age from the bridge network be-fore application to the output means~ an amplifier me~ns may be coupled to the output of the bridge network. In order to maint~in isolation of the bridge network from the remainder of the t~o--wire transmitter and the remote power supply, the power supply for the amplifier is derived from the rectifying means coupled to the bridge network.
It is another object of this invention to provide for stable cali-bration of the admittance measurement.
In accordance with this object, the RF signal generator comprises an RF oscillator and a regulating circuit for the oscillator for maintain~ng the amplitude of the RF signal substantially constant. The regulating circuit comprises means for full wave rectifying the output of the oscillator and a capacitor coupled to the full wave rectifying means which is charged by current flow through the rectifying means. A voltage divider is connected between the capacitor and a control input of the oscillator so as to maintain the amplitu-de of the RF signal from the oscillator and the voltage across the capacitor substantially constant. By maintaining the amplit~de of the RF signal substan-tially constant despite changes in the ~operating characteristics of the trans-lstors within the oscillator and despite resistive loading from the sensing electrode of the probe to ground, stable calibration of the admittance measure-ment is attained.
In a particularly preferred embodiment, the RF oscillator comprises ~LOS01~1L4 a class C oscillator including a multi-vibrator and a resonant circuit so as to develop an undistorted RF sinusoidal signal while still limiting power con-sumption. The resonant circuit may ~omprise the aforesaid first trans~ormer and the admittance in the bridge network.
In further accordance with the object of stable calibration and mi-nimizing power consumption, the output means of the two-wire transmitter com-prises a phase-sensitive detector including a chopper and ac'chopper drive means for generating a chopper trigger signal for application to the chopper means.
The chopper drive means comprises a pair of field effect transistors having first and second channel electrodes and a gate electrode respectively and fur-ther comprises a pair of channel resistors. The first channel electrodes are interconnected and the second channel electrodes are connected to a source of regulated voltage through the channel resistors. The resonant circuit of the RF oscillator is coupled to the gate electrodes of each of the fi~d effect transistors so as to render the field effect transistors alternately conduct-ive. The channel resistors minimize power consumption by limiting the current flow through the pair of field effect transistors should the transistors be simultaneously conductive. In addition, the channel resistors which reduce the output voltage from channel e~lectrode-to-channel electrode provide~ a shar-per kaee in the input-output cur~e at the threshold voltage of the field effect transistors ~o as to produce a more nearly square wave at the output and also limit or preclude any shift in the threshold voltage with termperature thereby enhancing the stability of the calibration. A feedback resistor between the interconnected channel electrodes and the gate electrodes is provided to achie-ve a duty fractor of 50~, The chopper drive means fu~ther comprises a second pair of field effect transistors comprising first and second channel electrodes and a gate electrode respectively with the first channel electrodes interconnected and the second channel electrodes being connected directly to the source of regu-lated voltage. A second pair of field effect transistors produces a square ~LIDS()~
wave having a greater peak-to-peak voltage than the square wave output of the first-pair of field ef~ect transistors for use in driving the chopper. In or-der to minimize the power consumption, the secand pair of field effect tran-sistors are biased just above the threshold voltage of each transistor such that switching occurs at the zero crossing of the square wave generated by the first pair of field e~fect transistors. Since the second pair of field effect tT~nsistors are not on at the same time except for the instant of tr~nsition~
virtually all current used by the second pair of field effect transistors is needed to drive the chopper and power consumption due to wasted current is minimized.
It is another object of the invention to provide an output means which maintains the stable current output for all current levels representing the admittance measurement.
In the preferred embodiment of the invention, the output means com-prises an output a~plifier including a voltage feedback ne~work connected to a resistor through which the 4-20 milliamp DC current drawn by the two-wire transmitter flows so as to stabilize the flow of the 4-20 milliamp DC current at all current levels.
It is yet another object of this invention to provide a two-wire transmitter having a pair of terminials which may be interchangeably connected to the two t~ansmission wires without damaging or adversely affecting the two-wire transmitter.
In accordance with this object of the invention, the input of the two-wire transmitter includes a full wave rectifying bridge permitting current flow through one pair of diodes when the terminals are connected to the trans-mission wires with one polarity and current flows through the other pair of diodes when the terminals are connected to the transmission wires with the op-posite polarity.
It is also an object of this invention to provide for linear calibra-tion of the admittance measurement.
~OSO~
In accordance with this object of the invention, the bridge network includes a span capacitance across which the unbalance of the bridge network is measured where the span capacitance is substantially larger than the capaci-tance of the admittance being measured. In a particularly preferred embodi-ment of the invention, the span capacitance is at least 10 times and prefer-ably 25 times the capacitance of the admittance being measured.
It is a further object of this invention to provide for RF admittance measurements wherein the length of the cable connecting the probe electrode to the bridge network does not affect the measurement of the admittance.
In accordance with this object, the probe electrode may include a guard electrode juxtaposed to and shielding the probe electrode so as to main-tain the potential of the guard electrode at substantially the same potential as the probe electrode for a given operating point where the probe electrode is connected to one side of the span capacitance through the axial conductor of a coaxial conductor and the guard electrode is connected to the other side of the span capacitance through the shield of the coaxial conductor In accordance with an~herobject of this invention, the system may employ various types of probes including linear and non-linear immersion probes utilizing a guard electrode as well as a probe elect~ode.
In accordance with a still further object of this invention, the overall system is adapted for use in a battery operated mode or an AC supply mode.
Figure 1 is a block diagram of a two-wire transmitter embodying the invention;
Figure 2 is a schematic circuit diagram of an RF signal generator em-bodying one important aspect of the invention;
Bigures-~:2(a-c) are waveform diagrams utilized in describing the oper-ation of the circuit of Figure 2;
Figure 3 is 2 schematic circuit diagram of acchopper drive circuit embodying another important aspect of the invention;
~C~501~L4 Figure 4 is a schematic circuit diagram of an outpùt amplifier embo-dying another important aspect of the inven~ion;
Figure 5 is a schematic representation of the bridge network includ-ing a mechanical representation of the probe;
Figure 6 is an equivalent circuit of the bridge network of Figure 5;
~ igures 7(a-c) are schematic representations of various p~obes immer-sed in various materials;
Figures 8(a-c) are equivalent circuits of the admittance measured by the probes of Figures 7(a-c~ respectively;
Figure 9 is an equivalent circuit of the admittance of Figures 8~a-c);
and Figure 10 is a schematic diagram of a battery-powered output ampli-fier.
As shown in Figure 1, a two-w~re transmitter 10 is connected in se-ries with a power supply 12 and a load represented by a resistor 14 through transmission wires 16 and 18 connected to the terminals 20 and 22 of the two-wire transmitter 10. In accordance with this inventionJ the transmitter 10 is adapted to measure and draw a signal current representing an unknown measured admittance 24 which may represent the condition of materials sensed by the probe. The measured admittance 24 which represents the capacitance 24c and the resistance 24r from a probe electrode to ground forms one arm of a bridge network 26 also comprising a capacitor 28 and wlndings 30 and 32 of a secon-dary 34 in a tr~nsformer 36. The bridge network 26 is driven by an oscillator 38 having an output connected to the primary 40 of the transformer 36, In accordance with this invention, the voltage across the admittance 24 is limited to a level so as to assure adequate power for the two-wire trans-mitter in view of the power consumption by the unknown resistance 24r. As will now be described in detail, the voltage is limited to less than ~ where V is the voltage across the two-wire transmitter and the current drawn by the two-wire transmitter varies from 4-20 milliamps.
Heretofore, it has been assumed tha~ the unknown resistance 24r of the unknown admittance 24 being measured may vary over a wide range. ~f course, for a ~ixed voltage, if the resistance 24r should become very small, a good deal of power would be consumed in that resist~nce. In a conventional two-wire transmitter, the sole source of power is the current flow th~ough the transmission wires 16 and 18 which is conventionally at levels of 4-20 milli-amps. If it is assumed that the power supply produces an output voltage of 24 volts, the voltage across the terminals 20 and 22 of the two-wire tr~nsmit-ter may, for exampler be 12 ~olts where the total voltage drop across the load 14 plus the drop across ea~h of the wires 16 and 18 is 12 volts, This means that when the two-wire transmitter is drawing 4 milliamps, the total power available to operate the two-wire transmitter is P = VI = 48 milliwatts. This would mean that extremely small shunt resistances 24r would require extremely sma~ll .v'o~tages across the unknown admittance 24 to permit the two-wire trans-mitter to operate from the available power at the 4 milliamp level.
It has however been discovered, as will be described subsequently, that the resistance 24r, in almost all applications regardless of the type of probe utilized, will not fall below 500 ohms, Thus, by only moderately limit-ing the voltage across the unknown admittance 24 and thus the voltage across the unknown resistance 24r, sufficient power is available to the two-wire trans-mitter even at the 4 milliamp current level. Having once recognized that the magnitude of the resistance 24r will not, in almost all aplications, fall be-low 500 ohms, the magnitude of the voltage across the resistance 24r may be readily computed for a 4-20 milliamp two-wire transmitter from the following equation:
V2 ,C VIm (1) r-2-4 wh~re V = the voltage across the transmitterj v = the rms voltage across resistance 24r;
~LOS(~
Im = ~he minimum current flow through the two~wire transmitter 10;
and r24= the resistance in ohms of the resistance 24.
For I equal 4 milliamps and r24 equal 500 ohms, then v = ~ (2) If V equals 12 volts7 then v equals~F or less than 5 volts rms. Of course, the two-wire transmitter itself requires someppower to operate. Therefore, in the preferred embodiment where Im = 4 milliamps and V = l~ volts~ v = ap-proximately 2.2 volts rms, or substantially less th~nJ~.
In further accorda~ce with this invention, the oscilla~or 38 of the class C type, i.e.~ the collector current of eachaof the two transistors in the oscillator 38 which drive the tank circuit flows through an angle less than 180 of the 360 cycle of the RF sinusoidal signal applied to the bridge network 26. However, cl~s C operation may produce distortion in the inten-ded sinusoidal signal. Therefore, in further accordance with this invention, the oscillator 38 comprises a resonant circuit in the form o~ a tank circuit including the transformer 36 as well as the measured admittance 24 as will sub-sequently be described in detail with reference to Figure 2. Since the ad-mittance 24 is part of the resonant circuit, little additional current is re-quired to drive additional admittance between the probe and ground.
As also shown in ~igure 1, an AC error signal representing the unba-lance of the bridge network 26 and thus the u~known measured admittance 24 is applied to an error amplifier 42. The error amplifier 42 permits the use of \ relatively low AC voltages in the bridge ne~work 26 in accorda~ce with this in-vention. The output from the error amplifier 42 is then applied to a phase sensitive detector comprising a chopper 44 which is triggered by a chopper drive 46.
In accordance with another importan~ aspect of the invention, the bridge network 26 and the error amplifier 42 are isolated from the power sup-ply by the first transformer 36 and the second transformer 48 which couples the outp~t of the error amplifier 42 to the input of the chopper 44 In other ~LOS0~L~4 words, the power supply is allowed to float wi~h respect to the probe. This permits the use of a probe for measuring the admittance 24 between the probe electrode and ground without being concerned with the manner in which the power supply 12 is connected to ground. Note that this power supply 12 is at a re-mote location with respect to the two-wire transmitter 10 and the manner in which the power supply 12 is connected to ground may not be readily disce~i-ble at the two-wire tra~smitter 10. The isolation provided by the transfor-mers 36 and 48 also allows either terminal 20 or 22 of the two-wire transmit-ter 10 to be maintained at a very substantial AC or DC voltage with respect to 10 ground without any high voltage breakdown.
In order to provide ~-solation for the bridge network 26 while still providing a DC power supply for the error amplifier 42 which is directly coup-led to the bridge network 24, diodes 50 and 52 are provlded to rectify the RF
sinusoidal signal from the secondary 34 of the transformer 36. Diodes 50 and 52 are then connected to a terminal 54 of the amplifier 42 so as to provide a DC power supply therefore which is isolated from the power supply 12 In contrast, the DC power supply voltages for the RF oscillator 38, the chopper drive 46, the chopper 44 and an ou~tput amplifier 56 are provided by a voltage regulator 58 with a positive power supply terminal ~Vl In ad-20 dition, a negakive power supply voltage is provided by a voltage regulatingcircuit in the RF oscillator 38 at a terminal -V2, The chopper drive 46, the chopper 44 and the output amplifier 56 are also connected to the circuit com-mon terminal C of the voltage regulator 58.
In order to permit the bridge to be zeroed with a capacitance 24c from probe to ground which is differenttfrom the zeroing capacitance 28, the number of windings 30 differs from the number of windings 32 For example"
the number of windings 30 may be three times as large as the nulnber of windings 32 so as to allow the bridge to be zeroed when the measured capacitance 24c from probe to ground is three times as great as the zeroing capacitance 28 30 In addition, the bridge network 26 includes a variable span capacitor 60, By adjusting the span capacitor 60, the measured capacitance 24c necessary to pro-duce a predetermined current through the transmission wires 16 and 18 may be varied. In addition, the output amplifier 56 may be provided with a gain ad-~ustment which provideS fine span control.
In order to provide spark protection for the transmitter 10, a pair of series connected, reversed poled Zener diodes 62 and 64 are connected be-tween one terminal o~ the span capacitor 60 and ground. A neon bulb 66 is con-nected between the other terminal of the span c~apa--~itor 60 and ground. The protection afforded by the diodes 62 and 64 and the bulb 66 allow the trans-mitter 10 to withstand spikes ~ several thousandsffol~s across the admittance 24 with no component failure or unbalancing of the bridge network 26.
As also shown in Figure 1, a tap on the primary 68 of the transfor-mer 48 is connected to the input of thee~r~r amplifier 42. This connection provides feedback to the amplif~er 42 so as to control the gain thereof. Of course, changing the location of the tap 68 will change the gain of the ampli-fier 42 and thus the magnitude of the output applied to the chopper 44.
As theeQutput from the chopper 44 varies and is compared with the voltage across a res~stor 57 connected to the wire 22, the signal current out-put from the amplifier 56 is tansmitted through the wires 16 and 18. The cur-rent h&ving a magnitude which represents the admittance 24 and the condition o the materials being measured is utilized to drive the load 14.
In accordance with one aspect of the invention, the input of the two wire transmitter 10 comprises a fullwave rectifying bridge network comprising diode pairs 70 and 72 which conduct the 4-20 milliamp current when the terminal 20 is positive with respect to the terminal 22. Similarly,~the pair of diodes 74 and 76 conduct when the terminal 22 is positive with respect to the terminal 20 o~l22 to be connected to ei~her transmission wire without damaging or affec-ting the operation of the transmitter~
The class C RF oscillator will now be described in detail with refe-rence to Figure 2. The oscillator comprises a multivibrator such as a pulsed ~S0~14 amplifier including a pair of transistors 100 and 102 which are alternately cond~ctive so as to drive a resonant tank circui~ comprising the transformer 36 and a capacitor 104 which is connected in parallel with the primary 40 or the transformer 36 as well as the measured admittance A in the bridge net-work 26. The base drive for the transistor 100 of the multivibrator is pro-vided by the capacitor 106 and resistors 108 and 110 where the resistor lQ0 is connected to a transistor 112 in a base current regulating circuit. Simi-larly, a c~apacitor 114 and resistors 116 and 118 provideda base drive for the transistor 102. The base current of the transistors 100 and 102 charge the capacitors 106 and 114 to a positive voltage higher than the supply voltage thereby cutting off the transistors 100 and 102 during most of the cycle so as to achieve class C operation Diodes 120 and 122 which are connected in the base circuits of the transistors 100 and 102 respectively provide protec-tion ~or the bases of the transistors by blocking current flow when the junc-tion of the resistors 108 and 110 and the junction of the ~esistors 116 and 118 are driven positive.
As mentioned previously, the transistor 112 is part of a regulating circuit. The regulation afforded by the transistor 112 maintains the ampli-tude of the RF sinu-soidal signals substantially constant despite any chang~
in the operating characteristics of transistors within the oscillator and des-pite resistive loading due to the resistance 24r, In this connection, the '~
base of the transistor 112 is connected to a tap in the voltage divider com-prising resistors 124 and 126 with~one terminal of the voltage divider con-nected to the +Vl power supply terminal of the voltage regulator and the other terminal of the voltage divider connected to a capacitor 128 which is connec-ted to circuit common through a discharge resistor 130 which may be potted with the capacitor 128 to providedintrinsic safety.
The capacitor 128 is charged to a negative potential with respect to circuit conLmon by full wave rectifying diodes 127 and 129 connected across the tank circuit such that the tap of the voltage divider connected to the base of ~s~
the transistor 112 is maintained at an operating point of approximately zero volts which is just enough to render the collector-emitter circuit of the tran-sistor 112 conductive. The emitter of the transistor 112 is maintained slight-ly negative by a resistor 132 and a diode 134. Diode 134 compensates for the base emitter voltage of the transistor 112 and partially compensates for chan-ges in the base emitter îroltage of the transistor 112 with t~ rature so as to assure stable calibration. As clearly shown in Figure 2, the negative volt-age of the capacitor 128 is utilized to provide a negative power supply volt-age -V2 for the chopper 44 and the output amplifier 56 as shown in Figure 1.
The regulating circuit as previously described including the transis-tor 112 operates in the following manner to maintain the amplitude of the RF
sinusoidal signal at the transformer 36 substantially constant. The voltage across the transformer 36 which is the voltage across the tank circuit of the oscillator is, in effect, detected by the diodes 127 and 129 which charge the capacitor 128. The resulting negative DC voltage on the capacitor is then com-pared to the voltage of the regulator 48 at the resistive voltage divider com-prising the resistors 124 and 126 so as to maintain the intermediate tap at approximately circuit common, As the characteristics of the transistors chan-ge with temperature and the probe is resistively loaded as represented by the resistance 24r, the transistor 112 leaks bias off the capacitors 106 and 114 so as to maintain the ampl.itude of the oscill~tor and the corresponding voltage across the capacitor at the same potential.
In order to eliminate any distortion in the RF sinusoidal signal, a relatively large choke inductor 136 provides a high impedance load to the tank circuit thereby avoiding~any sharp current pulse which might distort the RF
sinusoidal waveform. An inductor 140 and a capacitor 142 provides a power sup-ply filter network.
The class C mode of operation for the oscillator 38 will now be des-cribed with reference to the waveforms of Figures 2(a-c). As shown in Figure 3Q 2a, the output voltage from the collector to circuit common which is applied -1~-1050~14 across the primary 40 of the transformer 36 is substantially sinusoidal due to the resonant action of the primary 40 with the capacitor 104 and the image of the bridge capacitors 24C and 28 (shown on Figure 6) reflected through trans-former 40. However, the diode 120 is biased off by the voltage o~ capacitor 106 for most of the cycle, producing a voltage pulse as shown in Figure 2C at the anode of diode 120. Thus, the collector current which flows through the transistor 100 is intermi~tent as shown in Figure ~b. In fact, only a brief surge of collector current flows assshown in Figure 2b during the 360 degree cycle depicted in Figure 2a. ~I~ actuality, some current continues to flow during the remainder of the cycle but this current is small relative to the surge of current flow and has not therefore been depicted in the drawing,) As shown in Figure 2b, the substantial or surge of collector current flows for substanti~lly less than 90 degrees of the 360 degree cycle which is of course substantially less than 180 degrees flow of current which still falls within the rsalm of class C operation. Note that the surge of current corresponds in time with the peak voltages for Figures 2a and 2C to assure that the maximum power is derived from the current flow.
As shown in Figures 1 and 2, the tank circuit is connected to the chopper drive 46 through a switch 144 which is c.apable of connecting the chop-per drive toweither terminal of the primary 40, By moving the switch frDm one position to the otherJ the phase of the chopper drive is reversed 180 degrees and the phase sensitive detection performed by the chopper 44 is changed by 180 degrees to permit the transmitter~"to operate in a high level or low level failsafe mode. As will now be described in detail with reference to Figure 3, the chopper drive 46 generates a square wave trigger signal for the chopper 44 while minimizing power consumption and optimizing stable, accurate calibration consistent with this invention.
To achieve these objectives, chopper drive 46 as sh~wn in Figure 3 comprises a first pair of field ef~ect transistors 200 and 202 having gate electrodes connected to the tank circuit through a capacitor 204. The first ~S6~
channel (drain) electrodes of the transistors 200 and 202 are interconnected and the second channel tsource) electrodes are connected between circuit com-mon and the regulated supply voltage +Vl. In accordance with the objectives of this invention, the second channel electrodes are connected to the power supply voltage ~Vl and circuit common through resistors 206 and 208.
The sinusoidal output from the oscillator 38 as shown in Figure 1 is applied to a capacitive divider network including the capacitor 204 and capacitors 228 and 230 connected between the capacitor 204 and circuit common.
The capacitively divided sinusoidal signal across the capacitors 228 and 230 is then applied to the gate electrodes of the transistors 200 and 202 to al-ternately gate the transistors between the conductive states.
It will be understood that the re~istors 206 and 208 play a particu-larly important role in assuring low power consumption and accuracy in the phase detection at the chopper 44. In this connection, it will be understood that the resistors 206 and 208 serve to limit the voltage across the channel electrodes of each of the transistors 200 and 202 which in turn sharpens the knee of the input voltage-output voltage transfer characteristics of the field effect transistors. As shown in curve a of Figure 3aJ large output voltages from channel-electrode-to-channel-electrode of a field effect transistor give a rounded knee to the output voltage-input voltage transfer characteristic while limiting the output voltage as shown in curve b sharpens the knee of the output voltage-input voltage characteristic. This tends to produce a more nearly square wave signal which is of the utmost importance in achieving,relia-bility in the ph~se detection at the chopper 44.
M~reover, as shown in Figure 3b, limiting the output voltage of chan nel electrode to channel electrode of the field effect transistor tends to im-munize the field efect transistor to ch~lges in the output voltage-input volt-age transfer characteristic with temperature. As shown in waveforms c and d of Figure 3b where~;curve c represents the output-input voltage characteristic at a temperature of -55C. and curve d represents the output-input voltage cha-racteristic at a temperature of ~25C. ~lus, a large channel electrode-to-channel-electrode voltage makes for a very substantial difference in curves c and d which affect the stability of the calibrations ~or the systemi On the other hand, limiting the output voltage as shown in curves e and f renders the -55C. curve e substantially identical to the ~25C. curve f.
In addition, the channel resistors tend to limit curren~ flow through the transistors 200 and 202 when the transistors 200 and 202 are simultaneous-ly conductive between the first and second channel electrodes. This assures that the power consumption by the transistors 200 and 202 will not be excessive as in the case where both of the transistors 200 and 202 conduct simul*aneously.
The output ~from the interconnected first channel electrodes is a square wave v~ltage riding above circuit common. In order to assure that the waveform is square, a feedback resistor 210 is provided between the first chan-nel electrodes and the gate electrode so as to raise the gate electrode to the average DC voltage at the first channel electrodes~ The resistor 210 assures a duty factor of 50% thereby compensating for small differences in the thres~
hold voltages of the field effect transistors. Capacitors 212 and 214 provide a low impedance to drive the gate capacitance of the succeeding stage with the square wave signal generated by the field effect transistors 200-and 202, Thus, the first state o~ the chopper drive generates a voltage waveform ~hich is square. However, the square voltage waveform is of insuffi-d~nt peak-to-peak voltage to drive the chopper because of the voltage drop across the channel resistors 206 and 208.
Therefore, the succeeding or second stage of the chopper drive, which is AC coupled to the preceding stage through capacitors 217 and 219, com-prises another or second pair of field effect transistors 216 and 218 which ^
are biased near their respective threshold voltages by resistors 220, 222 and 224 which are connected to the gate electrodes thereof. By biasing the transis-tors 216 and 218 near their threshold voltages the transistors turn on very near the zero crossing of th~ square wave signal generated by the transistors 1050~L14 200 and 202. As a result~ the ~uty factor of each of tllo transistors 216 and 218 more closely appro~ch~s 50~ thoroby c1ilninating ~ny ph~se uncortainty 50 as to assurc rc1iabl~ phasc dctcction at the choppcr 44. Since the transis-tors 216 and 218 do not conduct simultaneously except for the inst.~nt of tran-_ siticn~ there is littlc or no power wasted by the second stage.
Note that the transistors 216 and 218 are conn~cted directly across thc power supply voltage ~Vi and circuit common so that the ou put to thc c}-op-per 44 is alternately switched between IVl an~ circuit common. This produces a low output impedance in the chopp~r dri~e to assure fast rise and fall times ; 10 of the resulting s~uare wave output signal without the necessity of dissipating - large amoun~s of power in the chopper drive. Accordingly, the square wave out-put signal generated by the field effect transistors 216 and 218 connected be-tween the supply volta~e Vl and circuit common very closely approaches a per-fect square wave so as to assure phase stability in the phase sensitive detec-tion without sacrificing efficiency of the chopper drive.
~ ~Yhere a probe is utilized to measure the level of liquids and the - liquids tend to coa~ the probe, it is desirable to provide means by which the phasing of the chopper drive square wa~e signal may be altered by a 45 lead.
In this connection, it will be understood that long coatings on a probe às described in the aforesaid U~S. patent 3,706,980 appear as an infinite transmission line and the conductive and susceptive components of the coating are equal so as to produce a 45 lag. By detecting at a 45 phase angle, the conductive component and the susceptive component ~ill cancel leaving only the susceptance due to the change in capacit mce of the li~uid level being measured and no susceptance du~ to the coating itself. In this connection, capacitor 226 and series resistor 234 or the capacitor 228 may be optionally connected in parallelwith a capacitor 230.
In accordance with another important aspect of the invention, the output amplifier 56 comprises a vo1tage feed-back netwolk connected to a resis-tor 57 as shown in ~igure 1 through which the 4-20 millian~ DC current drawn .. . .. .
105al1~4 by the two-wire transmitter flows so as to stabilize the flow of the 4-20 mil-liamp DC current at all current levels. As shown in Figure 4,~the~output am-plifier 56 is divided into the following sections~ a voltage ~eedback divider network 300, a first differential amplifier stage 302, a second differential stage 30~, a voltage to current gain stage 306 and an output amplifier stage 308 which is shown as including the resistor 57 connected between circuit com-mon and the terminal 22 in Figure 1.
The voltage feedback divider network 300 includes an independent ~
point adjustment potentiometer 310 connected in series with resistors 312 and 314. A tap 316 on the potentiometer 310 is set so that when the bridge net-work 26 shown in Figure 1 is at balance, the current drawn by the two-wire tr~nsmitter is 4 milliamps when na current is flowing through the gain adjust-ment network compris~ng a potentiometer 318 in serres with a resistor 320 and having an adjustable tap 322 connected to the input of the first differential stage 302 through a resistor 324. When there is no current flowing through the gain adJustment network, the voltage with respect to circuit common C at the tap 322 remains at zero volts throughout the entire range of gain control.
The differential amplifier stage 302 comprises a first transistor 326 having a base connected to the output from the chopper 44 and the voltage feedback network 300. The base of a second transistor 330 is connected to circuit common C through a resistor 332. The differential amplifier stage 302 includes biasing resistors 334, 336 and 338 which are connected between the positive power supply terminal +Vl and the negative power supply terminal -V2, The second ampl~fier stage 304 comprises a first transistor 340 having a base connected to t~he collector of the transistor 328 and a second transistor 342 having a base connected to the collector of the transistor 330.
Biasing resistors 344, 346 and 348 are connected between the positive power supply terminal +Vl and circuit common.
The collectors of the transistors 340 and 342 are connected to the bases of a pair of transistors 350 and 352 of the voltage to current stage 306, 10~0114 The collector-emitter circuits of the transi~tors 350 and 352 are connected in series with a resistor 354 between the power supply terminal +Vl and the nega-tive power supply terminal -V2.
The output stage co~rises a pair of transistors 356 and 358 where the base of the transistor 356 is connected to the junction of the resistor 354 and the collector of the transistor 352 in the voltage to current gain stage 306. The output current from the output stage 308 is connected to the resistor 57 through a resistor 360. Resistors 362 and 364 connect the collec-tor and emitter of the resistors 356 and 358 respectively to the terminal 20 of the two-wire transmitter, ~hen an unbalance is created at the bridge network 26~ the voltage output from the chopper 44 increases which tends to make the base of the tran-sistor 328 more positive. This renders the transistor 328 more conductive and the transistor 330 less conductive which in turn causes the voltage at the col-lector of the transistor 328 to decrease and the voltage of the collector of the transistor 330 to rise, The voltages at the collectors of the transistors
It is an overa~l objec~ of this invention to monitor ~he condition of materials at a remote location.utilizing RF admittance measurements.
It is a more specific object of this invention to minimize the power consumption necessary in making the RF admittance measurements.
It is also a more specific object of the invention to provide an in-trinsically safe system for measuring RF admittance measurements.
It is a still more specific object of this invention to provide a two-wire transm~tter which is capable of operating from the power supplied by 4-20 milliamps of current which flows through the two transmission wires con-necting the two-wire transmitter to a remotely located power supply.
In accordance with these objects, a particularly preferred embodi-ment of the invention comprises an admittance sensing probe including a sensing electrode adapted to detect the admittance of materials for monitoring the con-dition of the materials, an RF signal generator and a bri~dge network coupled to the RF signal generator. The bridge network includes the admitbance detec-ted by the probe such that the unbalance of the network corresponds to the con-dition of the materials being monitored while the RF signal generator applies a voltage of less than the ~r~ rms across the admittance detected, where V is the voltage across the two-wire transmitter. Output means are coupled to `the bridge network for changing the current flow drawn by the two-wire transmitter from ~ milliamps to 20 milliamps in response to the unbalænce of the bridge net-work so as to represent the condition of the materials.
It is also a specific object of this invention to provide isolation between a floating power supply and ~he probe so as to permit the admittance of the materials to be measured between the sensing electrodes and a grounded mem-ber.
It is a further object of this invention to provide DC isolation 10S01~4 which is not subject to high voltage breakdown.
In accordance with these objects, the preferred embodiment of the in-vention comprises means for DC isolating the bridge network from the RF signal generator and the output means. The DC isolating means may comprise a first transformer having a primary connected to the RF signal generator and a secon-dary forming part of the bridge network. The DC isolating means may further comprise a second transformer having a primary connected to the bridge network and a secondary connected to the output means.
In order to increase the output vol~age from the bridge network be-fore application to the output means~ an amplifier me~ns may be coupled to the output of the bridge network. In order to maint~in isolation of the bridge network from the remainder of the t~o--wire transmitter and the remote power supply, the power supply for the amplifier is derived from the rectifying means coupled to the bridge network.
It is another object of this invention to provide for stable cali-bration of the admittance measurement.
In accordance with this object, the RF signal generator comprises an RF oscillator and a regulating circuit for the oscillator for maintain~ng the amplitude of the RF signal substantially constant. The regulating circuit comprises means for full wave rectifying the output of the oscillator and a capacitor coupled to the full wave rectifying means which is charged by current flow through the rectifying means. A voltage divider is connected between the capacitor and a control input of the oscillator so as to maintain the amplitu-de of the RF signal from the oscillator and the voltage across the capacitor substantially constant. By maintaining the amplit~de of the RF signal substan-tially constant despite changes in the ~operating characteristics of the trans-lstors within the oscillator and despite resistive loading from the sensing electrode of the probe to ground, stable calibration of the admittance measure-ment is attained.
In a particularly preferred embodiment, the RF oscillator comprises ~LOS01~1L4 a class C oscillator including a multi-vibrator and a resonant circuit so as to develop an undistorted RF sinusoidal signal while still limiting power con-sumption. The resonant circuit may ~omprise the aforesaid first trans~ormer and the admittance in the bridge network.
In further accordance with the object of stable calibration and mi-nimizing power consumption, the output means of the two-wire transmitter com-prises a phase-sensitive detector including a chopper and ac'chopper drive means for generating a chopper trigger signal for application to the chopper means.
The chopper drive means comprises a pair of field effect transistors having first and second channel electrodes and a gate electrode respectively and fur-ther comprises a pair of channel resistors. The first channel electrodes are interconnected and the second channel electrodes are connected to a source of regulated voltage through the channel resistors. The resonant circuit of the RF oscillator is coupled to the gate electrodes of each of the fi~d effect transistors so as to render the field effect transistors alternately conduct-ive. The channel resistors minimize power consumption by limiting the current flow through the pair of field effect transistors should the transistors be simultaneously conductive. In addition, the channel resistors which reduce the output voltage from channel e~lectrode-to-channel electrode provide~ a shar-per kaee in the input-output cur~e at the threshold voltage of the field effect transistors ~o as to produce a more nearly square wave at the output and also limit or preclude any shift in the threshold voltage with termperature thereby enhancing the stability of the calibration. A feedback resistor between the interconnected channel electrodes and the gate electrodes is provided to achie-ve a duty fractor of 50~, The chopper drive means fu~ther comprises a second pair of field effect transistors comprising first and second channel electrodes and a gate electrode respectively with the first channel electrodes interconnected and the second channel electrodes being connected directly to the source of regu-lated voltage. A second pair of field effect transistors produces a square ~LIDS()~
wave having a greater peak-to-peak voltage than the square wave output of the first-pair of field ef~ect transistors for use in driving the chopper. In or-der to minimize the power consumption, the secand pair of field effect tran-sistors are biased just above the threshold voltage of each transistor such that switching occurs at the zero crossing of the square wave generated by the first pair of field e~fect transistors. Since the second pair of field effect tT~nsistors are not on at the same time except for the instant of tr~nsition~
virtually all current used by the second pair of field effect transistors is needed to drive the chopper and power consumption due to wasted current is minimized.
It is another object of the invention to provide an output means which maintains the stable current output for all current levels representing the admittance measurement.
In the preferred embodiment of the invention, the output means com-prises an output a~plifier including a voltage feedback ne~work connected to a resistor through which the 4-20 milliamp DC current drawn by the two-wire transmitter flows so as to stabilize the flow of the 4-20 milliamp DC current at all current levels.
It is yet another object of this invention to provide a two-wire transmitter having a pair of terminials which may be interchangeably connected to the two t~ansmission wires without damaging or adversely affecting the two-wire transmitter.
In accordance with this object of the invention, the input of the two-wire transmitter includes a full wave rectifying bridge permitting current flow through one pair of diodes when the terminals are connected to the trans-mission wires with one polarity and current flows through the other pair of diodes when the terminals are connected to the transmission wires with the op-posite polarity.
It is also an object of this invention to provide for linear calibra-tion of the admittance measurement.
~OSO~
In accordance with this object of the invention, the bridge network includes a span capacitance across which the unbalance of the bridge network is measured where the span capacitance is substantially larger than the capaci-tance of the admittance being measured. In a particularly preferred embodi-ment of the invention, the span capacitance is at least 10 times and prefer-ably 25 times the capacitance of the admittance being measured.
It is a further object of this invention to provide for RF admittance measurements wherein the length of the cable connecting the probe electrode to the bridge network does not affect the measurement of the admittance.
In accordance with this object, the probe electrode may include a guard electrode juxtaposed to and shielding the probe electrode so as to main-tain the potential of the guard electrode at substantially the same potential as the probe electrode for a given operating point where the probe electrode is connected to one side of the span capacitance through the axial conductor of a coaxial conductor and the guard electrode is connected to the other side of the span capacitance through the shield of the coaxial conductor In accordance with an~herobject of this invention, the system may employ various types of probes including linear and non-linear immersion probes utilizing a guard electrode as well as a probe elect~ode.
In accordance with a still further object of this invention, the overall system is adapted for use in a battery operated mode or an AC supply mode.
Figure 1 is a block diagram of a two-wire transmitter embodying the invention;
Figure 2 is a schematic circuit diagram of an RF signal generator em-bodying one important aspect of the invention;
Bigures-~:2(a-c) are waveform diagrams utilized in describing the oper-ation of the circuit of Figure 2;
Figure 3 is 2 schematic circuit diagram of acchopper drive circuit embodying another important aspect of the invention;
~C~501~L4 Figure 4 is a schematic circuit diagram of an outpùt amplifier embo-dying another important aspect of the inven~ion;
Figure 5 is a schematic representation of the bridge network includ-ing a mechanical representation of the probe;
Figure 6 is an equivalent circuit of the bridge network of Figure 5;
~ igures 7(a-c) are schematic representations of various p~obes immer-sed in various materials;
Figures 8(a-c) are equivalent circuits of the admittance measured by the probes of Figures 7(a-c~ respectively;
Figure 9 is an equivalent circuit of the admittance of Figures 8~a-c);
and Figure 10 is a schematic diagram of a battery-powered output ampli-fier.
As shown in Figure 1, a two-w~re transmitter 10 is connected in se-ries with a power supply 12 and a load represented by a resistor 14 through transmission wires 16 and 18 connected to the terminals 20 and 22 of the two-wire transmitter 10. In accordance with this inventionJ the transmitter 10 is adapted to measure and draw a signal current representing an unknown measured admittance 24 which may represent the condition of materials sensed by the probe. The measured admittance 24 which represents the capacitance 24c and the resistance 24r from a probe electrode to ground forms one arm of a bridge network 26 also comprising a capacitor 28 and wlndings 30 and 32 of a secon-dary 34 in a tr~nsformer 36. The bridge network 26 is driven by an oscillator 38 having an output connected to the primary 40 of the transformer 36, In accordance with this invention, the voltage across the admittance 24 is limited to a level so as to assure adequate power for the two-wire trans-mitter in view of the power consumption by the unknown resistance 24r. As will now be described in detail, the voltage is limited to less than ~ where V is the voltage across the two-wire transmitter and the current drawn by the two-wire transmitter varies from 4-20 milliamps.
Heretofore, it has been assumed tha~ the unknown resistance 24r of the unknown admittance 24 being measured may vary over a wide range. ~f course, for a ~ixed voltage, if the resistance 24r should become very small, a good deal of power would be consumed in that resist~nce. In a conventional two-wire transmitter, the sole source of power is the current flow th~ough the transmission wires 16 and 18 which is conventionally at levels of 4-20 milli-amps. If it is assumed that the power supply produces an output voltage of 24 volts, the voltage across the terminals 20 and 22 of the two-wire tr~nsmit-ter may, for exampler be 12 ~olts where the total voltage drop across the load 14 plus the drop across ea~h of the wires 16 and 18 is 12 volts, This means that when the two-wire transmitter is drawing 4 milliamps, the total power available to operate the two-wire transmitter is P = VI = 48 milliwatts. This would mean that extremely small shunt resistances 24r would require extremely sma~ll .v'o~tages across the unknown admittance 24 to permit the two-wire trans-mitter to operate from the available power at the 4 milliamp level.
It has however been discovered, as will be described subsequently, that the resistance 24r, in almost all applications regardless of the type of probe utilized, will not fall below 500 ohms, Thus, by only moderately limit-ing the voltage across the unknown admittance 24 and thus the voltage across the unknown resistance 24r, sufficient power is available to the two-wire trans-mitter even at the 4 milliamp current level. Having once recognized that the magnitude of the resistance 24r will not, in almost all aplications, fall be-low 500 ohms, the magnitude of the voltage across the resistance 24r may be readily computed for a 4-20 milliamp two-wire transmitter from the following equation:
V2 ,C VIm (1) r-2-4 wh~re V = the voltage across the transmitterj v = the rms voltage across resistance 24r;
~LOS(~
Im = ~he minimum current flow through the two~wire transmitter 10;
and r24= the resistance in ohms of the resistance 24.
For I equal 4 milliamps and r24 equal 500 ohms, then v = ~ (2) If V equals 12 volts7 then v equals~F or less than 5 volts rms. Of course, the two-wire transmitter itself requires someppower to operate. Therefore, in the preferred embodiment where Im = 4 milliamps and V = l~ volts~ v = ap-proximately 2.2 volts rms, or substantially less th~nJ~.
In further accorda~ce with this invention, the oscilla~or 38 of the class C type, i.e.~ the collector current of eachaof the two transistors in the oscillator 38 which drive the tank circuit flows through an angle less than 180 of the 360 cycle of the RF sinusoidal signal applied to the bridge network 26. However, cl~s C operation may produce distortion in the inten-ded sinusoidal signal. Therefore, in further accordance with this invention, the oscillator 38 comprises a resonant circuit in the form o~ a tank circuit including the transformer 36 as well as the measured admittance 24 as will sub-sequently be described in detail with reference to Figure 2. Since the ad-mittance 24 is part of the resonant circuit, little additional current is re-quired to drive additional admittance between the probe and ground.
As also shown in ~igure 1, an AC error signal representing the unba-lance of the bridge network 26 and thus the u~known measured admittance 24 is applied to an error amplifier 42. The error amplifier 42 permits the use of \ relatively low AC voltages in the bridge ne~work 26 in accorda~ce with this in-vention. The output from the error amplifier 42 is then applied to a phase sensitive detector comprising a chopper 44 which is triggered by a chopper drive 46.
In accordance with another importan~ aspect of the invention, the bridge network 26 and the error amplifier 42 are isolated from the power sup-ply by the first transformer 36 and the second transformer 48 which couples the outp~t of the error amplifier 42 to the input of the chopper 44 In other ~LOS0~L~4 words, the power supply is allowed to float wi~h respect to the probe. This permits the use of a probe for measuring the admittance 24 between the probe electrode and ground without being concerned with the manner in which the power supply 12 is connected to ground. Note that this power supply 12 is at a re-mote location with respect to the two-wire transmitter 10 and the manner in which the power supply 12 is connected to ground may not be readily disce~i-ble at the two-wire tra~smitter 10. The isolation provided by the transfor-mers 36 and 48 also allows either terminal 20 or 22 of the two-wire transmit-ter 10 to be maintained at a very substantial AC or DC voltage with respect to 10 ground without any high voltage breakdown.
In order to provide ~-solation for the bridge network 26 while still providing a DC power supply for the error amplifier 42 which is directly coup-led to the bridge network 24, diodes 50 and 52 are provlded to rectify the RF
sinusoidal signal from the secondary 34 of the transformer 36. Diodes 50 and 52 are then connected to a terminal 54 of the amplifier 42 so as to provide a DC power supply therefore which is isolated from the power supply 12 In contrast, the DC power supply voltages for the RF oscillator 38, the chopper drive 46, the chopper 44 and an ou~tput amplifier 56 are provided by a voltage regulator 58 with a positive power supply terminal ~Vl In ad-20 dition, a negakive power supply voltage is provided by a voltage regulatingcircuit in the RF oscillator 38 at a terminal -V2, The chopper drive 46, the chopper 44 and the output amplifier 56 are also connected to the circuit com-mon terminal C of the voltage regulator 58.
In order to permit the bridge to be zeroed with a capacitance 24c from probe to ground which is differenttfrom the zeroing capacitance 28, the number of windings 30 differs from the number of windings 32 For example"
the number of windings 30 may be three times as large as the nulnber of windings 32 so as to allow the bridge to be zeroed when the measured capacitance 24c from probe to ground is three times as great as the zeroing capacitance 28 30 In addition, the bridge network 26 includes a variable span capacitor 60, By adjusting the span capacitor 60, the measured capacitance 24c necessary to pro-duce a predetermined current through the transmission wires 16 and 18 may be varied. In addition, the output amplifier 56 may be provided with a gain ad-~ustment which provideS fine span control.
In order to provide spark protection for the transmitter 10, a pair of series connected, reversed poled Zener diodes 62 and 64 are connected be-tween one terminal o~ the span capacitor 60 and ground. A neon bulb 66 is con-nected between the other terminal of the span c~apa--~itor 60 and ground. The protection afforded by the diodes 62 and 64 and the bulb 66 allow the trans-mitter 10 to withstand spikes ~ several thousandsffol~s across the admittance 24 with no component failure or unbalancing of the bridge network 26.
As also shown in Figure 1, a tap on the primary 68 of the transfor-mer 48 is connected to the input of thee~r~r amplifier 42. This connection provides feedback to the amplif~er 42 so as to control the gain thereof. Of course, changing the location of the tap 68 will change the gain of the ampli-fier 42 and thus the magnitude of the output applied to the chopper 44.
As theeQutput from the chopper 44 varies and is compared with the voltage across a res~stor 57 connected to the wire 22, the signal current out-put from the amplifier 56 is tansmitted through the wires 16 and 18. The cur-rent h&ving a magnitude which represents the admittance 24 and the condition o the materials being measured is utilized to drive the load 14.
In accordance with one aspect of the invention, the input of the two wire transmitter 10 comprises a fullwave rectifying bridge network comprising diode pairs 70 and 72 which conduct the 4-20 milliamp current when the terminal 20 is positive with respect to the terminal 22. Similarly,~the pair of diodes 74 and 76 conduct when the terminal 22 is positive with respect to the terminal 20 o~l22 to be connected to ei~her transmission wire without damaging or affec-ting the operation of the transmitter~
The class C RF oscillator will now be described in detail with refe-rence to Figure 2. The oscillator comprises a multivibrator such as a pulsed ~S0~14 amplifier including a pair of transistors 100 and 102 which are alternately cond~ctive so as to drive a resonant tank circui~ comprising the transformer 36 and a capacitor 104 which is connected in parallel with the primary 40 or the transformer 36 as well as the measured admittance A in the bridge net-work 26. The base drive for the transistor 100 of the multivibrator is pro-vided by the capacitor 106 and resistors 108 and 110 where the resistor lQ0 is connected to a transistor 112 in a base current regulating circuit. Simi-larly, a c~apacitor 114 and resistors 116 and 118 provideda base drive for the transistor 102. The base current of the transistors 100 and 102 charge the capacitors 106 and 114 to a positive voltage higher than the supply voltage thereby cutting off the transistors 100 and 102 during most of the cycle so as to achieve class C operation Diodes 120 and 122 which are connected in the base circuits of the transistors 100 and 102 respectively provide protec-tion ~or the bases of the transistors by blocking current flow when the junc-tion of the resistors 108 and 110 and the junction of the ~esistors 116 and 118 are driven positive.
As mentioned previously, the transistor 112 is part of a regulating circuit. The regulation afforded by the transistor 112 maintains the ampli-tude of the RF sinu-soidal signals substantially constant despite any chang~
in the operating characteristics of transistors within the oscillator and des-pite resistive loading due to the resistance 24r, In this connection, the '~
base of the transistor 112 is connected to a tap in the voltage divider com-prising resistors 124 and 126 with~one terminal of the voltage divider con-nected to the +Vl power supply terminal of the voltage regulator and the other terminal of the voltage divider connected to a capacitor 128 which is connec-ted to circuit common through a discharge resistor 130 which may be potted with the capacitor 128 to providedintrinsic safety.
The capacitor 128 is charged to a negative potential with respect to circuit conLmon by full wave rectifying diodes 127 and 129 connected across the tank circuit such that the tap of the voltage divider connected to the base of ~s~
the transistor 112 is maintained at an operating point of approximately zero volts which is just enough to render the collector-emitter circuit of the tran-sistor 112 conductive. The emitter of the transistor 112 is maintained slight-ly negative by a resistor 132 and a diode 134. Diode 134 compensates for the base emitter voltage of the transistor 112 and partially compensates for chan-ges in the base emitter îroltage of the transistor 112 with t~ rature so as to assure stable calibration. As clearly shown in Figure 2, the negative volt-age of the capacitor 128 is utilized to provide a negative power supply volt-age -V2 for the chopper 44 and the output amplifier 56 as shown in Figure 1.
The regulating circuit as previously described including the transis-tor 112 operates in the following manner to maintain the amplitude of the RF
sinusoidal signal at the transformer 36 substantially constant. The voltage across the transformer 36 which is the voltage across the tank circuit of the oscillator is, in effect, detected by the diodes 127 and 129 which charge the capacitor 128. The resulting negative DC voltage on the capacitor is then com-pared to the voltage of the regulator 48 at the resistive voltage divider com-prising the resistors 124 and 126 so as to maintain the intermediate tap at approximately circuit common, As the characteristics of the transistors chan-ge with temperature and the probe is resistively loaded as represented by the resistance 24r, the transistor 112 leaks bias off the capacitors 106 and 114 so as to maintain the ampl.itude of the oscill~tor and the corresponding voltage across the capacitor at the same potential.
In order to eliminate any distortion in the RF sinusoidal signal, a relatively large choke inductor 136 provides a high impedance load to the tank circuit thereby avoiding~any sharp current pulse which might distort the RF
sinusoidal waveform. An inductor 140 and a capacitor 142 provides a power sup-ply filter network.
The class C mode of operation for the oscillator 38 will now be des-cribed with reference to the waveforms of Figures 2(a-c). As shown in Figure 3Q 2a, the output voltage from the collector to circuit common which is applied -1~-1050~14 across the primary 40 of the transformer 36 is substantially sinusoidal due to the resonant action of the primary 40 with the capacitor 104 and the image of the bridge capacitors 24C and 28 (shown on Figure 6) reflected through trans-former 40. However, the diode 120 is biased off by the voltage o~ capacitor 106 for most of the cycle, producing a voltage pulse as shown in Figure 2C at the anode of diode 120. Thus, the collector current which flows through the transistor 100 is intermi~tent as shown in Figure ~b. In fact, only a brief surge of collector current flows assshown in Figure 2b during the 360 degree cycle depicted in Figure 2a. ~I~ actuality, some current continues to flow during the remainder of the cycle but this current is small relative to the surge of current flow and has not therefore been depicted in the drawing,) As shown in Figure 2b, the substantial or surge of collector current flows for substanti~lly less than 90 degrees of the 360 degree cycle which is of course substantially less than 180 degrees flow of current which still falls within the rsalm of class C operation. Note that the surge of current corresponds in time with the peak voltages for Figures 2a and 2C to assure that the maximum power is derived from the current flow.
As shown in Figures 1 and 2, the tank circuit is connected to the chopper drive 46 through a switch 144 which is c.apable of connecting the chop-per drive toweither terminal of the primary 40, By moving the switch frDm one position to the otherJ the phase of the chopper drive is reversed 180 degrees and the phase sensitive detection performed by the chopper 44 is changed by 180 degrees to permit the transmitter~"to operate in a high level or low level failsafe mode. As will now be described in detail with reference to Figure 3, the chopper drive 46 generates a square wave trigger signal for the chopper 44 while minimizing power consumption and optimizing stable, accurate calibration consistent with this invention.
To achieve these objectives, chopper drive 46 as sh~wn in Figure 3 comprises a first pair of field ef~ect transistors 200 and 202 having gate electrodes connected to the tank circuit through a capacitor 204. The first ~S6~
channel (drain) electrodes of the transistors 200 and 202 are interconnected and the second channel tsource) electrodes are connected between circuit com-mon and the regulated supply voltage +Vl. In accordance with the objectives of this invention, the second channel electrodes are connected to the power supply voltage ~Vl and circuit common through resistors 206 and 208.
The sinusoidal output from the oscillator 38 as shown in Figure 1 is applied to a capacitive divider network including the capacitor 204 and capacitors 228 and 230 connected between the capacitor 204 and circuit common.
The capacitively divided sinusoidal signal across the capacitors 228 and 230 is then applied to the gate electrodes of the transistors 200 and 202 to al-ternately gate the transistors between the conductive states.
It will be understood that the re~istors 206 and 208 play a particu-larly important role in assuring low power consumption and accuracy in the phase detection at the chopper 44. In this connection, it will be understood that the resistors 206 and 208 serve to limit the voltage across the channel electrodes of each of the transistors 200 and 202 which in turn sharpens the knee of the input voltage-output voltage transfer characteristics of the field effect transistors. As shown in curve a of Figure 3aJ large output voltages from channel-electrode-to-channel-electrode of a field effect transistor give a rounded knee to the output voltage-input voltage transfer characteristic while limiting the output voltage as shown in curve b sharpens the knee of the output voltage-input voltage characteristic. This tends to produce a more nearly square wave signal which is of the utmost importance in achieving,relia-bility in the ph~se detection at the chopper 44.
M~reover, as shown in Figure 3b, limiting the output voltage of chan nel electrode to channel electrode of the field effect transistor tends to im-munize the field efect transistor to ch~lges in the output voltage-input volt-age transfer characteristic with temperature. As shown in waveforms c and d of Figure 3b where~;curve c represents the output-input voltage characteristic at a temperature of -55C. and curve d represents the output-input voltage cha-racteristic at a temperature of ~25C. ~lus, a large channel electrode-to-channel-electrode voltage makes for a very substantial difference in curves c and d which affect the stability of the calibrations ~or the systemi On the other hand, limiting the output voltage as shown in curves e and f renders the -55C. curve e substantially identical to the ~25C. curve f.
In addition, the channel resistors tend to limit curren~ flow through the transistors 200 and 202 when the transistors 200 and 202 are simultaneous-ly conductive between the first and second channel electrodes. This assures that the power consumption by the transistors 200 and 202 will not be excessive as in the case where both of the transistors 200 and 202 conduct simul*aneously.
The output ~from the interconnected first channel electrodes is a square wave v~ltage riding above circuit common. In order to assure that the waveform is square, a feedback resistor 210 is provided between the first chan-nel electrodes and the gate electrode so as to raise the gate electrode to the average DC voltage at the first channel electrodes~ The resistor 210 assures a duty factor of 50% thereby compensating for small differences in the thres~
hold voltages of the field effect transistors. Capacitors 212 and 214 provide a low impedance to drive the gate capacitance of the succeeding stage with the square wave signal generated by the field effect transistors 200-and 202, Thus, the first state o~ the chopper drive generates a voltage waveform ~hich is square. However, the square voltage waveform is of insuffi-d~nt peak-to-peak voltage to drive the chopper because of the voltage drop across the channel resistors 206 and 208.
Therefore, the succeeding or second stage of the chopper drive, which is AC coupled to the preceding stage through capacitors 217 and 219, com-prises another or second pair of field effect transistors 216 and 218 which ^
are biased near their respective threshold voltages by resistors 220, 222 and 224 which are connected to the gate electrodes thereof. By biasing the transis-tors 216 and 218 near their threshold voltages the transistors turn on very near the zero crossing of th~ square wave signal generated by the transistors 1050~L14 200 and 202. As a result~ the ~uty factor of each of tllo transistors 216 and 218 more closely appro~ch~s 50~ thoroby c1ilninating ~ny ph~se uncortainty 50 as to assurc rc1iabl~ phasc dctcction at the choppcr 44. Since the transis-tors 216 and 218 do not conduct simultaneously except for the inst.~nt of tran-_ siticn~ there is littlc or no power wasted by the second stage.
Note that the transistors 216 and 218 are conn~cted directly across thc power supply voltage ~Vi and circuit common so that the ou put to thc c}-op-per 44 is alternately switched between IVl an~ circuit common. This produces a low output impedance in the chopp~r dri~e to assure fast rise and fall times ; 10 of the resulting s~uare wave output signal without the necessity of dissipating - large amoun~s of power in the chopper drive. Accordingly, the square wave out-put signal generated by the field effect transistors 216 and 218 connected be-tween the supply volta~e Vl and circuit common very closely approaches a per-fect square wave so as to assure phase stability in the phase sensitive detec-tion without sacrificing efficiency of the chopper drive.
~ ~Yhere a probe is utilized to measure the level of liquids and the - liquids tend to coa~ the probe, it is desirable to provide means by which the phasing of the chopper drive square wa~e signal may be altered by a 45 lead.
In this connection, it will be understood that long coatings on a probe às described in the aforesaid U~S. patent 3,706,980 appear as an infinite transmission line and the conductive and susceptive components of the coating are equal so as to produce a 45 lag. By detecting at a 45 phase angle, the conductive component and the susceptive component ~ill cancel leaving only the susceptance due to the change in capacit mce of the li~uid level being measured and no susceptance du~ to the coating itself. In this connection, capacitor 226 and series resistor 234 or the capacitor 228 may be optionally connected in parallelwith a capacitor 230.
In accordance with another important aspect of the invention, the output amplifier 56 comprises a vo1tage feed-back netwolk connected to a resis-tor 57 as shown in ~igure 1 through which the 4-20 millian~ DC current drawn .. . .. .
105al1~4 by the two-wire transmitter flows so as to stabilize the flow of the 4-20 mil-liamp DC current at all current levels. As shown in Figure 4,~the~output am-plifier 56 is divided into the following sections~ a voltage ~eedback divider network 300, a first differential amplifier stage 302, a second differential stage 30~, a voltage to current gain stage 306 and an output amplifier stage 308 which is shown as including the resistor 57 connected between circuit com-mon and the terminal 22 in Figure 1.
The voltage feedback divider network 300 includes an independent ~
point adjustment potentiometer 310 connected in series with resistors 312 and 314. A tap 316 on the potentiometer 310 is set so that when the bridge net-work 26 shown in Figure 1 is at balance, the current drawn by the two-wire tr~nsmitter is 4 milliamps when na current is flowing through the gain adjust-ment network compris~ng a potentiometer 318 in serres with a resistor 320 and having an adjustable tap 322 connected to the input of the first differential stage 302 through a resistor 324. When there is no current flowing through the gain adJustment network, the voltage with respect to circuit common C at the tap 322 remains at zero volts throughout the entire range of gain control.
The differential amplifier stage 302 comprises a first transistor 326 having a base connected to the output from the chopper 44 and the voltage feedback network 300. The base of a second transistor 330 is connected to circuit common C through a resistor 332. The differential amplifier stage 302 includes biasing resistors 334, 336 and 338 which are connected between the positive power supply terminal +Vl and the negative power supply terminal -V2, The second ampl~fier stage 304 comprises a first transistor 340 having a base connected to t~he collector of the transistor 328 and a second transistor 342 having a base connected to the collector of the transistor 330.
Biasing resistors 344, 346 and 348 are connected between the positive power supply terminal +Vl and circuit common.
The collectors of the transistors 340 and 342 are connected to the bases of a pair of transistors 350 and 352 of the voltage to current stage 306, 10~0114 The collector-emitter circuits of the transi~tors 350 and 352 are connected in series with a resistor 354 between the power supply terminal +Vl and the nega-tive power supply terminal -V2.
The output stage co~rises a pair of transistors 356 and 358 where the base of the transistor 356 is connected to the junction of the resistor 354 and the collector of the transistor 352 in the voltage to current gain stage 306. The output current from the output stage 308 is connected to the resistor 57 through a resistor 360. Resistors 362 and 364 connect the collec-tor and emitter of the resistors 356 and 358 respectively to the terminal 20 of the two-wire transmitter, ~hen an unbalance is created at the bridge network 26~ the voltage output from the chopper 44 increases which tends to make the base of the tran-sistor 328 more positive. This renders the transistor 328 more conductive and the transistor 330 less conductive which in turn causes the voltage at the col-lector of the transistor 328 to decrease and the voltage of the collector of the transistor 330 to rise, The voltages at the collectors of the transistors
3~8 and 330 are then applied as input to the bases of the transistors 340 and 342 causing the voltages at the collectors of the transistors 340 and 342 to increase and decrease respectively,'~ This in turn causes the transistors 350 and 352 to become more conductive and increase the current flow through the re-sistor 354 thereby raising the base of the transistor 356 to a more positive voltage causing an increase in current flow from the output transistors 356 and 358.
Since all of the current from the output transistors 356 and 358 flows through the resistor S7, the voltage across the resistor 357 will increa-se with increasing c~rrent flow due to the unbalance of the bridge network thereby decreasing the voltage at the terminal 22 with respect to circuit com-mon C. This in turn increases the negative lvoltage which is applied to the base of the transistor 328 through the voltage feedback divider network until that vol~cage is again zero volts thereby establishing a stable condition at the higher output current.
1050~14 From the foregoing, it sh~ld be understood that the output ampli~ier 56 may be analogized to an operational amplifier having one input at the base of transistor 328 acting as a summ4ng junction for the voltage from the output of the chopper 44 and the voltage of the voltage feedback divider network 3aO
and the other input at the base of the transistor connected to circuit common.
In accordance with another important aspect of the invention, the length of the cables associated with the probe will not affect the admittance measurements.
As shown in Figure 5, a probe 400 is connected into the bridge net-work 26, The probe 400 includes a guard electrode 410 juxtaposed to and sur-rounding a probe electrode 412. 3~nSu]ation 414 surrounds the probe electrode 412 so as to insulate the guard electrode 410 from the probe electrode 412 and the guard electrode 410 from a grounded conductive vessel 418. A coaxial cable is utilized to connect the probe 400 into the bridge network 402 where the shield of the cable 420 is connected to the guard electrode 410 at one terminal of the span capacitor 60 and the axial conductor 422 connects the probe electrode 412 to the vther terminal of the span capacitor 60.
Reference to Figure 6, wherein the equivalent circuit o~ Figure 5 is shown, reveals that a variation in the cable length will have no effect on the admittance measurement. As shown, the probe electrode to ground admittance 24 is represented by a capacitance 24c and a resistance 24r. Since the axial conductor 422 is surrounded by the coaxial shield 420 which is connected to the opposite terminal of the span capacitance 60, any admittance between the coaxial shield 420 and the axial conductor 422 will be connected across the span capacitance 60 and will not affect the balance or unbalance of the bridge network. Similarly, any admittance between the coaxial shield 420 and ground as represented by a capacitance 426c and a resistance 426r will have no effect on the balance of the bridge network 26 since this admittance is in parallel with the secondary 24 of the transformer, In accordance with another important aspect of the invention, linear ~10~0~4 c~ r~ltion of the ~dl~ittance ~ suring system is ~ ieve~ ~y m~king th~ sp~n capacital-ce 60 l~rge rclativ~ to the cal~<lcit~nce of the ~mittance being measured as disclosed in United States patent 3,778,705 - Maltby, issued December 11, 1973. Preferably, the capacitance of th~ span capacitor 408 or the span capacit~r 26 is at least 10 times the capacitance of capacitance 424c or capacitance 24c. In a particularly preferred emkodiment, the span capacitanceis 25 times the capacit~nce beLng measured.
As shown in Figurc S, the pro'~e 400 comprises a probe elcctrode 412 which is completely surrounded with insulation 414, As also shown, the insu-lation 414 is coated with materials 428 contained within the vessel ~18. As - will now be explaincd, the probe electrode-to-ground resistance 24r will, in substantially all applications, be in excess of the previously mentioned 500 ohms even when the probe is covered with a coating 428 of conductive liquid ; 429 as shown in Figure 5.
Referring now to Figure 7a, a s~lematic representation oE the coat-ing 428 on the probe 400 of Figure S illustrates the nature of the probe-to-s ground resistance.- As shown therel the coating 428 may be represented by a series of small resistors 430 which are coex:tensive with the length of the coating. The j~ctian of these resistors 430 are connected to the probe elect-rode 414 by shunt capacitors 432 which represent the capacitance of the insu-lation 414. An equivalent circuit corresponding to the probe and coating of Figure 7a is illustrated in Figure 8a wherein the capacitor 432 is connected in shunt with the resistor 430, A capacitor 434 represents the capacitance through the insulation 414 from the conductive liquid below the coating 428 to the probe electrode 412. This equivalent circuit may in turn be represented as sl~own in l:igure 9 by the shunt resistor 424r and the shunt capacitor 424c.
It has been fo~ld that in substantially all applications where the resistance 424r as shown in Fîgure 9 is contr;l~uted by the coating 428 as represented by the series o resistors 430 shown in Figure 7a, the resistance 424r is more than 500 ohms.
Fi~ure 7~ r~rresents tlle in~Ul.lt~ prol)e 400 of Figurs S immersed in ~ s~n~i-con~uctive li(lui~ t~herein the li~ l itself is r~presented by ~ num-~er o sl~ult cai~citors ~6 ~1~ shunt resistors 438. 111e equivalent circuit for the i~umr~e~ pro~e o~ l:igure 7b is shown in Figure ~b wherein tl~c shunt capacitors 436 ~nd the shunt resistors 438 ar~ connect~d in par~llel and a ca-pacitor 434 again rc~resents the capacitanc~ through tllC insul~tion from the materials to the probc electro~e 412 The equivalent circuit of Figurc ~b m~y ' of course ~lso be depicted as a shunt resistor-c~pacitor co~ina~ion as shown in ~igure 9. Although the resistor 438 is now contributed by the semi-con-ductive material rath~r than ~hc coating as in the immersed probe of Figure 7a, it has neverthelcss been found that the equivalent resistance 42~r as depicted in Figu~e 9 will, iII substantially all cases, exceed 500 ohms for the immersed probc of Figure 7b. ~
Finally, Figure 7c dcpicts a bare probe 440 i~nmersed in semi-conduc-tive materials which may be represented by shunt capacitors 436 and shunt resis- ;
tors 438 which are depicted in schematic d ~cuit form by a resistance 442 and ;
a resistance 444 in Figure 8c. Once again, it has been found that the resist-ance 444 which represents the resistance 424r ~f Figure 9 in the bridge net~
work will exceed 500 ohms for almost all applications.
O As described in the foregoing, the invention may be utilized withinsulated as well as bare immersions probes including guard electrodes of the type described in U.S. patent 3,879,644 which issued on April 22, 1975. It will of course be appreciated that the invention is equally applicable to ~w~
terminal probes without a guard electrode. It will also be understood that the invention is applicable to non-linear probes wherein the probe electrode is characterized, i.e., the cross-sectional dimension of the probe electrode vaLies from one end o~ the probe electrode to the other. Probes of this type are disclosed in United States patent ~o. 3,269,180 issued August 30, 1966 to Schreiber which discloses a non-linear Ero'ce with~ut a guard electrcce and a -23~
., ~0501~4 non-linear probe with a guard elect~ode. Further~re, ~e ~nvention is apl?licableto non-iI[~rsible pro~es ~ich se~se the oc)ndition of an a~nittance material whe~ in close proximity therewith.
In the foregoing> the invention has becn dcscribcd in terms of a two-wire transnLittcr. It will of course be appreciated that many aspects of thc invelltion may be embodied in otller applications such as, for example a battery powered system, tYherein the power available is as limited if not more limited than the two-wire transmitter application.
In this connection, ano~her output amplifier 56 for use in a battery powered system will no~Y bc described with reference to Figure 10. As shown there, the output amplifier is in many respects similar to the output ampli-fier sho~m in Figure 4 and substantially iclentical circuit elements bear iden-tical reference characters.
However, the output amplifier of Figure 9 differs in that the voltage feedback from the resistor 57 is not applied to a summing junction in the first differential amplifier stage but rather to the other input o the differential ; amplifier at the base of the transistor 330. The output signal is represented by the current flow to and from output terminals 520 and 522 at the terminals of a diode 524 in the collector-emitter circuit of the transistor 358, In operation, a positive input at the base o the transistor 328 and a first differential a~plifier stage tends to increase the current flow through the resistor 57 This in turn raiscs the positive voltage applied to the base of the transistQr 3~0 of the voltage divider nctwork con~rising the resistors 310, 312 and a rcsistor 526. As a result, the current through thc rcsistor 57 and the output current terminals 520 ~ld 522 is stabilizcd at a highcr currcnt l~vc 1 .
It shoulcl be un(lsrstoo~l that thc output ampli~ier described is in cffcc~ ~sl ol)eratiol~al ~mplifier having one input conrlected to tile outyut of 105~)114 the chopper and the other input connected to a voltage feedback network as contrasted with the circuit of Figure ~ wherein one inyut served as a summing junction connected to the chopper output as well as the voltage feedback net-work and the other input was connected to circuit common.
Although the chopper 44 has not been ~hown in detail, it will be un-derstood that the chopper circuits and output amplifier circuits well known in the art are suitable for use in the two-wire transmitter s~s~em of this in-vention. For example, the chopper circuit disclosed in the aforesaid United States patent No. 3,778,7~5 may be utiliz~d. The output amplifier may com-prise ~ny o a number of commercially available differential amplifiers. It will also be understood that various resonant circuits may be utilized to re-place the tank circuit shown in Figure l.SiSimilarly, the voltage regulator circuit 58 may comprise a prior art voltage regulator well known in the art.
Although a preferred embodiment o~ the invention has been shown and ; described, it will be understood that various modifications may be made with-out departing from the true spi~it and scope of the invention as set forth in the appended claims.
Since all of the current from the output transistors 356 and 358 flows through the resistor S7, the voltage across the resistor 357 will increa-se with increasing c~rrent flow due to the unbalance of the bridge network thereby decreasing the voltage at the terminal 22 with respect to circuit com-mon C. This in turn increases the negative lvoltage which is applied to the base of the transistor 328 through the voltage feedback divider network until that vol~cage is again zero volts thereby establishing a stable condition at the higher output current.
1050~14 From the foregoing, it sh~ld be understood that the output ampli~ier 56 may be analogized to an operational amplifier having one input at the base of transistor 328 acting as a summ4ng junction for the voltage from the output of the chopper 44 and the voltage of the voltage feedback divider network 3aO
and the other input at the base of the transistor connected to circuit common.
In accordance with another important aspect of the invention, the length of the cables associated with the probe will not affect the admittance measurements.
As shown in Figure 5, a probe 400 is connected into the bridge net-work 26, The probe 400 includes a guard electrode 410 juxtaposed to and sur-rounding a probe electrode 412. 3~nSu]ation 414 surrounds the probe electrode 412 so as to insulate the guard electrode 410 from the probe electrode 412 and the guard electrode 410 from a grounded conductive vessel 418. A coaxial cable is utilized to connect the probe 400 into the bridge network 402 where the shield of the cable 420 is connected to the guard electrode 410 at one terminal of the span capacitor 60 and the axial conductor 422 connects the probe electrode 412 to the vther terminal of the span capacitor 60.
Reference to Figure 6, wherein the equivalent circuit o~ Figure 5 is shown, reveals that a variation in the cable length will have no effect on the admittance measurement. As shown, the probe electrode to ground admittance 24 is represented by a capacitance 24c and a resistance 24r. Since the axial conductor 422 is surrounded by the coaxial shield 420 which is connected to the opposite terminal of the span capacitance 60, any admittance between the coaxial shield 420 and the axial conductor 422 will be connected across the span capacitance 60 and will not affect the balance or unbalance of the bridge network. Similarly, any admittance between the coaxial shield 420 and ground as represented by a capacitance 426c and a resistance 426r will have no effect on the balance of the bridge network 26 since this admittance is in parallel with the secondary 24 of the transformer, In accordance with another important aspect of the invention, linear ~10~0~4 c~ r~ltion of the ~dl~ittance ~ suring system is ~ ieve~ ~y m~king th~ sp~n capacital-ce 60 l~rge rclativ~ to the cal~<lcit~nce of the ~mittance being measured as disclosed in United States patent 3,778,705 - Maltby, issued December 11, 1973. Preferably, the capacitance of th~ span capacitor 408 or the span capacit~r 26 is at least 10 times the capacitance of capacitance 424c or capacitance 24c. In a particularly preferred emkodiment, the span capacitanceis 25 times the capacit~nce beLng measured.
As shown in Figurc S, the pro'~e 400 comprises a probe elcctrode 412 which is completely surrounded with insulation 414, As also shown, the insu-lation 414 is coated with materials 428 contained within the vessel ~18. As - will now be explaincd, the probe electrode-to-ground resistance 24r will, in substantially all applications, be in excess of the previously mentioned 500 ohms even when the probe is covered with a coating 428 of conductive liquid ; 429 as shown in Figure 5.
Referring now to Figure 7a, a s~lematic representation oE the coat-ing 428 on the probe 400 of Figure S illustrates the nature of the probe-to-s ground resistance.- As shown therel the coating 428 may be represented by a series of small resistors 430 which are coex:tensive with the length of the coating. The j~ctian of these resistors 430 are connected to the probe elect-rode 414 by shunt capacitors 432 which represent the capacitance of the insu-lation 414. An equivalent circuit corresponding to the probe and coating of Figure 7a is illustrated in Figure 8a wherein the capacitor 432 is connected in shunt with the resistor 430, A capacitor 434 represents the capacitance through the insulation 414 from the conductive liquid below the coating 428 to the probe electrode 412. This equivalent circuit may in turn be represented as sl~own in l:igure 9 by the shunt resistor 424r and the shunt capacitor 424c.
It has been fo~ld that in substantially all applications where the resistance 424r as shown in Fîgure 9 is contr;l~uted by the coating 428 as represented by the series o resistors 430 shown in Figure 7a, the resistance 424r is more than 500 ohms.
Fi~ure 7~ r~rresents tlle in~Ul.lt~ prol)e 400 of Figurs S immersed in ~ s~n~i-con~uctive li(lui~ t~herein the li~ l itself is r~presented by ~ num-~er o sl~ult cai~citors ~6 ~1~ shunt resistors 438. 111e equivalent circuit for the i~umr~e~ pro~e o~ l:igure 7b is shown in Figure ~b wherein tl~c shunt capacitors 436 ~nd the shunt resistors 438 ar~ connect~d in par~llel and a ca-pacitor 434 again rc~resents the capacitanc~ through tllC insul~tion from the materials to the probc electro~e 412 The equivalent circuit of Figurc ~b m~y ' of course ~lso be depicted as a shunt resistor-c~pacitor co~ina~ion as shown in ~igure 9. Although the resistor 438 is now contributed by the semi-con-ductive material rath~r than ~hc coating as in the immersed probe of Figure 7a, it has neverthelcss been found that the equivalent resistance 42~r as depicted in Figu~e 9 will, iII substantially all cases, exceed 500 ohms for the immersed probc of Figure 7b. ~
Finally, Figure 7c dcpicts a bare probe 440 i~nmersed in semi-conduc-tive materials which may be represented by shunt capacitors 436 and shunt resis- ;
tors 438 which are depicted in schematic d ~cuit form by a resistance 442 and ;
a resistance 444 in Figure 8c. Once again, it has been found that the resist-ance 444 which represents the resistance 424r ~f Figure 9 in the bridge net~
work will exceed 500 ohms for almost all applications.
O As described in the foregoing, the invention may be utilized withinsulated as well as bare immersions probes including guard electrodes of the type described in U.S. patent 3,879,644 which issued on April 22, 1975. It will of course be appreciated that the invention is equally applicable to ~w~
terminal probes without a guard electrode. It will also be understood that the invention is applicable to non-linear probes wherein the probe electrode is characterized, i.e., the cross-sectional dimension of the probe electrode vaLies from one end o~ the probe electrode to the other. Probes of this type are disclosed in United States patent ~o. 3,269,180 issued August 30, 1966 to Schreiber which discloses a non-linear Ero'ce with~ut a guard electrcce and a -23~
., ~0501~4 non-linear probe with a guard elect~ode. Further~re, ~e ~nvention is apl?licableto non-iI[~rsible pro~es ~ich se~se the oc)ndition of an a~nittance material whe~ in close proximity therewith.
In the foregoing> the invention has becn dcscribcd in terms of a two-wire transnLittcr. It will of course be appreciated that many aspects of thc invelltion may be embodied in otller applications such as, for example a battery powered system, tYherein the power available is as limited if not more limited than the two-wire transmitter application.
In this connection, ano~her output amplifier 56 for use in a battery powered system will no~Y bc described with reference to Figure 10. As shown there, the output amplifier is in many respects similar to the output ampli-fier sho~m in Figure 4 and substantially iclentical circuit elements bear iden-tical reference characters.
However, the output amplifier of Figure 9 differs in that the voltage feedback from the resistor 57 is not applied to a summing junction in the first differential amplifier stage but rather to the other input o the differential ; amplifier at the base of the transistor 330. The output signal is represented by the current flow to and from output terminals 520 and 522 at the terminals of a diode 524 in the collector-emitter circuit of the transistor 358, In operation, a positive input at the base o the transistor 328 and a first differential a~plifier stage tends to increase the current flow through the resistor 57 This in turn raiscs the positive voltage applied to the base of the transistQr 3~0 of the voltage divider nctwork con~rising the resistors 310, 312 and a rcsistor 526. As a result, the current through thc rcsistor 57 and the output current terminals 520 ~ld 522 is stabilizcd at a highcr currcnt l~vc 1 .
It shoulcl be un(lsrstoo~l that thc output ampli~ier described is in cffcc~ ~sl ol)eratiol~al ~mplifier having one input conrlected to tile outyut of 105~)114 the chopper and the other input connected to a voltage feedback network as contrasted with the circuit of Figure ~ wherein one inyut served as a summing junction connected to the chopper output as well as the voltage feedback net-work and the other input was connected to circuit common.
Although the chopper 44 has not been ~hown in detail, it will be un-derstood that the chopper circuits and output amplifier circuits well known in the art are suitable for use in the two-wire transmitter s~s~em of this in-vention. For example, the chopper circuit disclosed in the aforesaid United States patent No. 3,778,7~5 may be utiliz~d. The output amplifier may com-prise ~ny o a number of commercially available differential amplifiers. It will also be understood that various resonant circuits may be utilized to re-place the tank circuit shown in Figure l.SiSimilarly, the voltage regulator circuit 58 may comprise a prior art voltage regulator well known in the art.
Although a preferred embodiment o~ the invention has been shown and ; described, it will be understood that various modifications may be made with-out departing from the true spi~it and scope of the invention as set forth in the appended claims.
Claims (26)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. In a two-wire transmitter system comprising a power supply and a load at one location and a two-wire transmitter at another location intercon-nected by a pair of transmission lines carrying a current ranging from 4 milli-amps at a minimum to 20 milliamps at a maximum and powered solely by said power supply, the improvement residing in said two-wire transmitter comprising: an admittance sensing probe including a probe electrode adapted to detect the ad-mittance of materials for monitoring the condition of the materials; an RF sig-nal generator; a bridge network coupled to said RF signal generator, said brid-ge network including the admittance detected by said probe such that the unba-lance of the network corresponds to the condition of the materials being moni-tored, said RF signal generator applying a voltage of less than the ? rms across the admittance detected where V is the voltage across the two-wire transmitter; and output means coupled to said bridge network for changing the current flow through the transmission lines from 4 milliamps to 20 milliamps in response to the unbalance of the bridge network so as to represent the con-dition of the materials.
2, The two-wire transmitter system of claim 1 including means for DC
isolating said bridge network from said RF signal generator and said output means.
isolating said bridge network from said RF signal generator and said output means.
3. The two-wire transmitter system of claim 2 including amplifier means DC coupled to said bridge network and DC isolated from said RF signal genera-tor and said output means.
4. The two-wire transmitter system of claim 3 including rectifying means coupled between said amplifier means and said bridge network for provid-ing a DC power supply voltage to said amplifier while maintaining isolation be-tween said amplifier and said power supply.
5. The two-wire transmitter system of claim 1 wherein said RF signal generator comprises an RF oscillator of the class C type including a resonant circuit so as to generate an RF signal characterized by a frequency equal to the resonant frequency of the resonant circuit.
6. The two-wire transmitter system of claim 5 wherein said DC isola-tion means comprises a transformer coupling the output of said oscillator to said bridge network, said transformer having a primary winding and a secondary winding, said primary winding, said secondary winding and the detected admit-tance forming part of said resonant circuit.
7. The two-wire transmitter system of claim 1 wherein said RF signal generator comprises an RF oscillator and a regulating circuit for said oscil-lator for maintaining the amplitude of said RF sinusoidal signals substantial-ly constant despite resistive loading.
8. The two-wire transmitter system of claim 7 wherein said transmitter further comprises a voltage regulator and said regulating circuit includes means for full wave rectifying the output of said oscillator, a capacitor coupled to said full wave rectifying means and charged by current flow through said rectifying means, a voltage divider connected between said capacitor and said voltage regulator and control means coupled to said voltage divider and said oscillator so as to maintain the amplitude of said RF signal and the vol-tage across said capacitor substantially constant regardless of resistive loading across the detected admittance.
9. The two-wire transmitter system of claim 1 wherein said bridge net-work includes a span capacitance substantially greater than the capacitive com-ponent of the admittance detected by said probe so as to assure linear cali-bration of said transmitter.
10. The two-wire transmitter system of claim 1 wherein said output means comprises: chopper means; and chopper drive means having an input coupled to said oscillator and an output coupled to said chopper for applying a chopper trigger signal to said chopper.
11. The two-wire transmitter system of claim 10 wherein said chopper drive means comprises a first pair of field effect transistors comprising first and second channel electrodes and a gate electrode respectively, and a pair of channel resistors, said first channel electrodes being interconnected and said second channel electrodes being connected across a source of voltage through said channel resistors so as to apply a reduced voltage across said first pair of field effect transistors, said RF signal generator being coupled to the gate electrodes of each of said field effect transistors so as to ren-der said field effect transistors alternately conductive thereby generating a substantially square wave signal.
12. The two-wire transmitter system of claim 11 wherein said chopper drive means further comprises another pair of field effect transistors compri-sing first and second channel electrodes and a gate electrode respectively, said pair of channel electrodes being interconnected, said second channel elec-trodes being connected across said source of voltage and said gate electrode being connected to said first pair of transistors, said chopper drive means further comprising means for biasing said other pair of field effect transis-tors near the threshold values thereof such that the square wave signal gene-rated by said first pair of field effect transistors is capable of turning said other pair of field effect transistors on at or near the zero crossing of said square wave signal without substantial simultaneous conduction of said other pair of transistors.
13. The two-wire transmitter system of claim 1 wherein said output means comprises: a phase sensitive detector for generating a DC signal representing the unbalance of said bridge network; and output amplifier means responsive to said DC signal and including a voltage feedback network responsive to the 4-20 milliamp DC current, said feedback network stabilizing the flow of said 4-20 milliamp DC current at all levels.
14. The two-wire transmitter system of claim 1 further comprising a full wave rectifying bridge at the input of said two-wire transmitter for con-nection to said pair of transmission lines so as to permit said two-wire trans-mitter to operate regardless of the polarity of the current applied to said two-wire transmitter.
15. The two-wire transmitter system of claim 1 wherein said sensing probe comprises a guard electrode surrounding said probe electrode, said guard electrode being connected to said bridge network so as to drive said guard electrode at substantially the same potential as said probe electrode at a given operating point.
16. The two-wire transmitter system of claim 15 further comprising a coaxial cable having an axially extending conductor surrounded by a coaxial shield, said shield connecting said guard electrode to one side of said bridge network and said axial conductor connecting said probe electrode to the other side of said bridge network.
17. The two-wire transmitter system of claim 16 wherein said bridge network further comprises a span capacitor connected from said one side of said bridge network to said other side of said bridge network, said output means being coupled to said bridge network to cross said span capacitance.
18. In a two-wire transmitter system comprising a power supply at a first location, a system for monitoring the condition of materials at a second location comprising: an admittance sensing probe including a probe electrode adapted to detect the admittance of materials, an admittance responsive network including the admittance detected by said probe output means coupled to said network for generating a signal upon a wire of the two-wire transmitter system in response to the condition of the materials.
19. The system of claim 18 including an amplifier means DC coupled to said network and DC isolated from said output means.
20. The system of claim 19 including rectifying means coupled between said amplifier means and said bridge network for providing a DC power supply voltage to said amplifier while maintaining isolation between said amplifier and said power supply.
21. The two-wire transmitter system of claim 18 wherein said DC isola-ting means includes a transformer having a primary coupled to said RF signal generator and a secondary forming part of said bridge network.
22. The two-wire transmitter system of claim 21 wherein said RF signal generator comprises an oscillator including a resonant circuit, said primary, said secondary and said detected admittance comprising part of said resonant circuit.
23. A system for monitoring the condition of materials in a two-wire system comprising a power supply and a load at one location and a two-wire transmitter at another location interconnected by a pair of transmission lines carrying a variable current, the improvement residing in said two-wire trans-mitter comprising an admittance sensing probe including a probe electrode adapted to detect the admittance of materials for monitoring the condition of the materials; an RF signal generator comprising an oscillator including a re-sonant circuit, said resonant circuit including a bridge network including the admittance detected by said probe such that the unbalance of the bridge net-work corresponds to the condition of the materials being monitored; and output means coupled to said bridge network for changing the current flow through the transmission lines in response to the unbalance of the bridge network so as to represent the condition of the materials.
24. A system for monitoring the condition of materials comprising: an admittance sensing probe including a probe electrode adapted to detect the ad-mittance of materials for monitoring the condition of the materials; a bridge network including the admittance detected by said probe when said unbalance of the bridge network corresponds to the condition of materials being monitored;
an RF signal generator coupled to said bridge network and comprising an RF
oscillator and a regulating circuit for said oscillator for maintaining the am-plitude of said RF signals substantially constant, said regulating circuit in-cluding means for full wave rectifying the output of said oscillator, a capa-citor coupled to said full wave rectifying means and charged by current flow through said rectifying means, a voltage divider connected between said capa-citor and a regulated source of voltage and control means coupled to said voltage divider and said oscillator so as to maintain the amplitude of said RF signal and the voltage across said capacitor substantially constant; and output means coupled to said bridge network for generating an output signal in response to the unbalance of said bridge network so as to represent the condition of the materials being monitored.
an RF signal generator coupled to said bridge network and comprising an RF
oscillator and a regulating circuit for said oscillator for maintaining the am-plitude of said RF signals substantially constant, said regulating circuit in-cluding means for full wave rectifying the output of said oscillator, a capa-citor coupled to said full wave rectifying means and charged by current flow through said rectifying means, a voltage divider connected between said capa-citor and a regulated source of voltage and control means coupled to said voltage divider and said oscillator so as to maintain the amplitude of said RF signal and the voltage across said capacitor substantially constant; and output means coupled to said bridge network for generating an output signal in response to the unbalance of said bridge network so as to represent the condition of the materials being monitored.
25. The system of claim 24 wherein said RF oscillator comprises a mul-tivibrator and a resonant circuit coupled to the output of said multivibrator.
26. In a two-wire transmitter system comprising a power supply and a load at one location and a two-wire transmitter at another location intercon-nected by a pair of transmission lines carrying a variable DC current, the im-provement residing in said two-wire transmitter comprising: a pair of input terminals connected to said pair of transmission lines respectively; an RF
signal generator; an admittance sensing probe including a probe electrode adap-ted to detect the admittance of materials for monitoring the condition of the materials; a bridge network coupled to said RP signal generator, said bridge network including the admittance detected by said probe such that the unbalan-ce of the network corresponds to the condition of the materials being monito-red; output means coupled to said bridge network for varying the DC current flow through the transmission lines; and a full wave rectifying bridge compri-sing four rectifying means coupling said input terminals to said output means, one pair of said rectifying means conducting when current flow through said transmission lines is of one polarity and the other pair of said rectifying means conducting when the current flow through said transmission lines is of the other polarity.
signal generator; an admittance sensing probe including a probe electrode adap-ted to detect the admittance of materials for monitoring the condition of the materials; a bridge network coupled to said RP signal generator, said bridge network including the admittance detected by said probe such that the unbalan-ce of the network corresponds to the condition of the materials being monito-red; output means coupled to said bridge network for varying the DC current flow through the transmission lines; and a full wave rectifying bridge compri-sing four rectifying means coupling said input terminals to said output means, one pair of said rectifying means conducting when current flow through said transmission lines is of one polarity and the other pair of said rectifying means conducting when the current flow through said transmission lines is of the other polarity.
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US05/507,540 US3993947A (en) | 1974-09-19 | 1974-09-19 | Admittance measuring system for monitoring the condition of materials |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1050114A true CA1050114A (en) | 1979-03-06 |
Family
ID=24019038
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA235,144A Expired CA1050114A (en) | 1974-09-19 | 1975-09-10 | Admittance measuring system for monitoring the condition of materials |
Country Status (5)
| Country | Link |
|---|---|
| US (2) | US3993947A (en) |
| JP (2) | JPS6346366B2 (en) |
| CA (1) | CA1050114A (en) |
| DE (1) | DE2541908A1 (en) |
| GB (1) | GB1528167A (en) |
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-
1974
- 1974-09-19 US US05/507,540 patent/US3993947A/en not_active Expired - Lifetime
-
1975
- 1975-09-10 CA CA235,144A patent/CA1050114A/en not_active Expired
- 1975-09-17 GB GB38248/75A patent/GB1528167A/en not_active Expired
- 1975-09-18 JP JP50113544A patent/JPS6346366B2/ja not_active Expired
- 1975-09-19 DE DE19752541908 patent/DE2541908A1/en active Granted
-
1976
- 1976-11-22 US US05/743,618 patent/US4146834A/en not_active Expired - Lifetime
-
1988
- 1988-04-27 JP JP63102808A patent/JPS63289423A/en active Granted
Also Published As
| Publication number | Publication date |
|---|---|
| US3993947A (en) | 1976-11-23 |
| DE2541908A1 (en) | 1976-04-08 |
| US3993947B1 (en) | 1992-07-14 |
| DE2541908C2 (en) | 1987-08-20 |
| JPS5182651A (en) | 1976-07-20 |
| GB1528167A (en) | 1978-10-11 |
| JPS63289423A (en) | 1988-11-25 |
| US4146834B1 (en) | 1992-09-22 |
| JPS6346366B2 (en) | 1988-09-14 |
| US4146834A (en) | 1979-03-27 |
| JPH0248844B2 (en) | 1990-10-26 |
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