CA1194138A - Terminal arrangement for a duplex transmission system - Google Patents
Terminal arrangement for a duplex transmission systemInfo
- Publication number
- CA1194138A CA1194138A CA000430509A CA430509A CA1194138A CA 1194138 A CA1194138 A CA 1194138A CA 000430509 A CA000430509 A CA 000430509A CA 430509 A CA430509 A CA 430509A CA 1194138 A CA1194138 A CA 1194138A
- Authority
- CA
- Canada
- Prior art keywords
- equalizer
- arrangement
- transmission
- alpha
- reception
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L5/00—Arrangements affording multiple use of the transmission path
- H04L5/14—Two-way operation using the same type of signal, i.e. duplex
- H04L5/1423—Two-way operation using the same type of signal, i.e. duplex for simultaneous baseband signals
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03006—Arrangements for removing intersymbol interference
- H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
- H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
- H04L25/03057—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception with a recursive structure
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L25/00—Baseband systems
- H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
- H04L25/40—Transmitting circuits; Receiving circuits
- H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
Abstract
ABSTRACT:
"Terminal arrangement for a duplex transmission system".
Terminal arrangement for a duplex transmission system for digital signals, comprising an equalizer for reception and an equalizer for transmission.
In order to realise a terminal arrangement which automatically adapts itself to the transmission path the terminal arrangement comprises an adaptive quantized feedback equalizer (ER) for reception and a pulse-shaping equalizer for transmission (ET) and a converter arrange-ment (OM) for converting the coefficients (C1, C2, ...) from the equalizer for reception into the correction factors ( ? 1, ...) for the equalizer for transmission.
Use : digital subscriber lines.
"Terminal arrangement for a duplex transmission system".
Terminal arrangement for a duplex transmission system for digital signals, comprising an equalizer for reception and an equalizer for transmission.
In order to realise a terminal arrangement which automatically adapts itself to the transmission path the terminal arrangement comprises an adaptive quantized feedback equalizer (ER) for reception and a pulse-shaping equalizer for transmission (ET) and a converter arrange-ment (OM) for converting the coefficients (C1, C2, ...) from the equalizer for reception into the correction factors ( ? 1, ...) for the equalizer for transmission.
Use : digital subscriber lines.
Description
i~94"1 3~3 PHN 10.377 l 25.5.1983 "Terminal arrangement for a duplex transmission system".
A. Background of -the invention.
A (1~ Field of the invention.
The invention relates to a terminal arrangement for a duplex transmission system for digital signals, comprising a transmitting arrangement and a receiving arrangement and a coupling arrangement for coupling the transmitting and receiving arrangements to a duplex transmission path, the receiving arrangement comprising an equalizer for reception and the transmitting arrangement comprising an equalizer for transmission and means being provided for adjusting the equalizer for transmission in response to the received digital signals in combination with the equalizer for reception.
The terminal arrangement is, for example, intended for a digital subscriber connection in a two-wire full-duplex system, which has for its object to replace and improve existing analog subscriber connections, the existing subscribers' lines being utilised.
The frequency band required for such a system may 20 be between approximately 0,1 - 200 kHz.
When equipment for the system is being installed, only a few details will be known of the subscribers~ lines, such as, for example: length, diameter, cross-ta]k signals, interferences and the number of stubs, and equalization will be required which can automatically adapt itself for an optimum reception.
An adapti~e quantized feedback equalizer (AQF-equalizer) satisfies this require~ent. This equalizer adapts itself continuously to a maximum eye opening of 30 the digital signal.
In principle such an eq-ualizer would be required at the sub.scriber set as well as in the exchange.
~94~38 PHN 10.377 2 25.5.1983 In order to keep the dissipation in the subscriber set as low as possible 9 and to prevent the subscriber set from being unnecessarily expensive, methods to have the pulse correction be effected in the exchange only are sought for.
A (2) Description of the prior art.
Generally, cable losses are corrected by means of an equalizing network or a pulse-shaping method.
Equalization may, in principle, be effected at the begin-ning or at the end of a transmission path, while pulse-shaping is usually effected afterwards 7 that is to say at the receiving side.
In its most simple form an equalizing ne-twork is an RC-circuit with which the attenuation variation of the cable is indeed equalized, but an unwanted phase shift is introduced at the same time. As a result thereof, the correction of digital signals will not be optimal.
The cross-talk signals from adjacent wire pairs in a cable are the stronger according as the frequency of the noise signals becomes higher. Fqualization by means of a network now has the effect that at high frequencies these noise signals will be still more amplified, relatively so that the signal-to-noise ratio is considerably affected.
A g:reat disadvantage of this equalization is therefore the decrease in the signal-to-noise ratio, as a result of which the maximum useful cable length decreases.
For digital signals the quality of the trans-mission is determined by the number of bit errors produced during regeneration. This number of errors is determined by the extent to which the so-called eye is still sufficiently open.
In order to ensure that the intersymbol inter-ference is as low as possible, it is a requirement that at the sampling moments which are located at n-times the 35 bit period (t = n r ), the signal obtained in response to the preceding data pulses must be as small as possible.
The equalizer for transmission provides that 119~38 PHN 10.377 3 25.5.1983 intersymbol interference at the receiving end is as small as possible. This method is used at the transmitter instead of at the receiver.
A terminal arrangement of the type described in A(l) is disclosed in Japanese Patent Application~
publication number 9083/77 in which the equalization for the return direction is derived from the receiving arrangement, it being assumed that the return line is equal to the receiving line.
B,Summary of the invention.
The invention has for its object to provide a simple construction of the terminal arrangement of the type set forth in the opening paragraph.
According to the invention, the terminal arrangemen-t is characterized in that the equalizer for reception is provided by an adaptive quantize~ feedback equalizer having adaptively adjustable coefficients ~Cl~ C2, C3, ...) and that the equalizer for transmission is provided by a pulse-shaping equalizer which generates a number (n) of variably dela~ed versions of the digital input signal and after multiplication of these versions by an associated correction factor (CG n) adds them to the digital input signal for forming a predistorted digital transmit signal and that a converter arrangement is pro-vided for converting the adaptively adjustable coefficientsof the said feedback equalizer into the correction factors for the pulse-shaping equalizer in accordance with the recursive expressions:
C1 + OC 1 = O
C2 ~ ~ 1C1 + ~ 2 =
.......
Cn + ~ 1Cn-1 + ~ 2 Cn-2 + ... + ~ n = 0 The correction method for the transmit signal utilizes the coefficients found in the transversal filter of the AQF equaliser. The coefficients found correspond to the values of the samples of the implllse response of the transmission path.
1194~3~
PHN 10.377 ~ 25.5.1983 As generall~- a cable behaves as a RC-network whose impulse response has an e t/RC character, a first correction of the transmit signal by ~ 1 = -C1 is in that case sufficient to realise an impulse response having a small residual value at the subsequent sampling moments.
When this method is used, the noise signals and the cross-talk signals are not amplified, so that the signal-to-noise ratio is not detoriated.
C. Short description of the Figures.
Figure 1 shows the block diagram of an embodiment of the terminal arrangement according to the invention, Figure ~ illustrates the impulse response of a cable;
Figs, 3a-b show a Table and a set of equations deriving the correction factors for the equalizer for transmission;
Figs. 4 and 5 show corrected impulse responses for different bit rates;
Fig. 6 shows the block diagram of a second embodiment of the terminal arrangement in accordance with the invention for bi-phase modulation; and Fig. 7 illustrates an example of the corrected bi-phase wave form.
D, Reference.
Equalisation of Data Transmission Channels bv Means of Adaptive Quantized Feedback, Werner ~drexser, AEU, V~0 34 (1980), No. 7/8 pp. 287-292.
E. Description of the embodiments.
The terminal arrangement shown -n Figo 1 comprises a transmitting arrangement ~, a receiving arran-gement 0 and a coupling arrangement K for coupling the transmission and receiving arrangements to a duple~
transmission path P.
Depending on the transmission mode used, the coupling arrangement K can be implemented in different manners. When the burst method is used the coupling 3~3 PHN 10.377 5 25,5.1983 arrangement K is in the form of a change~over switch which alternately connects the transmi-tting arrangement and the receiving arrangement to the transmission path. When simultaneous transmission in both signal directions is used, the coupling arrangement is in the form of a hybrid in combination with an echo canceller. These components are known and need no further description.
The receiving arrangement 0 comprises a recei-ving filter F and an equalizer for reception ER, which has an output U for the regenerated digital signal~
The transmitting arrangement has an input I
for a digital signal, an encoder arrangement C and an equalizer for transmission ET. Depending on the transmis-sion code used, the encoAing arrangement C may be implemen-ted in different ways~ When the NRZ-signal form is used, considered over the symbol interval, the encoding arrangement C produces a signal which does not change during the symbol period (T), In the event of bi-phase encoding the encoding arrangement C produces the bi-phase waveform 20 in each symbol interval. This waveform has a transition halfway along the symbol interval: the solid line in Fig. 7 illustrates this waveform.
The equalizer for reception ER is provided by an adaptive quantised feedback equalizer of a type which 25 is known ~ se~ for example from the sub. D. reference.
In a difference producer 1 a compensation signal issubtracted from the output signal of filter F. The output signal from difference producer 1 is applied to a regenerator 2, which produces a regenerated digital signal 30 at the output U.
The regenerated digital signal is applied to a cascade arrangement of delay se ~ions ( ~~ ). After having been multiplied by the adaptive, adjustable coefficients C1, C2, C3 and C4 in the correspondingly referenced multipliers, 35 the output signals of these sections are added together in an adder circuit 3 for forming the compensation signal.
A difference producer 4 derives from the input 119~ 13~
PHN 1O.377 6 25.5.1983 and the output of the regenerator 2 an error signal which is applied to a set of multipliers (x) for multiplication by the output signals of the delay sections ( ~ ). The coefficients C1, ... C4 are derived from the output signals of the amplifiers (x).
The coefficients C1, .., C4 automatically adjust themselves such that the impulse response of the trans-mission path P which is sampled by regenerator 2 at the instant t = O is compensated for at the i~tants t = n r .
Eig, 2 illustrates the impulse response of a subscriber line to a NRZ pulse. I~erein Co is the signal value at the sampling instant t = O and C1, ... C4 are the signal values at the instant t = n r . These values correspond to the coefficient values C1~ ... Cl~ of the equalizer for reception ER, (it being assumed that Co = 1~
The equalizer for transmission ET is formed by a pulse-shaping equalizer of a known type.
The output signal of encoder arrangement C
is applied to a cascade arrangement of delay section ( r ).
20 After having been multiplied by the correction factors ~ 1, ~ 2 and cG3 in a set of multipliers (x) the output signals of these sections are added together and added by adder arrangement 5 to the output signal of encoder arrangement C for forming a predistorted digital transmit signal. This predistorted digital output signal is applied to the coupling arrangement K for transmission over the transmission path P.
The terminal arrangement shown in fig. 1 compri-ses a converter arrangement OM, which converts the 30 coefficients C1, .,. C3 from the equalizer for l~eception ER into the correction factors ~ 1, ... C~ 3 for the equalizer for transmission ET, The equalizer for transmission ET has for its object to provide by predistortion of the transmitted signals a response at the receiving end which approximates to zero at the instants tn = n r (n ~ O).
The Table of Fig, 3a shows for the instants 1~94~38 PIIN 1O.377 7 25.5.19~3 tn = n ~ the response of the transmission path to a predistorted output pulse from equalizer ET. ~ig. 3b shows the recurrent expressions which the correction factors ~ n must satisfy in order to make the response of the transmission path to the predistorted transmitting pulse equal to zero at the instant tn = n r (n ~ O).
The signal values C1, ... Cn of the impulse response of the transmission path correspond to the adaptively adjusted coefficients C1, ,,, Cn of the equalizer for reception ER. By converting by means of the converting arrangement OM the coefficients C1, ... Cn of the equalizer ER into the correction factorDc n in accordance with the expressions of Fig,3b, an equalizer for transmission operating to the desired goal will then lS be realised.
In practice it has been found possible to suffice with only the correction factor ~ 1 in the equalizer for transmission ET, that is to say to use only one correction pulse. For the case in which a subscriber cable having a length of 2 km and a diameter of O 5O mm is used, the impulse response is illustrated in Fig. 4 by curve A1 and in Fig. 5by curve A2. Fig, 4 relates to a bit rate of 333 kB/s ( ~~ =O.3/us) on the transmission path and Fig. 5 to a bit rate of 1 MB/s ( ~ = 0 1/us). For these cases the impulse response to the correction pul e is illustrated by means of the curves Bl and B2 and the ultimate response of the subscriber cable tothe predis-torted transmitting pulse is illustrated by means of the curves D1 and D2. From this it appears that the o-verall response at the instants t = n ~ (n ~ O) approximates already to a very satisfa tory extent to the value zero.
If necessary a second correction pulse can be used at the distance 2 ~ (correction factor ~ 2) in order to reduce still further the cable response at the instant t = 2~ and the subsequent sampling instants.
Fig 6 shows an embodiment of -the terminal arrangement in accordance ~ith the invention for bi-phase 1~9~
PHN 10.377 ~ 25.5.1983 modulation. Components corresponding Irith those shown in Fig. 1 are referenced correspondingly.
The spectrum of bi-phase modulation extends from O Hz to twice the bit frequency (2/T), wherein T
represents the svmbol period in sec. In accordance with the sampling theorem, such a signal must be sampled with a frequency fs = l/ r = 4/T ~ so that here it must hold that r = T/4.
l`he equalizer for transrnission ET comprises only one delay section ~ = T/4 and an analog multiplier (x). The converter arrang~ment OM is here provided by a simple inverter, which realizes the relation ~ 1 = - C1.
A D/A converter converts the value of ~ 1 into an analog signal and applies it to the analog multiplier (x) in the equalizer for transmission ET.
In Fig. 7 the solid line ( - ~ illustrates the bi-phase signal waveform at the output of encoding arrangement C. The broken line (------) illustrates the correction pulse (CG 1 = -2 ) which is shifted a period of time r = T/4 and the dot-and-dash line (-.-.-) illu-strates the predistorted output pulse of equalizer ET.
It will be seen that the above-described prin-ciples can be used irrespective of whether binary signal pulses or signal pulses having more than two values are transmitted. For the latter case the system components must however be arranged for processing multilevel pulses and, for example, the regenerator 2 shown in Fig. 1 must be capable of regenerating multi-level signals.
It will further be seen that the described principles can be used independently of the code produced by encoder arrangement C. Account must only be taken of the bit rate at the output of encoder arrangement C, which when, for example, bi-phase modulation is used is doubled with respect to the original bit rate. ~Ioreover, the line code at the transmitting end need not be the same as the line code at the receiving end,
A. Background of -the invention.
A (1~ Field of the invention.
The invention relates to a terminal arrangement for a duplex transmission system for digital signals, comprising a transmitting arrangement and a receiving arrangement and a coupling arrangement for coupling the transmitting and receiving arrangements to a duplex transmission path, the receiving arrangement comprising an equalizer for reception and the transmitting arrangement comprising an equalizer for transmission and means being provided for adjusting the equalizer for transmission in response to the received digital signals in combination with the equalizer for reception.
The terminal arrangement is, for example, intended for a digital subscriber connection in a two-wire full-duplex system, which has for its object to replace and improve existing analog subscriber connections, the existing subscribers' lines being utilised.
The frequency band required for such a system may 20 be between approximately 0,1 - 200 kHz.
When equipment for the system is being installed, only a few details will be known of the subscribers~ lines, such as, for example: length, diameter, cross-ta]k signals, interferences and the number of stubs, and equalization will be required which can automatically adapt itself for an optimum reception.
An adapti~e quantized feedback equalizer (AQF-equalizer) satisfies this require~ent. This equalizer adapts itself continuously to a maximum eye opening of 30 the digital signal.
In principle such an eq-ualizer would be required at the sub.scriber set as well as in the exchange.
~94~38 PHN 10.377 2 25.5.1983 In order to keep the dissipation in the subscriber set as low as possible 9 and to prevent the subscriber set from being unnecessarily expensive, methods to have the pulse correction be effected in the exchange only are sought for.
A (2) Description of the prior art.
Generally, cable losses are corrected by means of an equalizing network or a pulse-shaping method.
Equalization may, in principle, be effected at the begin-ning or at the end of a transmission path, while pulse-shaping is usually effected afterwards 7 that is to say at the receiving side.
In its most simple form an equalizing ne-twork is an RC-circuit with which the attenuation variation of the cable is indeed equalized, but an unwanted phase shift is introduced at the same time. As a result thereof, the correction of digital signals will not be optimal.
The cross-talk signals from adjacent wire pairs in a cable are the stronger according as the frequency of the noise signals becomes higher. Fqualization by means of a network now has the effect that at high frequencies these noise signals will be still more amplified, relatively so that the signal-to-noise ratio is considerably affected.
A g:reat disadvantage of this equalization is therefore the decrease in the signal-to-noise ratio, as a result of which the maximum useful cable length decreases.
For digital signals the quality of the trans-mission is determined by the number of bit errors produced during regeneration. This number of errors is determined by the extent to which the so-called eye is still sufficiently open.
In order to ensure that the intersymbol inter-ference is as low as possible, it is a requirement that at the sampling moments which are located at n-times the 35 bit period (t = n r ), the signal obtained in response to the preceding data pulses must be as small as possible.
The equalizer for transmission provides that 119~38 PHN 10.377 3 25.5.1983 intersymbol interference at the receiving end is as small as possible. This method is used at the transmitter instead of at the receiver.
A terminal arrangement of the type described in A(l) is disclosed in Japanese Patent Application~
publication number 9083/77 in which the equalization for the return direction is derived from the receiving arrangement, it being assumed that the return line is equal to the receiving line.
B,Summary of the invention.
The invention has for its object to provide a simple construction of the terminal arrangement of the type set forth in the opening paragraph.
According to the invention, the terminal arrangemen-t is characterized in that the equalizer for reception is provided by an adaptive quantize~ feedback equalizer having adaptively adjustable coefficients ~Cl~ C2, C3, ...) and that the equalizer for transmission is provided by a pulse-shaping equalizer which generates a number (n) of variably dela~ed versions of the digital input signal and after multiplication of these versions by an associated correction factor (CG n) adds them to the digital input signal for forming a predistorted digital transmit signal and that a converter arrangement is pro-vided for converting the adaptively adjustable coefficientsof the said feedback equalizer into the correction factors for the pulse-shaping equalizer in accordance with the recursive expressions:
C1 + OC 1 = O
C2 ~ ~ 1C1 + ~ 2 =
.......
Cn + ~ 1Cn-1 + ~ 2 Cn-2 + ... + ~ n = 0 The correction method for the transmit signal utilizes the coefficients found in the transversal filter of the AQF equaliser. The coefficients found correspond to the values of the samples of the implllse response of the transmission path.
1194~3~
PHN 10.377 ~ 25.5.1983 As generall~- a cable behaves as a RC-network whose impulse response has an e t/RC character, a first correction of the transmit signal by ~ 1 = -C1 is in that case sufficient to realise an impulse response having a small residual value at the subsequent sampling moments.
When this method is used, the noise signals and the cross-talk signals are not amplified, so that the signal-to-noise ratio is not detoriated.
C. Short description of the Figures.
Figure 1 shows the block diagram of an embodiment of the terminal arrangement according to the invention, Figure ~ illustrates the impulse response of a cable;
Figs, 3a-b show a Table and a set of equations deriving the correction factors for the equalizer for transmission;
Figs. 4 and 5 show corrected impulse responses for different bit rates;
Fig. 6 shows the block diagram of a second embodiment of the terminal arrangement in accordance with the invention for bi-phase modulation; and Fig. 7 illustrates an example of the corrected bi-phase wave form.
D, Reference.
Equalisation of Data Transmission Channels bv Means of Adaptive Quantized Feedback, Werner ~drexser, AEU, V~0 34 (1980), No. 7/8 pp. 287-292.
E. Description of the embodiments.
The terminal arrangement shown -n Figo 1 comprises a transmitting arrangement ~, a receiving arran-gement 0 and a coupling arrangement K for coupling the transmission and receiving arrangements to a duple~
transmission path P.
Depending on the transmission mode used, the coupling arrangement K can be implemented in different manners. When the burst method is used the coupling 3~3 PHN 10.377 5 25,5.1983 arrangement K is in the form of a change~over switch which alternately connects the transmi-tting arrangement and the receiving arrangement to the transmission path. When simultaneous transmission in both signal directions is used, the coupling arrangement is in the form of a hybrid in combination with an echo canceller. These components are known and need no further description.
The receiving arrangement 0 comprises a recei-ving filter F and an equalizer for reception ER, which has an output U for the regenerated digital signal~
The transmitting arrangement has an input I
for a digital signal, an encoder arrangement C and an equalizer for transmission ET. Depending on the transmis-sion code used, the encoAing arrangement C may be implemen-ted in different ways~ When the NRZ-signal form is used, considered over the symbol interval, the encoding arrangement C produces a signal which does not change during the symbol period (T), In the event of bi-phase encoding the encoding arrangement C produces the bi-phase waveform 20 in each symbol interval. This waveform has a transition halfway along the symbol interval: the solid line in Fig. 7 illustrates this waveform.
The equalizer for reception ER is provided by an adaptive quantised feedback equalizer of a type which 25 is known ~ se~ for example from the sub. D. reference.
In a difference producer 1 a compensation signal issubtracted from the output signal of filter F. The output signal from difference producer 1 is applied to a regenerator 2, which produces a regenerated digital signal 30 at the output U.
The regenerated digital signal is applied to a cascade arrangement of delay se ~ions ( ~~ ). After having been multiplied by the adaptive, adjustable coefficients C1, C2, C3 and C4 in the correspondingly referenced multipliers, 35 the output signals of these sections are added together in an adder circuit 3 for forming the compensation signal.
A difference producer 4 derives from the input 119~ 13~
PHN 1O.377 6 25.5.1983 and the output of the regenerator 2 an error signal which is applied to a set of multipliers (x) for multiplication by the output signals of the delay sections ( ~ ). The coefficients C1, ... C4 are derived from the output signals of the amplifiers (x).
The coefficients C1, .., C4 automatically adjust themselves such that the impulse response of the trans-mission path P which is sampled by regenerator 2 at the instant t = O is compensated for at the i~tants t = n r .
Eig, 2 illustrates the impulse response of a subscriber line to a NRZ pulse. I~erein Co is the signal value at the sampling instant t = O and C1, ... C4 are the signal values at the instant t = n r . These values correspond to the coefficient values C1~ ... Cl~ of the equalizer for reception ER, (it being assumed that Co = 1~
The equalizer for transmission ET is formed by a pulse-shaping equalizer of a known type.
The output signal of encoder arrangement C
is applied to a cascade arrangement of delay section ( r ).
20 After having been multiplied by the correction factors ~ 1, ~ 2 and cG3 in a set of multipliers (x) the output signals of these sections are added together and added by adder arrangement 5 to the output signal of encoder arrangement C for forming a predistorted digital transmit signal. This predistorted digital output signal is applied to the coupling arrangement K for transmission over the transmission path P.
The terminal arrangement shown in fig. 1 compri-ses a converter arrangement OM, which converts the 30 coefficients C1, .,. C3 from the equalizer for l~eception ER into the correction factors ~ 1, ... C~ 3 for the equalizer for transmission ET, The equalizer for transmission ET has for its object to provide by predistortion of the transmitted signals a response at the receiving end which approximates to zero at the instants tn = n r (n ~ O).
The Table of Fig, 3a shows for the instants 1~94~38 PIIN 1O.377 7 25.5.19~3 tn = n ~ the response of the transmission path to a predistorted output pulse from equalizer ET. ~ig. 3b shows the recurrent expressions which the correction factors ~ n must satisfy in order to make the response of the transmission path to the predistorted transmitting pulse equal to zero at the instant tn = n r (n ~ O).
The signal values C1, ... Cn of the impulse response of the transmission path correspond to the adaptively adjusted coefficients C1, ,,, Cn of the equalizer for reception ER. By converting by means of the converting arrangement OM the coefficients C1, ... Cn of the equalizer ER into the correction factorDc n in accordance with the expressions of Fig,3b, an equalizer for transmission operating to the desired goal will then lS be realised.
In practice it has been found possible to suffice with only the correction factor ~ 1 in the equalizer for transmission ET, that is to say to use only one correction pulse. For the case in which a subscriber cable having a length of 2 km and a diameter of O 5O mm is used, the impulse response is illustrated in Fig. 4 by curve A1 and in Fig. 5by curve A2. Fig, 4 relates to a bit rate of 333 kB/s ( ~~ =O.3/us) on the transmission path and Fig. 5 to a bit rate of 1 MB/s ( ~ = 0 1/us). For these cases the impulse response to the correction pul e is illustrated by means of the curves Bl and B2 and the ultimate response of the subscriber cable tothe predis-torted transmitting pulse is illustrated by means of the curves D1 and D2. From this it appears that the o-verall response at the instants t = n ~ (n ~ O) approximates already to a very satisfa tory extent to the value zero.
If necessary a second correction pulse can be used at the distance 2 ~ (correction factor ~ 2) in order to reduce still further the cable response at the instant t = 2~ and the subsequent sampling instants.
Fig 6 shows an embodiment of -the terminal arrangement in accordance ~ith the invention for bi-phase 1~9~
PHN 10.377 ~ 25.5.1983 modulation. Components corresponding Irith those shown in Fig. 1 are referenced correspondingly.
The spectrum of bi-phase modulation extends from O Hz to twice the bit frequency (2/T), wherein T
represents the svmbol period in sec. In accordance with the sampling theorem, such a signal must be sampled with a frequency fs = l/ r = 4/T ~ so that here it must hold that r = T/4.
l`he equalizer for transrnission ET comprises only one delay section ~ = T/4 and an analog multiplier (x). The converter arrang~ment OM is here provided by a simple inverter, which realizes the relation ~ 1 = - C1.
A D/A converter converts the value of ~ 1 into an analog signal and applies it to the analog multiplier (x) in the equalizer for transmission ET.
In Fig. 7 the solid line ( - ~ illustrates the bi-phase signal waveform at the output of encoding arrangement C. The broken line (------) illustrates the correction pulse (CG 1 = -2 ) which is shifted a period of time r = T/4 and the dot-and-dash line (-.-.-) illu-strates the predistorted output pulse of equalizer ET.
It will be seen that the above-described prin-ciples can be used irrespective of whether binary signal pulses or signal pulses having more than two values are transmitted. For the latter case the system components must however be arranged for processing multilevel pulses and, for example, the regenerator 2 shown in Fig. 1 must be capable of regenerating multi-level signals.
It will further be seen that the described principles can be used independently of the code produced by encoder arrangement C. Account must only be taken of the bit rate at the output of encoder arrangement C, which when, for example, bi-phase modulation is used is doubled with respect to the original bit rate. ~Ioreover, the line code at the transmitting end need not be the same as the line code at the receiving end,
Claims
1. A terminal arrangement for a duplex transmission system for digital signals, comprising a transmitting arrangement and a receiving arrangement and a coupling arrangement for coupling the transmitting and receiving arrangements to a duplex transmission path, the receiving arrangement comprising an equalizer for reception and the transmitting arrangement comprising an equalizer for transmission and means being provided for adjusting the equalizer for transmission in response to the received digital signals in combination with the equalizer for reception, characterized in that the equali-zer for reception is provided by an adaptive quantized feedback equalizer having adaptively adjustable coeffi-cients (C1, C2, C3, ,.,)and that the equalizer for transmission is provided by a pulse-shaping equalizer which generates a number (n) of variably delayed versions of the digital input and after multiplication of these versions by an associated correction factor ( .alpha. n) adds them to the digital input signal for forming a predistor-ted digital transmit signal and that a converter arrange-ment is provided for converting the adaptively adjustable coefficients of the said feedback equalizer into the correction factors for the pulse-shaping equalizer in accordance with the recursive expressions:
C1 + .alpha.1 = O
C2 + .alpha.1C1 + .alpha. 2 = O
........
Cn + .alpha.1Cn-1 + .alpha. 2 Cn-2 + ,.,. + .alpha. n = 0.
C1 + .alpha.1 = O
C2 + .alpha.1C1 + .alpha. 2 = O
........
Cn + .alpha.1Cn-1 + .alpha. 2 Cn-2 + ,.,. + .alpha. n = 0.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| NL8202438 | 1982-06-16 | ||
| NL8202438A NL8202438A (en) | 1982-06-16 | 1982-06-16 | END DEVICE FOR A DUPLEX TRANSMISSION SYSTEM. |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1194138A true CA1194138A (en) | 1985-09-24 |
Family
ID=19839893
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000430509A Expired CA1194138A (en) | 1982-06-16 | 1983-06-16 | Terminal arrangement for a duplex transmission system |
Country Status (6)
| Country | Link |
|---|---|
| US (1) | US4535443A (en) |
| EP (1) | EP0096943B1 (en) |
| JP (1) | JPS594336A (en) |
| CA (1) | CA1194138A (en) |
| DE (1) | DE3369450D1 (en) |
| NL (1) | NL8202438A (en) |
Families Citing this family (25)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FR2571566B1 (en) * | 1984-10-09 | 1987-01-23 | Labo Electronique Physique | DIGITAL DATA RECEIVING DEVICE COMPRISING AN ADAPTIVE RHYTHM RECOVERY DEVICE |
| AR241298A1 (en) * | 1985-10-03 | 1992-04-30 | Siemens Ag | Adaptive transversal equalizer |
| US4745622A (en) * | 1986-07-29 | 1988-05-17 | Integrated Network Corporation | Equalizer for digital transmission systems |
| US4797898A (en) * | 1986-11-21 | 1989-01-10 | Racal Data Communications Inc. | Method and apparatus for equalization of data transmission system |
| CA2012914C (en) * | 1989-05-12 | 1999-05-04 | Vedat M. Eyuboglu | Trellis precoding for modulation systems |
| EP0411463A1 (en) * | 1989-08-03 | 1991-02-06 | Siemens Aktiengesellschaft | Digital transmission system |
| US5299230A (en) * | 1990-12-21 | 1994-03-29 | The Whitaker Corporation | Digital data transmission system with predistortion of transmitted pulses |
| US5394440A (en) * | 1991-02-06 | 1995-02-28 | General Datacomm, Inc. | High speed modem systems incorporating distribution preserving Tomlinson encoding and decoding for secondary channels |
| US5291520A (en) * | 1991-02-06 | 1994-03-01 | General Datacomm, Inc. | Methods and apparatus employing distribution preserving Tomlinson precoding in transmission of digital data signals |
| US5260971A (en) * | 1991-02-06 | 1993-11-09 | General Datacomm, Inc. | Apparatus and methods employing distribution preserving tomlinson precoding in transmission of digital data signals |
| US5253272A (en) * | 1991-03-01 | 1993-10-12 | Amp Incorporated | Digital data transmission system with adaptive predistortion of transmitted pulses |
| FI92357C (en) * | 1992-10-12 | 1994-10-25 | Nokia Mobile Phones Ltd | Channel equalizer for a bidirectional telecommunication system |
| US5440594A (en) * | 1993-12-09 | 1995-08-08 | Bell Communications Research, Inc. | Method and apparatus for joint optimization of transmitted pulse shape and receiver timing in digital systems |
| DE4342559A1 (en) * | 1993-12-14 | 1995-06-22 | Kommunikations Elektronik | Digital data transmission method |
| KR960011740B1 (en) * | 1994-01-18 | 1996-08-30 | 대우전자 주식회사 | Digital television system with updated equalizing function |
| EP0688116A1 (en) * | 1994-06-15 | 1995-12-20 | Laboratoires D'electronique Philips | Equatisation method for a bidirectional transmission system |
| US6470405B2 (en) * | 1995-10-19 | 2002-10-22 | Rambus Inc. | Protocol for communication with dynamic memory |
| US5724344A (en) * | 1996-04-02 | 1998-03-03 | Beck; William Federick | Amplifier using a single forward pilot signal to control forward and return automatic slope circuits therein |
| FR2748137B1 (en) * | 1996-04-24 | 1998-07-03 | Lewiner Jacques | PROCESS FOR OPTIMIZING RADIO COMMUNICATION BETWEEN A FIXED BASE AND A MOBILE |
| CA2206661C (en) * | 1997-05-29 | 2004-07-20 | Telecommunications Research Laboratories | A duplex decision feedback equalization system |
| US6266379B1 (en) | 1997-06-20 | 2001-07-24 | Massachusetts Institute Of Technology | Digital transmitter with equalization |
| US6741643B1 (en) | 1998-04-15 | 2004-05-25 | Telecommunications Research Laboratories | Asymmetric equalization system for data transmission |
| US7386053B2 (en) * | 2002-10-11 | 2008-06-10 | Synopsys, Inc | System and method of equalization of high speed signals |
| JP2006060451A (en) * | 2004-08-19 | 2006-03-02 | Matsushita Electric Ind Co Ltd | Power amplifier and delay measuring method for power combining system |
| US9755669B2 (en) * | 2015-09-01 | 2017-09-05 | Mediatek Inc. | Variation calibration for envelope tracking on chip |
Family Cites Families (10)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US3003030A (en) * | 1954-09-18 | 1961-10-03 | Kokusai Denshin Denwa Co Ltd | Transmission characteristic compensation system |
| US3356955A (en) * | 1964-05-22 | 1967-12-05 | Ibm | Digital automatic time domain equalizer |
| US3479458A (en) * | 1967-03-06 | 1969-11-18 | Honeywell Inc | Automatic channel equalization apparatus |
| BE758978A (en) * | 1969-11-20 | 1971-04-30 | Western Electric Co | TRANSMISSION SYSTEM FOR MULTI-LEVEL DIGITAL SIGNALS |
| JPS529083A (en) * | 1975-07-14 | 1977-01-24 | Tokyo Porimaa Kk | Method of receiving net of rubber composition |
| US4238779A (en) * | 1976-04-13 | 1980-12-09 | Intech Laboratories, Inc. | Data transmission and reception system |
| JPS5558612A (en) * | 1978-10-26 | 1980-05-01 | Kokusai Denshin Denwa Co Ltd <Kdd> | Delay circuit |
| FR2460075B1 (en) * | 1979-06-22 | 1988-12-09 | Cit Alcatel | ADAPTIVE ECHO CANCELLER FOR DUPLEX DATA TRANSMISSION |
| FR2491701A1 (en) * | 1980-10-08 | 1982-04-09 | Pays Gerard | METHOD AND DEVICE FOR MINIMIZING TELEDIAPHONIE BETWEEN DIGITAL TRANSMISSION LINES |
| US4377858A (en) * | 1980-10-23 | 1983-03-22 | International Telephone And Telegraph Corporation | Digital two-to-four wire converter for full duplex signals |
-
1982
- 1982-06-16 NL NL8202438A patent/NL8202438A/en not_active Application Discontinuation
-
1983
- 1983-06-14 JP JP58106587A patent/JPS594336A/en active Pending
- 1983-06-15 EP EP83200871A patent/EP0096943B1/en not_active Expired
- 1983-06-15 US US06/504,101 patent/US4535443A/en not_active Expired - Fee Related
- 1983-06-15 DE DE8383200871T patent/DE3369450D1/en not_active Expired
- 1983-06-16 CA CA000430509A patent/CA1194138A/en not_active Expired
Also Published As
| Publication number | Publication date |
|---|---|
| DE3369450D1 (en) | 1987-02-26 |
| NL8202438A (en) | 1984-01-16 |
| EP0096943B1 (en) | 1987-01-21 |
| US4535443A (en) | 1985-08-13 |
| EP0096943A2 (en) | 1983-12-28 |
| JPS594336A (en) | 1984-01-11 |
| EP0096943A3 (en) | 1985-01-30 |
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Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| MKEC | Expiry (correction) | ||
| MKEX | Expiry |