CA1295672C

CA1295672C - Quasi-resonant current mode power conversion method and apparatus

Info

Publication number: CA1295672C
Application number: CA000579608A
Authority: CA
Inventors: Deepakraj M. Divan
Original assignee: Wisconsin Alumni Research Foundation
Current assignee: Wisconsin Alumni Research Foundation
Priority date: 1987-10-16
Filing date: 1988-10-07
Publication date: 1992-02-11
Anticipated expiration: 2009-02-11
Also published as: WO1989004084A1; JPH02501974A; JP2636918B2; EP0344243A4; ATE90815T1; EP0344243B1; US4833584A; DE3881872D1; EP0344243A1; DE3881872T2

Abstract

QUASI RESONANT CURRENT MODE
POWER CONVERSION METHOD AND APPARATUS
ABSTRACT OF THE DISCLOSURE
An inverter has a resonant circuit composed of a parallel connected inductor and capacitor and a filter capacitor connected in series with the inductor which has a capacitance substantially greater than the resonant capacitor. A half bridge or full bridge switching circuit formed of pairs of gate controlled switching devices is connected to a DC power supply and to the resonant circuit and filter capacitor, with the switching devices being switched to provide a relatively high frequency, e.g., 20 KHz or higher, resonant current in the resonant circuit.
The filter capacitor is of a size such that the high frequency component of the current flowing in the resonant circuit does not result in a substantial voltage at the switching frequency appearing across the filter capacitor. In addition to the high frequency switching current in the resonant circuit, the switching frequency and the duration of switching is adjusted in a controlled manner such that a lower frequency AC component appears in the current flowing in the resonant circuit and through the filter capacitor such that a voltage at the lower frequency component appears across the filter capacitor.

Description

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QUASI-RESONANT CURRENT MoDE
STATIC POWER CONVEXSION METHOD AND APPARATUS
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This invention pertains generally to the field of static power converters and systems for the control of static power converters.

BACKGROUND OF THE INVENTION

he d~evelopment and commercial availability of gate ;turn-off devices capable o handling relatively large ;power~levels has~res~lted in a significant :change in power conver~sion technology. For example,~thyristors are now rarely u e~ in force-commutated systemsO To a large extent, the thyri~stor current source inverter has been replaced by GTO and transistor voltage source inverters at power ~ratings up to~l me~awatt (MW). The,voltage source inverter is particularly attractive because of its 15~ extremely simple power structure and the need for only six uni-directional gate turn-off devices (for three-phase ; load power). The antl-parallel diodes required across :: ::

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35~72 each of the gate turn-off devices are typically provided by the manufacturer in the same device packa~e for minimum lead inductance and ease of assembly. The con-trol strategy for such voltage source inverters is reasonably S simple and provides a fully regenerative interface between the DC source and the AC load.
Despite the clear advantages of the voltage source inverter structure, the inherent characteristics of available gate turn-off devices imposes several limitations on the performance of the inverters. For example, the high switching losses encountered in such inverters mandates the u~e of low switching frequencies, resulting in low amplifier bandwidth and poor load current waveform fidelity (unwanted harmonics). The rapid change of voltage with time on the output of the inverter generates interference due to capacitive coupling. The parallel diode reverse recovery and snubber interactions cause high device stresses under regeneration conditions.
In turn, the need to withstand the high device stresses reduces reliability and requires that the devices be overspecified. The relatively low switching frequencies required has also been observed to cause an acoustic noise problem because the switching frequency harmonics in the output power!generate noise at audible frequencies in the switching system and motor. And, in general, present nverter designs have poor regeneration capability into the AC supply Line, poor AC input line harmonics, requiring large DC link and AC side filters, and have poor fault recovery characteristics.
Ideally, a power converter should have essentially zero switching losses, a switching frequency greater than about 18 kHz ~above the audible range), small reactive components and the ability to transfer power ; ~ bi-directionally. The system should also be insensitive to second order parameters such as diode recovery times, device turn-off characteristics and parasitic reactiv~
~ elements. It is clear that present voltage source ::

~g~672 inverter designs do not achieve such optimum converter characteristics.
It is apparent that a substantial increase in inverter switching frequency would be desirable to minimize the lower order harmonics in pulse width modulated inverters.
Higher switching ~requencies have the aecompanying advantages of higher current regulator bandwidth, smaller reactive component size and, or frequencies above 18 kHz, acoustic noise which is not perceptible to humans.
Increases in pulse width modulated inverter switching frequencies achieved in the las-t several years (from about SOO Hz to 2 k~z for supplies rated from l to 25 XW) have generally been accomplished because of improvements in the speed and ratings of the newer devices. An alternative approach is to modify the switching circuit structure to make best use of the characteristics of available devices.
One well-established approacil is the use of snubber networks whicll protect the devices by diverting switching losses away ~rom the device itself. l'he most popular snubber configuration is a simple circuit strueture in which a small inductor provides turn-on protection while a shunt diode and capacitor across the device provide a polarized turn-of-f snubber. A resistor connected across the inductor, and diode provides a dissipative snubber diseharge path. Although the advantages of the use of snubbers in transistor inverters are well~known, paekaging problems and the cost of the additional sn~lbber components has made their commercial use infre~uent. For GTO
inverters, on the other hand, the snubber is absolutely essential for deviee protection and is o~ten erueial for reliable and success~ul inverter design. While snubbers adequately alleviate deviee switehing losses, the total switching losses do not change appreeiably when losses in the snubber are eonsidered, and can actually increase from the losses experienced in cireuits unproteeted by snubbers under eertain operating conditions. Thus, the increases in inverter switehing requency which have been obtained 67~

with the use of snubbers carry a serious penalty in terms o~ overall system efficiency.
Another alternative is a resonant mode converter employing a high frequency resonant circuit in the power transfer path. Two distinct categories of resonant inverters can be identified. The first category, of which induction heating inverters and DC/DC converters are examples, accomplish control of the power transfer through a modulation of the inverter swi-tching frequency. For lQ these ~ircuits, the frequency sensitive impedance of the resonant tank is the key to obtaininy a variable output.
~hile it is also possible to synthesi~e low frequency AC
waveforms using such fre~uency modulation principles, complexity of control, the large number of switching devices required, and the relatively large size of the resonant components limits the applications for such circuit structures.
The second type of resonant converter, sometimes referred to as a high frequency link converter, typically uses naturally commutated converters and cycloconverters with a high frequency AC link formed of a resonant LC tank circuit. The high frequency link converters are capable of AC/AC or DC/AC conversion with bi-directional power flow and adjustability of tl~e power factor presented to the AC supply. In contrast to the frequency modulation scheme of the first category of converters, the link freque~cy is not particularly important and output AC
waveform synthesis~is done throl~h modulation of the output stage. For naturally commutated switching devices, 3Q phase angle control is ordinarily used. The high frequency link converter is generalIy capable of switching at frequencies greater than 18 kHz using available devices at the multi-kilowatt power level. However, the technology has not been economically competitive and has not been widely used industrially for variable speed drive type applications. This may be attributed to several factors. In particular, the large number of 56~

bi-directional high speed, high power switches required must be realized using available uni-directional devices.
For example, as many as thirty-six thyristors may be required in addition to an excitation inverter in some configurations. The recovery characteristics of the devices used often necessitate the addition of snubber networks, lowering the efficiency of the overall system.
In addition, the LC resonant circuit handles the full load power which is transferred from input to output and has large circulating cuxrents, e.g., oEten up to six times the load current. Consequently, even though the total energy stored in the system is small, the volt-ampere rating of the resonant elements i~ quite high.
Furthermore, control of such systems is extremely complex given the simultaneous tasks of input and output control, high frequency bus regulation, and thyristor commutation for circuits employing naturally commutated thyristors.
These conventional approaches to voltage source inverter design assume an a priori relationship between the inverter losses and the inverter switching frequency.
Most of these commercial designs utilize gate turn-off devices and operate in the 1 to 2.5 ICHz frequency range for power levels between 1 and 50 kilowatts. For commerciall~ available devices, turn-on and turn-off tim~s of 1 to 2 microseconds are readily available, as are orage times of 5;to 15 microseconds, enabling these devices to switch at higher frequencies than used in conventional designs~. Although the exact switching frequency is a trade off between system performance and efflciency, commercially available designs tend to be thermally limited. In a typical design, approximately 30 to 50% of the total device losses derive from switching losses. Thus, inverter designs which reduce or eliminate 35~ ~ s~witching losses can yield several benefits. By decoupling the inverter losses from the switching ~frequency, better device utilization is permitted. Both the inverter switching frequency and the r~m.s. current :

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rating can be substantially increased befoxe thermal limitations occur. Tlle resonant converters described above can operate with lower ~witching losses but have not been widely utilized ~or the reasons discussed.
A resonant DC linX inverter design has been developed which overcomes the most serious objectio~s to the conventional resonant converters. This design is disclosed i~ U.S. patent No. 4,730,242 issued March 8, 1988 to the present applicant and entitled la Static Power Conversion Method and Apparatus Having Essentially Zero Switching Los~es. An LC resonant tank circuit i5 excited in such a way a~ to set up periodic oscillations on the inverter DC link. Under appropriate control, the DC link voltage can be made to go to zero for a controlled period of time during each cycle. During the the time that the DC link voltage goes to zero, the devices across the DC link can be turned on and of~ in a lossless manner. By eliminating device switching losses, the inverter switching frequencies can be raised to above 20 KHz at power ratings of 1 to 25 KW using commercially availible switching devices such as darlington bipolar junction transistors. Inverter operation is also compatible with uniformly sampled zero hystereis bang-bang controllers~ referred to as delta modulators. When operated with delta modulation strategies, resonant lin~
converters are capable of better perPormance than hard ; switched pulse width;modulated voltage source inverters.
The resonant DC link inverter also has a simple power structure and non-catastrophic fault mode which makes the inverter both rugged and reliable. The major limitation o the resonant DC link inverter is the imposition of device voltage stresses of 2.5 to 3 times the DC supply voltage. A discrete pulse modulation strategy ~or such resonant link inverters, such as sigma delta modulation, can also yield su~stantial spectral energy at ~requencies much lower than the resonant link Prequency. The device stresses in ~u~h resonant DC linX inverters can be reduced ' :L~9~

using clamping of the DC link voltages. The present lnvention provides an alternate circuit design to that disclosed, for example, in the aforesaid U.S. patent, which also realizes high level power conversion with essentially zero switching losses.

SUMMARY OF THE INVENTION

In accordance with the present invention, DC to AC and AC to AC power conversion is accomplished with substantially zero switching losses at switching requencies to 20 kilohertz and above over a wide range of power ratings. Current and voltage stresses imposed on the switching devices are moderate and substantial improvements are obtained in the spectral response obtained at the inverter outputs compared to conventional commercially available inverters. The apparatus cf the invention operate~ in a quasi-resonant manner, utilizing an inductor-capacitor resonant circuit, with the directly ; controlled parameter being the inductor current, thereby operating as a current mode inverter. The ~uasi-resonant current mode,inverter combines clamped voltage stresses on the switching devices with the availability of true pulse ~width modulation and sinusoidal output voltages.
In~the inverter apparatus of the invention, an ; : : ad~itional filter capacitor is connected in series with the inductor of the resonant circuit, the filter capacitor ;having a capacitance~which is substantially greater than the capacitance of the resonant capacitor so that the resonan~t frequ:ency will be determined primarily by the resonant circuit capacitor and inductor. The output power rom: the inverter is;ta~en across the filter capacitor, ; which filters out the high frequency switching components, leaving only an AC component, at a much lower frequency than the switching frequency, which has a substantially :: ~:::~ :: :

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~2~5i6~Z `-sinusoidal wave shape. A switching means is supplied with DC power and includes at least a pair of switching devices which are connected together at a node and to which the resonant circuit inductor and capacitor are also connected. Preferably, the resonant circuit capacitor is comprised of two separate capacitorfi with a resonant circuit capacitor connected across each one of the switching devices in the pair of switching devices.
For a single phase inverter, the power source may be composed of two separate voltage sources of substantially e~ual supply voltage which are connected in seri~s across the pair of switching devices and which are connected together at a node, with the resonant inductor and the filter capacitor connected between the node of the switching devices and the node of the power sources. When the switching devices are switched on, the voltage across the resonant circuit inductor and series connected filter capacitor is clamped to the voltage of one of the pair of voltage sourcesO The switch is turned off at a time when current is flowing in the switch, and not in an anti-parallel diode connected in parallel with the switch, and with zero voltage across the switch so that substantially no switching loss occurs. The current flow is then transferred to the resonant capacitor. ~ minimu~
inductor current is required at the time of switching to insure that the resonant circuit resonates to drive the voltage level back to the opposite clamping voltage, that is, to the voltage of the other of the two power sources the pair. When this voltage is reached, the second of ~the switches in the pair (which has zero voltage across it at this time) can be turned on to clamp the voltage across the series connected filter capacitor and resonant inductor at the voltage o~ the second power source. When a sufficient current is built up in the inductor to insure that the resonant circuit will resonate sufficiently to drive the voltage across the resonant circuit back to the :

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voltage of the first voltage source, the switch can be turned off and the resonant cycle repeated. For ideal inductor and capacitor elements, the minimum switching current in the inductor is chosen to be greater than or equal to the square root of the product of the voltage of each power source and the output voltage across the filter capacitor divided by the square root of the inductance of the resonant circuit inductor, all divided by the capacitance of the resonant circuit capacitor. The output voltage across the filter capacitor will be a relatively :10 slowly varying sine wave, e.g., a ~ine wave with a frequency from G0 to 400 Hz, while the voltage across the series connected induc~or and the filter capacitor, and the current through the inductor, may vary at a much higher switching frequency. For example, the current through the inductor will appear substantially as a triangular wave at a frequency which may be as much as 20,000 ~z or higher. Preferably, the switching frequency is relatively high, e.g., in the range of 18,000 Rz to 20,000 H~ or higher, so that the switching frequencies are above the audible range.

Under steady state operation when a low frequency voltage is generated across the filter capacitor, the switching fnequency of the pairs of switching devices and the duty cycle both vary continuously depending on the OlltpUt voltages across the filter capacitor and the ~5 minimum and maximum inductor curren~t envelopes which are selected to generate the desired low requency output current through the filter capacito~. These minimum and maximum current envelopes are selected to insure that the condition for sustained resonance is satisfied and so that the averaye of the minimum and maximum current envelopes approximates the desirad low frequency output current which will be impressed through the filter capacitor and which will appear as output voltage across the output terminals of the inverter. The capacitance of the resonant circ~lit capacitor or capacitors is pre~erably cbosen so that durlng the switching cycle, as the current ~2~tS6~

in the switching device which is turning off goes to zero, a relatively moderate voltage will be developed across tnat switching device, preferably as low as possible, thus reducing substantially the switching losses incurred in that switching device.
The inverter of the present invention can be extended to provide three phase output power by utilizing three pairs of switching devices connected in a bridge configuration, with the node joining each pair of switching devices being connected to a series connected resonant inductor and filter capacitor, and with the filter capacitor of each of the three phases connected together in either a wye or delta configuration. The output voltages for each of the three phases are again taken across the filter capacitors, and the resonant capacitors are preferably separate capacitors connected across each of the switching devices in the three pairs of switching devices.
A bridge configuration may also be utilized for single ; phase output, employing two pairs of switching devices and a single power supply, and with the resonant circuit and ; filter capacltor connected between the nodes joining the switching devices in each pair.
By providing AC output voltages in either of the single phase or three phase configurations, bidirectional power flow i5 allowed through the inverter and an AC to AC
converter can also be realized by using two current mode `inverters and operating them back-to-back off of the same DC bus.
E'urther objects, features and advantages of the present invention will be apparent from the following detailed description when taken in conjunction with the accompanying drawings.

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~5i67;~ -BRIEF DESCRIPTION OF THE VRAWINGS

In the drawings:
Fig. 1 is a schematic circuit diagram of a resonant pole circuit shown for purposes of illustrating the circuit operation principles of the present invention.
Fig. 2 are graphs of voltage and current waveforms in the circuit of Fig. 1.
Fig. 3 is a schematic circuit diagram of a single phase quasi-resonant current mode inverter in accordance with the present invention.
Fig~ 4 is a schematic circuit diagram of a single phase quasi-resonant current mode inverter in accordance with the invention having separate resonant circuit capacitors for each of the two switching devices.
Fig. 5 is a schematic circuit diagram of a three phase quasi-resonant current mode inverter in accordance with the invention.
Fig. 6 are graphs of current and voltage waveEorms for the inverter of Fig. 3.
Fig. 7 is an illustrative graph showing the mannner in which a desired inductor current can be synthesized by specifying minimum and maximum current envelopes for the inductor current, yielding an average low frequency AC
inductor current.
Fig~ 8 is a bIock diagram showing a controller which ~provides the switching control signals to the switching ~; ~ dsvices of a quasi-resonant current mode inverter circuit in accordance with the invention.
Fig. 9 is a schematic circuit diagram of an AC to AC
converter in accordance with the present invention which utllizes two quasi resonant current mode inverters which interface between two three phase systems.

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DESCRIPTION OF THE PREFERRED EMBODIMENTS

With reference to the drawin~s, a resonant pole :~: : :

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6~2 circuit help~ul in explaining the circuit principles of the present invention is shown in Fig. 1. This circuit can be used to provide dc to ac power conversion at the switching frequency and dc to dc power conversion but is not capable of dc to ac conversion at an output frequency lower than the switching frequency. The circuit includes two voltage sources 11 and 12 of essentially identical voltage level Vs, a pair of switching devices 13 and 14 (illustrated as bipolar transistors with anti-parallel 10 diod~s 15 an~l 16) configured in an inverter pole and a resonant circuit composed of a parallel inductor 17 and capacitor 17 connected between the node 17 at which the two switching devices 13 and 14 are connected together and the node 20 at which the two voltage sources 11 and 12 are connected. A load 21 is connected across the resonant circuit to receive the voltage across the inductor. The voltage level stresses imposed on the switching devices are clamped at the voltage Vs of the voltage supplies 11 and 12. In operation, the first switch 13 is turned on until the current iL in the inductor 17 is positive in the direction shown and equals a reference value I .
The switch 13 is turned off at this point to transfer the current in the inductor iL to the capacitor 1~, and the resonant pole voltage Vm, the voltage between the nodes 18 and 19, reverses to the level of the negative supply voltage, -Vs, in a resonant manner. By choosing an appropriate value for capacitance of the capacitor 18, a zero voltage turn off for the switch 13 can be attained.
When the pole voltage Vm reaches minus Vs~ the diode 3~ 16 in parallel with the switch 14 turns on, whereupon the the ~witch~14 can be turned on in a lossless manner. With the switch 14 and/or the anti-parallel diode 16 conducting, the inductor current iL decreases linearly until it reaches a value -Ip, at which time the switch 14 can be turned off, and the resonant cycle repeated.
Fig. 2 shows graphs illustrating the pole voltage Vm .

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wave form 23 and the inductor current iL wave form 24.
The frequency of the output voltage Vm which is delivered to the load 21 will thus be at the switching frequency of the devices 13 and 14. The load 21 can comprise a ~ull wave rectifier which delivers DC voltage to a power consuming load. However, the circuit of Fig. 1 does not provide variable low frequency output voltage to tne load and thus cannot serve generally as a DC to AC
inverter.
In accordance with the present invention, an inverter utilizir~g a re~t~nant pole for DC to low frequency P.C
inverter operation is shown generally at 30 in Fig. 3.
l'he inverter 30, which may be designated a quasi-resonant current mode inverter, includes power supplies 31 and 32 of substantially equal supply voltage Vs, an inverter pole composed of switching devices 34 and 35 (which as shown, may be composed, respectively, o~ gate controlled devices such as the bipolar transistor 37 and anti-parallel diode 38 and bipolar transistor 39 and 2Q anti-parallel diode 40), with the voltage sources 31 and 32 connected together at a node 42 and the.switching devices 34 and 35 connected together at a node 43. A
resonant circuit composed of a capacitor 45 and inductor 46 is conneoted in parallel between the nodes 43 and 42 and, in addition, a filter capacitor 47 is connected in series with the inductor 46 between the nodes 43 and 42.
The output voltage from the inverter is taken across the filter capacitor 47 and i~ supplied to a load : illustratively shown as a resistor 49. The phase voltage Vm between the nodes 43 and 42 is shown by the graph : labeled Sl in Fig. 6 and the current iL through the inductor 46 is shown ~y the graph labeled 52 in Fig. 6.
To illustrate the principles of operation of this circuit, the case may first be considered where it is desired to generate a DC level output voltage VO across the load resistance 49. If the positive and negative trip currents I~ at which switching ta~es place are changed so that the average inductor current is equal to Vo/R, (where R
is the resistance of the resistor 49) the output voltage will then be held at VO' It is apparent that this relationship holds true for all polarities of load voltage and current, indicatin~ that the circuit 30 can operate as an inverter. Furthermore, as the directly controlled parameter is the inductor current, the circuit 30 may be referred to as a current mode inverter.
The operation of the inverter 30 can be viewed as composed of two distinct modes, which repeat every half cycle, with Fig. 6 showing the wave ~orms or the synthesis of an AC current illustrating the relationships between phase voltage and inductor current during the two modes. In the first mode, during which one or the other of the switches 34 and 39 is turned on, the phase voltage Vm, which is also the voltaye imposed across the capacitor 45, is equal to the supply voltage Vs (or minus Vs). The state equations for the inductor current iL and output voltage VO are:
~ iL (VS - Vo)/L
O ( L O/ )/C~, where L is the inductance of the inductor 46, R is the resistance of the resistor load 49, and Cf is the capacitance of the filter capacitor 47.
In the second mode, the switches 34 and 35 are turned off, and the voltage across the resonant capacitor 45 is no longer clamped to the supply voltage. The state aquation for the phase voltage Vm is thus 3Q Vm = -iL/Cr, where Cr is the capacitance of the resonant capacitor 45. ~
The switching conditions are determined based on two constraints. The first constraint is zaro voltage switching, which requires that current be flowing in the switch to be turned off, and that the current be sufficient to insure that the pole voltage Vm reach the opposite voltage supply level. ~ssuming that the i6~2 `--capacitance of the filter capacitor is much greater than the capacitance of the resonant capacitor, by applying conservation of energy over the first mode, the minimum value ILmin of inductor current required to reverse the pole voltaye is:
IL~in = 2 _ ~ VSVO
z where Z0 = (L/Cr)l/2. This equation assumes lossless i~ductive and capacitive components. The actual value o~ ILmin w~uld have to be greater than the value computed by the equation above to ~mpensate for losses in the inductive and capacitive components.
A second constraint on system performance is the variation o~ the switching frequency with the output voltage V0. Assuming that the transitions between turn c f of one device and turn on of the other form a small part of the entire cycle, the inductor current can be assumed to be substantially triangular in form. Inasmuch as V0 varies slowly compared to the phase voltage V , for purposes of analyzing this constraint the output voltage can be assumed to be substantially DC. For an output current Io = Vo/RI the switching period T can be calculated to be at least (assuming large Z0):

T = 4 ~,z (V~ - V ) The equation above for T implie.s that for a given load curr~ent level, as the output voltage VO varies from 0 to 0.8 Vs,~the switching frequency F varies from a maximum, FmaX, to 0.36 FmaX. If the output voltage increases to a.s vs~ the switching frequency goes to 0.19 FmaX, ~and if the output voltage VO equals the ~supply voltage level Vs, zero frequency results. This indicates that the lowest desirable switching fre~uency sets a limit on the maximum output voltage (modulation index) obtainable.
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For a sinusoidally varying output voltage VO~ it can be seen that the instantaneous switching frequency would sweep from a maximum value at VO = 0 to a minimum at the peak voltage point.
A single p'hase quasi resonant current mode inverter which has a different circuit structure but is otherwise equivalent to the circuit o~ Fig. 3 is shown generally at 60 in Fig. 4. The circuit 60 has two power sources 61 and 62, each providing output voltage Vs which are connected together at a node 63, and an inverter pole composed of a pair of switching devices 64 and 65 which are connected together at a node 66. In the inverter 60 of Fig. 4, the switching device 64 is illustratively shown composed of a ~ower MOSFET 68 and anti-parallel diode 69 and the switching device 65 as a power MOSFET 70 with anti-parallel diode 71. ~s in the inverter 30 of Fig. 3, the resonant circuit inductor 73 is connected in series with a filter capacitor 74 across the phase terminals 66 and 63 and the output voltage VO is provided across the 2Q filter capacitor 74 to a load 75. However, in the circuit 60, the single resonant capacitor 45 of the circuit 30 of Fig. 3 is split into two substantially equal resonant capacitors 77 and 78 which are connected across, respectively,, the switching devices 68 and 70. It is apparent that the resonant circuit composed of the resonant capacitors 77 and 78 and the resonant inductor 73 is entirely equivalent to the resonant circuit in the inverter 30 composed of the resonant capacitor 45 and ~ resonant inductor 46 assuming that the power sources 61 and 62 are ideal and have substantially no internal impedance. It is seen that t'he capacitor 77 and 78 are e~fectively connected in paraIlel with one another across the phase nodes 66 and 63. Thus, if each of the capacitors 77 and 78 have one half of the capacitance of the resonant capacitor 45 of Fig. 3, an equivalent re.~onant frequency is obtained for bot'h of the inverters ~5;6~2 --30 and 60, assuming that the inductors 46 and 73 have the same inductance and that the filter capacitors 47 and 7~
are substantially larger in capacitance than the resonant capacitor so that the filter capacitors do not substantially affect the resonant frequency of the resonant circuit.
A three phase reali7ation of a quasi resonant current mode inverter is shown generally at 80 in Fig. 5. The inverter 80 has a pair of power sources 81 and 82 of voltage level Vs connected together at a node 83 and three pairs of switching devices 85-90 each composed of a bipolar transistor and anti-parallel diode, with the three pairs being connected together at nodes 92-94 in a bridge configuration. Resonant capacitors 96-101 are connected across the switching devices 85-90, respectively, and the three phase nodes 92 94 have resonant inductors 104-106 connected thereto and supply three phase output terminals 107-109 respectively. The output terminals 107-109 have respective filter capacitors 110-112 connected thereto, with each of the capacitors 110-112 being connected together to a common node line 114 which may optionally be connected by a conducting line 115, shown in dash lines in Fig. 5, to the node 83 joining the voltage sources 81 and 82. However,, the conducting line llS is not required, and if eliminated, the filter capacitors 110-112 may be connected together in either a wye configuration as shown, or a delta configuration. If the conducting line 115 is not present, the split voltage sources 81 and 82 may be combined into a single voltage source. The three phase circuit of Fig. 5 operates under the same conditions and ~ in the same manner as described above for the circuit of Fig. 3, with the~three phase currents being individually monitored to meet the conditions specified above. Of course, it is also apparent that a full bridge single phase output may be acheived in accordance with the ~5 present invention utilizing two of the pairs of switching 567~

devices of Fig. 5 rather than all three. For a single phase full bridge circuit, the output would be taken across two terminals, say terminals 107 and 108, in which case the resonant capacitors 110 and 111 may be combined into a single equivalent capacitor and the resonant inductors 104 and 105 ma~ be combined into a single equivalent inductor.
A further extension of the invention to an AC to AC
converter can be realized by utilizing two current mode inverters operating back-to-back off of the same DC bus.
Such an AC to AC conversion system is shown generally at 120 in Fig. 9. T'he converter L20 has a three phase output inverter having switching devices 121-126 connected in pairs with resonant capacitors 12~-133 connected across the switching devices. l'he three phase output from the connections between the pairs of switching devices is provided to resonant inductors 135-137 and filter capacitors laO-142 are connected across the three phase output terminals which are available for connection to a three phase load-source 143. The inverter receives DC
power from DC bus lines 145 and 146, across which is connected a large filter capacitor 147 to filter out the ripples on the DC bus. A rectiEying converter composed o~
switching devices 150-155 is also connected in pairs across the DC bus lines 14S and 146 and has resonant capacitors 157-162 connected across the switching devices. The nodes connecting the two switching devices in each pair are connected to resonant inductors 165-167 30~ and filter capacitors 168-170 are connected applied to the three phase input terminaIs which are connected to a three phase power source 172. The three phase converter receives the three phase power from the source 172 and ;3 ~ ~ converts it to, DC power across the bus lines 145 and 146.
Reverse directional transfer of energy from the load 143 to the source 172 can be obtained b~ reversing the fvnctlon of the invereer and converter so that the ~l2~567;2 inverter composed of the switching devices 121-126 functions to rectify power provided from the load 143 to DC power on the lines 145 and 146, and the converter composed of the devices 150-155 can be switched in a proper fashion to invert the DC power on the bus lines to AC power which is supplied to the source 172.
The proper control conditions for quasi resonant current mode inverters in accordance with the present invention can be further understood with reference to the single phase circuit of Fig. 3. As noted above, the capacitance of the filter capacitor 47 is sufficiently high that this capacitor essentially passes the high frequency component of the inductor current so that no substantial high frequency voltage is developed across the filter ca~acitor. Elowever, the output voltage across the filter capacitor 47, VO' has a substantial low frequency content which is the desired output voltage. Preferably, the capacitance of the filter capacitor 47 is at least an order of magnitude greater than the capacitance of the resonant capacitor 45. Such a condition insures that the high frequency ripple in the output voltage VO is maintained at reasonably low levels, which is an important consideration for proper modulation.
Because!the output voltage VO varies at a low frequency, it can be at a voltage level substantially different from zero over a large number of switching cycles. Consequently a net transfer of energy from the inductor to the filter capacitor 47 is required over each switching cycle. To generate a low frequency wave form across the filter capacitor and the load connected in parallel with it, the inductor current iL must be controlled such that the low frequency component of current through the filter capacitor generates the desired ~ output voltage. Thus, the capacitance of the filter capacitor is chosen so as to filter out the high switching frequency components bu-t still support the low frequency : ~:
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~2~ 2 ~20-component. The modulation strategy controlling the switching of the switches 34 and 35 must satis~y the requirement for maintaining the necessary minimum current iL~in to allow each resonant cycle to continue. ~
maximum current envelope ILmaX, then determines, with the minimum current envelope, the resulting low frequency output current I which is desired by the designer. The minimum and maximum current envelopes and the resulting low frequency output current wave form I are shown lo illustratively in Fig. 7. Under steady state operation, where a low frequency sinusoidal voltage is being generate~ across the filter capacitor 47, the switching frequency and the duty cycle will both vary continuously, as illustrated in Fig. 7, depending on both the desired output voltage VO and the ILmin and ILmax e p which are selected. These envelopes are chosen so as to insure the switching condition given above and to cause the average of the minimum and maximum envelopes to approximate the desired output current wave form I .
The value of the resonant capacitor or capacitors is chosen so that during the switching cycle, as the current in the device which is turning off goes to zero, a moderate voltage will be developed across the device tpreferably as low as possible) to reduce to as great an extent as possible the switching losses incurred in the device.
An exemplary controller arrangement for the inverter of the invention is shown in Fig. 8, wherein the controller receives as input variables the actual time varying inductor current IL (received from a current sensor in series with the inductor 46), the desired low frequency componet of the inductor current I , and the actual output voltage VO across the filter capacitor ; (received from a voltage sensor). The value of the output voltage VO is operated on at 180 to compute a minimum value of the inductor current required ILmin, which may ;

~5i6~2 be computed in accordance witll the criterion given above for the minimum required current level. Depending on the circuit conditions, the output of the block 180 is a minimum current ILmin which is selected to insure the desired maintenance of resonance in the circuit, and the value of ILmin and the desired low frequency component I are then utilized as input variables for a calculation at 181 of ILmaX. This calculation is carried out so as to yield an ILmaX which results in the *g ILmax and ILmin being approximately equal to I The ILmax value is then compared with the actual inductor current value IL in a comparator 182 to determine a maximum switching point-and the ILmin value is compared with the actual current IL in a comparator 183 to determine the minimum switching point. These values are then used in a conventional fashion to provide switching inputs to the switching devices 34 and 35.
Preferably, the controller of Fig. 8 is implemented as a microprocessor based programmable controller which carries out the calculations shown in the blocks 180 and 181 utilizing software algorithms in a conventional fashion well known in the art. Although a mircroprocessor system allowing reprogrammable software to be utilized is preferred, the blocks 180 and 181 may also be implemented in a conventional fashion using hard wired circuit components.
As an example of the implementation of the present invention, an inverter topology in accordance with the three phase implementation of Fig. 5 was fabricated and tested which operated off of a 150 volt DC bus at peak load currents of 30 amperes. Bipolar switching transistors with anti-parallel diodes were utilized as the ; switching devices. Each of the resonant inductors had an inductance of ~0 microhenries, each resonant capacitor had a capacitance of 0.25 microfarads, and each filter 3S capacitor had a capacitance of 30 microfarads. The ~2~5~7;~: -inverter was used as non-interruptable power supply inverter with sinusoidal output, being switched at approximately 25 KHz under no load conditions which reduced to 12 KHz at full load. A pulse width modulation (PWM) strategy was utilized as discussed above. The value i~ ILmin was preselected to correspond to the maximum value of VO to be obtained during normal operation. The lowest spectral content of the output voltage VO to the load was approximately 10 KHz, the average switching losses were less than 10% of the switching losses in a conventional PWM inverter, the maximum voltage stress imposed on the switching devices was e~ual to the supply voltage Vs and the maximum current stress was slightly more than two times the maximum output current while the r~m.s. current ratio was approximately 1.2 times the output current. The output voltage was substantially sinusoidal.
It is apparent that a wide variety of gate controlled switching ~evices may be utilized as the switching devices in the present invention. These include power MOSFETs, gate turn o~f thyristors, bipolar transistors, and bipolar darlington transistors, which may be commercially packaged with anti-parallel diodes or which may include inherent parasitic anti-parallel diodes.
It is also understood that the invention is not confined to the partlcular embodiments set forth herein, but embraces all such ~orms thereof as come within the scope of the following claims.

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Claims

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE PROPERTY
OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

1. Quasi-resonant current mode power conversion apparatus comprising:
(a) a resonant circuit having a resonant inductor and a resonant capacitor connected in a parallel configuration with respect to each other;
(b) a filter capacitor connected in series with the resonant inductor and having substantially greater capacitance than the resonant capacitor, the output voltage across the filter capacitor being the output voltage of the power conversion apparatus;
(c) a switching circuit including at least two switching devices, which are controllable to be turned on and off, connected together at a node to which the resonant inductor and capacitor are connected;
(d) control means for switching the switching devices to cause a stable resonant oscillation in the resonant circuit when the switching circuit is provided with a supply voltage such that a high frequency AC resonant voltage oscillation is maintained across the resonant circuit and a lower frequency sinusoidal oscillation is maintained across the filter capacitor, the control means switching the switching devices on and off when the inductor current reaches selected minimum and maximum envelope values such that the average value of the inductor current yields a desired waveform which is at a substantially lower frequency than the frequency of switching of the switching devices.

2. Quasi-resonant current mode power conversion apparatus comprising:
(a) power supply means for supplying DC output power;
(b) a resonant circuit having an inductor and at least one capacitor connected in a parallel configuration with respect to each other;

(c) a filter capacitor connected in series with the resonant inductor and having substantially greater capacitance than the resonant capacitor, the output voltage across the filter capacitor being the output voltage of the power conversion apparatus;
(d) A switching circuit including at least one pair of switching devices connected together which are controllable to be turned on and off, and connected to the power supply means and to the resonant circuit;
(e) control means for switching the switching devices at the proper times to cause the resonant circuit to oscillate at a high frequency which is filtered out by the filter capacitor and does not substantially appear as a voltage across the filter capacitor and to also oscillate at a second lower frequency which appears as an alternating voltage across the filter capacitor, the control means switching the switching devices on and off when the inductor current reaches selected minimum and maximum envelope values such that the average value of the inductor current yields a desired waveform which is at a substantially lower frequency than the frequency of switching of the switching devices.

3. The power conversion apparatus of claim 1 including power supply means for providing DC power to the switching circuit.

4. The power conversion apparatus of claim 1 or 2 wherein the resonant capacitor comprises a resonant capacitor connected across each switching device.

5. The power conversion apparatus of claim 1 or 2 wherein the switching devices in the switching circuit each include a gate controlled switching device.

6. The power conversion apparatus of claim 5 wherein the switching devices each comprise a bipolar junction transistor and an anti-parallel diode connected thereto.

7. The power conversion apparatus of claim 2 or 3 wherein the power supply means includes a split power supply having two substantially equal voltage sources connected in series at a node and wherein the switching circuit comprises two switching devices connected in series at a node, and wherein the resonant circuit and the filter capacitor are connected between the power supply node connection and the node connection between the switching devices.

8. The power conversion apparatus of claim 2 or 3 wherein the switching circuit comprises at least two pairs of series connected switching devices with each pair being connected at a node in a bridge configuration, the bridge of switching devices receiving the supply voltage from the power supply means, and wherein the resonant circuit and filter capacitor are connected between the nodes connecting the pairs of switching devices.

9. The power conversion apparatus of claim 2 or 3 wherein the power supply means comprises converter means connected to an AC power system for rectifying the AC power from the power system and supplying DC output power to the switching circuit.

10. The power conversion apparatus of claim 9 wherein the converter means includes controllable switching devices connected in a bridge configuration and controlled such that the converter means can selectively rectify power from the AC
power system and supply it to the switching circuit and selectively receive DC power from the switching circuit and convert it to AC power supplied to the AC power system.

11. The power conversion apparatus of claim 1 or 2 wherein switching of the switching devices in the switching circuit is controlled to take place at times when there is no less than a predetermined minimum current flowing through the resonant inductor, the minimum current being that which is required to complete the resonant cycle in the resonant circuit after switching off of the switching devices in the switching circuit.

12. The power conversion apparatus of claim 11 wherein the predetermined minimum current is greater than or equal to where ZO= (Lr/Cr)1/2, Vs is the DC voltage provided by the power supply means, V0 is the output voltage across the filter capacitor, Lr is the inductance of the resonant inductor and Cr is the capacitance of the resonant capacitor.

13. The power conversion apparatus of claim 2 or 3 wherein the switching circuit comprises three pairs of controllable switching devices connected in a bridge configuration and receiving the DC power from the power supply means across the bridge, the switching devices in each pair connected together at a node, and wherein the resonant circuit comprises resonant capacitors and resonant inductors connected between the nodes of the switching devices such that there is a resonant capacitor and resonant inductor connected between each two of the nodes.

14. The power conversion apparatus of claim 1 or 2 wherein each of the switching devices is a gate controlled switching device and the control means provides control signals to switch the switching devices on and off, the control means switching off the switching devices only when current is flowing through the switching devices.

15. The power conversion apparatus of claim 1 or 2 wherein the filter capacitor has a capacitance of at least ten times greater than the capacitance of each resonant capacitor.

16. A method for converting a DC supply voltage from a DC
power source at an AC voltage, utilizing a resonant circuit comprised of a resonant inductor and a resonant capacitor connected in parallel, and a filter capacitor connected in series with the resonant inductor which is substantially larger in capacitance than the resonant capacitor, comprising the steps of:
(a) applying the DC supply voltage across the series connected resonant inductor and filter capacitor at a first polarity for a time sufficient to build up a desired current level in the inductor;

(b) then removing the supply voltage from the series connected inductor and filter capacitor to cause the current in the inductor to flow into the resonant capacitor until the voltage across the resonant inductor and filter capacitor is equal to the supply voltage at the opposite polarity;
(c) then applying the DC supply voltage to the resonant conductor and filter capacitor at the opposite polarity which matches the polarity of the voltage across the resonant circuit and filter capacitor at that time for a time sufficient to build up a selected current level in the inductor;
(d) then removing the supply voltage from the resonant inductor and filter capacitor to cause the current in the inductor to flow through the resonant capacitor until the voltage across the resonant inductor and filter capacitor is equal to the supply voltage of opposite polarity; and (e) then repeating steps (a) through (d) above;
wherein steps (a) through (e) are carried out such that the time of application of the supply voltage in each polarity to the resonant inductor and filter capacitor and the frequency of switching from one polarity to the other are selected to result in a time varying voltage appearing across the filter capacitor at a frequency substantially lower than the frequency of switching between the two polarities of the power supply voltage, and such that substantially no voltage varying at the switching frequency appears across the filter capacitor.

17. The method of claim 16 wherein the switching of the polarity of the supply voltage to the resonant inductor and filter capacitor is done by gate controlled switching devices having anti-parallel diodes connected thereto, and wherein the steps of removing the supply voltage is done by switching off the switching devices only when current is flowing through the switching devices.