CA2049501C - Method for channel adaptive detecting/equalizing - Google Patents
Method for channel adaptive detecting/equalizingInfo
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- CA2049501C CA2049501C CA002049501A CA2049501A CA2049501C CA 2049501 C CA2049501 C CA 2049501C CA 002049501 A CA002049501 A CA 002049501A CA 2049501 A CA2049501 A CA 2049501A CA 2049501 C CA2049501 C CA 2049501C
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- equalizer
- detection
- detection algorithm
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/22—Demodulator circuits; Receiver circuits
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B7/00—Radio transmission systems, i.e. using radiation field
- H04B7/005—Control of transmission; Equalising
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- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
Abstract
In a communications device, two coherent detection al-gorithms (102 and 103), one of which has a decision feedback equalizer (103), and a detector selection algorithm (104) are used to dynamically select a detector depending on whether delay spread distortion is present. First the correlation of the detector without the equalizer (102) is measured. If this corre-lation is greater than a predetermined threshold, the data from that detector (102) is used by the communications device.
If the correlation is less than the threshold, the correlation of the detector with the equalizer (103) is measured. If this is less than the correlation of the detector without the equalizer (102), the data from the detector without the equalizer (102) is used, otherwise the data is taken from the equalizer (103).
If the correlation is less than the threshold, the correlation of the detector with the equalizer (103) is measured. If this is less than the correlation of the detector without the equalizer (102), the data from the detector without the equalizer (102) is used, otherwise the data is taken from the equalizer (103).
Description
METHOD FOR CHANNEL ADAPTIVE
DETECTING/EQUALIZING
Field of the Invention The present invention relates generally to communica-tion devices in a time varying propagation environment.
Ba~k~round of the Invention Any modulation method can be represented by a con-stell~tion. An e~mple of this is the eight point constellation illustrated in Fig. 14. This constellation is generated from dif-ferentially encoded QPSK (DEQPSK or ~/4 QPSK) which is a 1 5 subset of the four state QPSK constellation. Each state is characterized by a vector having the same magnitude, but a different phase angle. In a Rayleigh faded ~h~nnel, the con-stellation has continuous but slowly varying, with respect to the data rate, ~mplitude and phase modulation imposed on it.
DETECTING/EQUALIZING
Field of the Invention The present invention relates generally to communica-tion devices in a time varying propagation environment.
Ba~k~round of the Invention Any modulation method can be represented by a con-stell~tion. An e~mple of this is the eight point constellation illustrated in Fig. 14. This constellation is generated from dif-ferentially encoded QPSK (DEQPSK or ~/4 QPSK) which is a 1 5 subset of the four state QPSK constellation. Each state is characterized by a vector having the same magnitude, but a different phase angle. In a Rayleigh faded ~h~nnel, the con-stellation has continuous but slowly varying, with respect to the data rate, ~mplitude and phase modulation imposed on it.
2 0 A coherent detector has the ability to track the phase modula-tion and cancel it, generating the original eight point constel-lation. Although the envelope or amplitude modulation still rem~in~ imposed on the signal, it is not a problem for data re-covery since all the data information is contained in the phase 2 5 shifts.
In a delay spread field, the ch~nn~l has the same char-acteristics but the amount of distortion is increased due to the delayed versions of the signal adding at the receiver input with their own uncorrelated Rayleigh fading statistics. This 3 0 fading can be caused by a transmitted signal bolln~ing off a tall building or other structure. This condition is typically re-ferred to as multipath distortion or delay spread distortion.
This type of distortion results in a frequency null or notch in the spectrum of the modulation that severely degrades the per-3 5 formance of the communication system.
Coherent detectors are typically used in digital commu-nications for the symbol decision process. Coherent detection i8 described in BERNARD SKLAR, DIGITAL COMMUNICATIONS, FUNDAMENTALS AND APPLICATIONS, Chapter 3 (1988). In a 5 delay spread distortion environment, however, a detector alone cannot remove the distortion caused by the same signal being received at multiple times. This type of detection re-quires a detector with a channel equalizer. A typical çh~nn~l equalizer structure is the decision feedback equalizer (DFE).
1 0 The DFE can phase track and cancel the distortion caused by the delayed version of the signal. Once the distortion is can-celled, the eight point constellation is generated at the detector output. DFES are described in JOHN PROAKIS, DIGITAL
COMMUNICATIONS, Chapter 6 (1989). The ~h~nnel equalizer, 1 5 however, should not be used in an environment without delay spread distortion as it can cause sensitivity degradation com-pared to a coherent detector without the equalizer. There is a resulting need for a coherent detection algorithm that works equally well in an environment with and without delay spread 2 0 distortion.
S~lmm~qrv of the Invention The present invention is comprised of two coherent de-2 5 tection algorithms, one of which has a decision fee-lbiqck equalizer, and a detector selection algorithm. The process of the present invention is comprised of the steps of first measur-ing the correlation of the detector without the eqll~ er. If this correlation is greater than a predetermined threshold, 3 0 the data from that detector is used by the communications de-vice. If the correlation is less than the threshold, the correla-tion of the detector with the equalizer is measured. If this i8 less than the correlation of the detector without the equalizer, the data from the detector without the equalizer is used, 3 5 otherwise the data is taken from the equalizer.
204950 ~
Brief Description of the Drawin~s Fig. 1 shows the preferred emboAiment of the present 5 invention.
Fig. 2 shows the symbol format of the data input to the detectors.
Fig. 3 shows a flowchart of the present invention.
Fig. 4 shows an alternate embodiment of the present in-10 vention process.
Fig. 5 shows a flowchart in accordance with the alter-nate emboAiment of Fig. 4.
Fig. 6 shows a typical application of the preferred em-bodiment of the present invention in a radiotelephone.
1 5 Figs. 7 - 13 show other alternate embodiments of the present invention.
Fig. 14 shows the eight point constellation for a recov-ered sign~l, 2 0 Detailed Description of the Preferred Embodiment The different coherent detectors of the present invention enable relatively distortion-free operation of a digital commu-nication device in environments with and without delay 2 5 spread distortion on the communication channel. By using multiple coherent detection algorithms, one having a channel equalizer, the detector with the highest constellation correla-tion can be used.
As illustrated in Fig. 1, the ~refe.. ed embodiment of the 3 0 present invention is comprised of two coherent detectors (102 and 103) coupled to the outputs of AID converters (101). There are two A/D converters (101) that are fed from two separate outputs from the Zero IF receiver (107). The data output format of the A/D co-lve- lers (101) that is fed into each detector 3 5 (102 and 103), illustrated in Fig. 2, is in a series of frames of 20495û ~
, 160 symbols each, each frame having an initial sync sequence (201 and 202). Each detector (102 and 103) has a separate path to the AID collvel lers (101) and one of the detectors (102 and 103) has a decision feedback equalizer (DFE) algorithm (103).
The detectors (102 and 103) provide correlation information (105 and 106) to a selection algorithm (104) that determines which detector output is to be used.
The selection algorithm (104) is based on the quad-riphase shift keying (QPSK) signal constellation correlation 1 0 between the coherently detected constellation points and a valid eight-point QPSK constellation. The signal constellation corresponds to differentially encoded QPSK(DEQPSK or ~J4 QPSK) which is a subset of the four state QPSK constellation.
Of course, if an alternate modulation method were used, a 1 5 signal constellation correlation could be obtained relating to that form of modulation. A quality factor is determined by al-lowing each detector to accumulate a QPSK constellation cor-relation history for a number of symbols. The number of points chosen to be in the correlation history is a trade-off de-2 0 cision. If the number of estimates used in the averaging pro-cess is too small, the correlation estimate becomes noisy. By increasing the length of the estimate, the contribution to the error signal due to noise is reduced. Also, since the çhAnnel is dyn~mic~lly ~h~nging, the individual detectors will reflect 2 5 this in terms of their performance. If too much averaging is used, the switch will be based on past est;mAt~s that do not apply to the present performance of the individual detectors as the ch~nnel changes and switching errors become more likely. So the choice of the window length should be based 3 0 upon the channel conditions encountered.
The amount of delay in the delay blocks (108 and 109) depends on the length of the QPSK correlation history. If the correlation history is chosen to be one frame, then the delay would be 160 symbol times. If a switch selection is made N
3 5 times per frame, then the delay would be 160/N symbol times.
In a delay spread field, the ch~nn~l has the same char-acteristics but the amount of distortion is increased due to the delayed versions of the signal adding at the receiver input with their own uncorrelated Rayleigh fading statistics. This 3 0 fading can be caused by a transmitted signal bolln~ing off a tall building or other structure. This condition is typically re-ferred to as multipath distortion or delay spread distortion.
This type of distortion results in a frequency null or notch in the spectrum of the modulation that severely degrades the per-3 5 formance of the communication system.
Coherent detectors are typically used in digital commu-nications for the symbol decision process. Coherent detection i8 described in BERNARD SKLAR, DIGITAL COMMUNICATIONS, FUNDAMENTALS AND APPLICATIONS, Chapter 3 (1988). In a 5 delay spread distortion environment, however, a detector alone cannot remove the distortion caused by the same signal being received at multiple times. This type of detection re-quires a detector with a channel equalizer. A typical çh~nn~l equalizer structure is the decision feedback equalizer (DFE).
1 0 The DFE can phase track and cancel the distortion caused by the delayed version of the signal. Once the distortion is can-celled, the eight point constellation is generated at the detector output. DFES are described in JOHN PROAKIS, DIGITAL
COMMUNICATIONS, Chapter 6 (1989). The ~h~nnel equalizer, 1 5 however, should not be used in an environment without delay spread distortion as it can cause sensitivity degradation com-pared to a coherent detector without the equalizer. There is a resulting need for a coherent detection algorithm that works equally well in an environment with and without delay spread 2 0 distortion.
S~lmm~qrv of the Invention The present invention is comprised of two coherent de-2 5 tection algorithms, one of which has a decision fee-lbiqck equalizer, and a detector selection algorithm. The process of the present invention is comprised of the steps of first measur-ing the correlation of the detector without the eqll~ er. If this correlation is greater than a predetermined threshold, 3 0 the data from that detector is used by the communications de-vice. If the correlation is less than the threshold, the correla-tion of the detector with the equalizer is measured. If this i8 less than the correlation of the detector without the equalizer, the data from the detector without the equalizer is used, 3 5 otherwise the data is taken from the equalizer.
204950 ~
Brief Description of the Drawin~s Fig. 1 shows the preferred emboAiment of the present 5 invention.
Fig. 2 shows the symbol format of the data input to the detectors.
Fig. 3 shows a flowchart of the present invention.
Fig. 4 shows an alternate embodiment of the present in-10 vention process.
Fig. 5 shows a flowchart in accordance with the alter-nate emboAiment of Fig. 4.
Fig. 6 shows a typical application of the preferred em-bodiment of the present invention in a radiotelephone.
1 5 Figs. 7 - 13 show other alternate embodiments of the present invention.
Fig. 14 shows the eight point constellation for a recov-ered sign~l, 2 0 Detailed Description of the Preferred Embodiment The different coherent detectors of the present invention enable relatively distortion-free operation of a digital commu-nication device in environments with and without delay 2 5 spread distortion on the communication channel. By using multiple coherent detection algorithms, one having a channel equalizer, the detector with the highest constellation correla-tion can be used.
As illustrated in Fig. 1, the ~refe.. ed embodiment of the 3 0 present invention is comprised of two coherent detectors (102 and 103) coupled to the outputs of AID converters (101). There are two A/D converters (101) that are fed from two separate outputs from the Zero IF receiver (107). The data output format of the A/D co-lve- lers (101) that is fed into each detector 3 5 (102 and 103), illustrated in Fig. 2, is in a series of frames of 20495û ~
, 160 symbols each, each frame having an initial sync sequence (201 and 202). Each detector (102 and 103) has a separate path to the AID collvel lers (101) and one of the detectors (102 and 103) has a decision feedback equalizer (DFE) algorithm (103).
The detectors (102 and 103) provide correlation information (105 and 106) to a selection algorithm (104) that determines which detector output is to be used.
The selection algorithm (104) is based on the quad-riphase shift keying (QPSK) signal constellation correlation 1 0 between the coherently detected constellation points and a valid eight-point QPSK constellation. The signal constellation corresponds to differentially encoded QPSK(DEQPSK or ~J4 QPSK) which is a subset of the four state QPSK constellation.
Of course, if an alternate modulation method were used, a 1 5 signal constellation correlation could be obtained relating to that form of modulation. A quality factor is determined by al-lowing each detector to accumulate a QPSK constellation cor-relation history for a number of symbols. The number of points chosen to be in the correlation history is a trade-off de-2 0 cision. If the number of estimates used in the averaging pro-cess is too small, the correlation estimate becomes noisy. By increasing the length of the estimate, the contribution to the error signal due to noise is reduced. Also, since the çhAnnel is dyn~mic~lly ~h~nging, the individual detectors will reflect 2 5 this in terms of their performance. If too much averaging is used, the switch will be based on past est;mAt~s that do not apply to the present performance of the individual detectors as the ch~nnel changes and switching errors become more likely. So the choice of the window length should be based 3 0 upon the channel conditions encountered.
The amount of delay in the delay blocks (108 and 109) depends on the length of the QPSK correlation history. If the correlation history is chosen to be one frame, then the delay would be 160 symbol times. If a switch selection is made N
3 5 times per frame, then the delay would be 160/N symbol times.
It is also conceivable to make a switch selection based on a constellation history over a plurality of past frames.
The decision on whether to use the limited output of the coherent detector is done in the following m~nner. The output 5 of the coherent detector without the equalizer (102) is defined as vt(t) and the detector with the equalizer (103) is defined as ve(t). The detected output of the the first detector is therefore represented by vd(t) and the latter by ve(t). Now let 1 0 Cd(t) = E(Vd(t)~d(t)) Ce(t) = E~ve(t)~e(t)}
where Ct and Ce are the correlation functions between the de-tected DEQPSK constellation points and the actual received 1 5 signal and E{ } denotes the en~emble average. The above true correlations can be estimated from a finite length data se-quence:
No d(t) N p ~ vd(n)V*d(n-t) 0 n=P+l No Ce (t) N - P ~ Ve (n)ve*(n-t) n=P+ 1 where No represents the window length (number of past sam-ples to use in the average), P represents the correlation esti-2 5 mate generated for the current symbol, and v* is the complexconjugate of v. This conjugation produces a subtraction in terms of the phase angle between two complex numbers.
A channel that has only Rayleigh fading has a Cd(t) that is close to unity and a ~e(t) that is lower than the Cd(t) even 3 0 though both vd(t) and ve(t) have low distortion. This is due to an inherent sensitivity loss in the equalizer when a Rayleigh channel is encountered. A channel that has delay spread dis-tortion has a distorted vd(t) and a ve(t) with very little distortion due to the equalizer. This results in a larger Ce(t) than ~d(t)~
showing a high degree of correlation between the valid 5 DEQPSK constellation and the signal from the equalizer. The detector selection algorithm (104), therefore, will select the vd(t) output when a Rayleigh ch~nnel is encountered and the ve(t) output when a delay spread ~h~nnel is encountered.
The selection process is illustrated in Fig. 3. This pro-1 0 cess uses a threshold, a typical value being 0.975, that is ameans by which the switch can be biased towards the coherent detector, without the equalizer, in flat fading. Although the equalizer has a sensilivily loss associated with it in flat fading, it still performs adequately to occasionally fool the switch into 1 5 thinking that it is pelr."mi~lg equally to the coherent detector.
By biasing the switch toward the coherent detector, the performance of the switch converges towards the coherent detector's performance curve in flat fading while its delay spread performance rem~ins unchanged. The reason for this 2 0 is that the coherent detector is always working below the threshold in the delay spread environment. The correlation of the detector is first tested (301). If this correlation is less than or equal to the threshold (302), a ch~qnnel with delay spread distortion is indicated (303). If this correlation is greater than 2 5 the threshold, a normal Rayleigh ~h~nnel exists and data is taken from the detector without the equalizer (304). If a ~h~nnçl with delay spread distortion is present, a test is done to determine if the distortion is enough to warrant taking the data from the equalizer (305). If the correlation for the 3 0 equalizer is greater than the correlation for the detector, data is taken from the equalizer (307). Otherwise, the distortion is so slight that the data is taken from the detector (304).
An alternate embodiment of the present invention is il-lustrated in Fig. 4. This embodiment uses three QPSK detec-3 5 tors (402 - 404) controlled by the channel selection algorithm 20~9~DI
The decision on whether to use the limited output of the coherent detector is done in the following m~nner. The output 5 of the coherent detector without the equalizer (102) is defined as vt(t) and the detector with the equalizer (103) is defined as ve(t). The detected output of the the first detector is therefore represented by vd(t) and the latter by ve(t). Now let 1 0 Cd(t) = E(Vd(t)~d(t)) Ce(t) = E~ve(t)~e(t)}
where Ct and Ce are the correlation functions between the de-tected DEQPSK constellation points and the actual received 1 5 signal and E{ } denotes the en~emble average. The above true correlations can be estimated from a finite length data se-quence:
No d(t) N p ~ vd(n)V*d(n-t) 0 n=P+l No Ce (t) N - P ~ Ve (n)ve*(n-t) n=P+ 1 where No represents the window length (number of past sam-ples to use in the average), P represents the correlation esti-2 5 mate generated for the current symbol, and v* is the complexconjugate of v. This conjugation produces a subtraction in terms of the phase angle between two complex numbers.
A channel that has only Rayleigh fading has a Cd(t) that is close to unity and a ~e(t) that is lower than the Cd(t) even 3 0 though both vd(t) and ve(t) have low distortion. This is due to an inherent sensitivity loss in the equalizer when a Rayleigh channel is encountered. A channel that has delay spread dis-tortion has a distorted vd(t) and a ve(t) with very little distortion due to the equalizer. This results in a larger Ce(t) than ~d(t)~
showing a high degree of correlation between the valid 5 DEQPSK constellation and the signal from the equalizer. The detector selection algorithm (104), therefore, will select the vd(t) output when a Rayleigh ch~nnel is encountered and the ve(t) output when a delay spread ~h~nnel is encountered.
The selection process is illustrated in Fig. 3. This pro-1 0 cess uses a threshold, a typical value being 0.975, that is ameans by which the switch can be biased towards the coherent detector, without the equalizer, in flat fading. Although the equalizer has a sensilivily loss associated with it in flat fading, it still performs adequately to occasionally fool the switch into 1 5 thinking that it is pelr."mi~lg equally to the coherent detector.
By biasing the switch toward the coherent detector, the performance of the switch converges towards the coherent detector's performance curve in flat fading while its delay spread performance rem~ins unchanged. The reason for this 2 0 is that the coherent detector is always working below the threshold in the delay spread environment. The correlation of the detector is first tested (301). If this correlation is less than or equal to the threshold (302), a ch~qnnel with delay spread distortion is indicated (303). If this correlation is greater than 2 5 the threshold, a normal Rayleigh ~h~nnel exists and data is taken from the detector without the equalizer (304). If a ~h~nnçl with delay spread distortion is present, a test is done to determine if the distortion is enough to warrant taking the data from the equalizer (305). If the correlation for the 3 0 equalizer is greater than the correlation for the detector, data is taken from the equalizer (307). Otherwise, the distortion is so slight that the data is taken from the detector (304).
An alternate embodiment of the present invention is il-lustrated in Fig. 4. This embodiment uses three QPSK detec-3 5 tors (402 - 404) controlled by the channel selection algorithm 20~9~DI
(401). The ch~nnel detection algorithm (401) is illustrated in Fig. 5. The first detector (402) of the alternate embo-liment is a DFE. The second detector (403) is a closed loop coherent detec-tor such as a digital decision directed phase lock loop (PLL). A
PLL is a common closed loop coherent detector. PLLs are de-scribed in LEE AND MESSERSCHMIT, DIGITAL
COMMUNICATIONS, Chapters 13 and 14 (1988). The third de-tector (404) is an inst~nt~neous carrier estimate detector.
This detector uses the phase difference between its output at t 1 0 = (n-1) and the symbol decision at t = (n-1) as the demodulation phase at t = n.
As in the preferred embo-liment the channel selection algorithm in the alternate embo-liment selects the a~rol.l;ate detector based on the measured ch~nn~l conditions. A chan-1 5 nel sonn~ling (501), that is performing a correlation to obtain an estimate of the symbol-rate sampled ~h~nnel impulse re-sponse, {h(n)~, is done when a sync sequence (201 and 202) is received. The elements of ~h(n)~, h(0), h(-1), h(1), etc., indicate whether delay spread distortion is present and therefore 2 0 whether eqll~ tion is necessary. An initial selection (502), between detectors (402) and (403), is next done based on if Ih(n)l < k h(0)l ~ n ~ o where k is a small positive constant that allows tolerances of small inaccuracies in the estimate of {h(n)~. These inaccuracies come mainly from the noise in the 2 5 rece*er. If no noise is present in the received si n~l, k = 0.
To allow for noise in the estimate of {h(n)~, k is set to a small positive constant such as 0.05 or 0.10. The exact value chosen for k depends on the aspects of the specific system, such as sync sequence length and expected range of signal to noise ra-tios.
If the above expression is true, detector (403) is selected, otherwise detector (402) is selected. Detector (404) is not con-sidered for initial selection because detector (403) performs bet-ter in the majority of non-delay spread distortion l h~nnel~
- 3 5 and detector (402) is chosen for delay spread channels.
PLL is a common closed loop coherent detector. PLLs are de-scribed in LEE AND MESSERSCHMIT, DIGITAL
COMMUNICATIONS, Chapters 13 and 14 (1988). The third de-tector (404) is an inst~nt~neous carrier estimate detector.
This detector uses the phase difference between its output at t 1 0 = (n-1) and the symbol decision at t = (n-1) as the demodulation phase at t = n.
As in the preferred embo-liment the channel selection algorithm in the alternate embo-liment selects the a~rol.l;ate detector based on the measured ch~nn~l conditions. A chan-1 5 nel sonn~ling (501), that is performing a correlation to obtain an estimate of the symbol-rate sampled ~h~nnel impulse re-sponse, {h(n)~, is done when a sync sequence (201 and 202) is received. The elements of ~h(n)~, h(0), h(-1), h(1), etc., indicate whether delay spread distortion is present and therefore 2 0 whether eqll~ tion is necessary. An initial selection (502), between detectors (402) and (403), is next done based on if Ih(n)l < k h(0)l ~ n ~ o where k is a small positive constant that allows tolerances of small inaccuracies in the estimate of {h(n)~. These inaccuracies come mainly from the noise in the 2 5 rece*er. If no noise is present in the received si n~l, k = 0.
To allow for noise in the estimate of {h(n)~, k is set to a small positive constant such as 0.05 or 0.10. The exact value chosen for k depends on the aspects of the specific system, such as sync sequence length and expected range of signal to noise ra-tios.
If the above expression is true, detector (403) is selected, otherwise detector (402) is selected. Detector (404) is not con-sidered for initial selection because detector (403) performs bet-ter in the majority of non-delay spread distortion l h~nnel~
- 3 5 and detector (402) is chosen for delay spread channels.
After the initial selection, each detector in the alternate embodiment i8 allowed to accumulate a QPSK constellation correlation history over nblock symbols (503). nblock is the (i) number of times that the Rn equation is updated (after the sync sequence) before a detector switch is allowed. Since Rn is reset to zero at the beginning of each new time slot, nblock samples must be accumulated to average out any noise or dis-tortion. In other words, Rn is an exponentially windowed av-erage constellation correlation and samples must be accumu-1 0 lated before the average can be valid. The value of nblock would typically be determined by how rapidly the channel is ~h~3n~ing, this could be in the range of 16 to 30. After nblock symbols, the best detector output is selected at every symbol time based on the magnitude of the constellation correlations.
The constell~tion correlation, R, i9 computed according to the recul~ive, exponentially weighted correlation equation:
RD = ~R( ) + ej~ e ~d 2 0 where: n = symbol number (n = 1 is the first symbol after sync sequence);
~ = memory coefficient, 0 < ~ < 1;
i = detector number (1 - 3 in this embo-liment);
~r = phase of the output of detector i at time n;
2 5 ~d = phase of the decision at time n.
The detector (402 - 404) having the largest const~ tion corre-lation at the current symbol time is selected (504) as an input to the decision device (405) that quantizes the signal to the 3 0 nearest valid QPSK conste11~tion point. The output of the deci-sion device (405) is fed back to the detectors (402 - 404) for up-~049501 dating the valid constellation points (505). The above describedprocess is repeated for each received sync sequence (201 and 202).
The two different correlation methods discussed above 5 may be used interchangeably between the preferred and alter-nate embodiments. The preferred embodiment correlation method is the rectangular window method while the alternate embodiment discussion teaches the exponentially decaying window method. The exponential window method tracks 10 channel variations by using an exponentially decaying, con-tinuous window. The rectangular window method tracks rh~nnel variations by dividing the time slot into smaller sub-blocks and uses a rectangular window for averaging over these sub-blocks.
1 5 Other alternate embo-liments of the present invention are based on different measurement criteria of the ch~nnel.
A first of these alternate embodiments is illustrated in Fig. 7.
This is a frequency spectrum method based on the identifica-tion of a spectral null with the band. The data modulation is 2 0 assumed to be r~ntlom noise that generates a flat spectra across the çh~nnel frequency band. In a Rayleigh fading rhz~nnel, the whole spectrum across the band is attenuated.
In a delay spread rh~nnel, however, certain regions of the spectrum are attenuated and will show a notch. The short 2 5 term spectra calculation is used to identify the channel condi-tions to activate the equalizer. The implementation of the fre-quency spectrum calculation can be based on a Fast Fourier Transform approach or a filter bank implementation.
Another alternate embodiment is illustrated in Fig. 8.
3 0 This embo-limçnt uses the sync word to identify the rh?,nnel.
The complex correlation of the received data with a sync de-fines the rh~nnel impulse response that can be used to iden-tify the ch~nnel~ This embodiment can be used to perform a single snapshot identification since the sync word is only pre-3 5 sent at the beginning of the TDM frame.
204950~
Still another alternate embodiment of the present inven-tion is illustrated in Fig. 9. This embodiment makes a selec-tion decision based on:
Cde(t) = E{vd(t)ve(t)}
where Cde is the cross-correlation function between the coher-ent detector and the equalizer. If the correlation is above some threshold, then both detectors are performing well (implying 1 0 flat Rayleigh fading) and the coherent detector should be cho-sen. Otherwise, the chAnnel is probably a delay spread çhAnnel and the equalizer should be chosen.
Still another alternate embodiment of the present inven-tion, illustrated in Fig. 10, uses the meA~ square error output 1 5 from both detectors to identify the ~hAnnel. The correlation approach is applied to identify signal quality. The detector with the highest quality is used for the data detection. This embodiment is sensitive to received RF signal level since the level will be reflected in the error signal. A form of normal-2 0 ization or automatic gain control can be used to reduce thelevel depen-iAnce.
The DFE equalizer coefficients can be used to identify the chAnnel in another embodiment of the present invention, illustrated in Fig. 11. This embodiment requires extensive 2 5 pattern mAt(-hing to define the ch~Annel type from a wide range of possihle coefficient values. RF levels will be reflected in the coefficients.
Another embo-limçnt of the present invention, illus-trated in Fig. 12, adjusts the memory coefficient of the equal-3 0 izer to dirrel el-t values in a Rayleigh and delay spread chan-nel. The memory coefficient in the equalizer structure deter-mines the trAçking performance of the eqllAli7er; a value closer to unity is desirable for operation in a Rayleigh chan-nel. A lower value, however, is needed for trA~king the dy-3 5 namic delay spread channel.
20495 ûl Fig. 13 illustrates another alternate embodiment of the present invention based on channel identification directly.
One possible method of identification is to implement a chan-nel estimation based on the LMS or any other type of adaptive 5 algorithm to identify the çh~nnel Fig. 6 shows the present invention as used in a typical application, such as a radiotelephone. The signal is received by the ~ntenn~ (601) and is mixed to baseband by the local oscillator (602) and filtered by the rece*e filter ~603). This 1 0 signal is then sampled and converted to a digital stream of data by the Analog to Digital (A/D) converter (604). The coherent detectors and selection algorithm (100) of the present invention then processes the data as described above.
Although a QPSK detection algorithm is the preferred 1 5 embodiment of the present invention, the present invention will work with any modulation method.In sllmm~ry, the pre-sent invention enables adaptive switching between an equal-izer and a coherent detector or detectors to recover data from a received signal. The switching is accomplished dynamically, 2 0 depentling on whether the receiver is in a delay spread distor-tion enviro~ment, as the signal is received.
The constell~tion correlation, R, i9 computed according to the recul~ive, exponentially weighted correlation equation:
RD = ~R( ) + ej~ e ~d 2 0 where: n = symbol number (n = 1 is the first symbol after sync sequence);
~ = memory coefficient, 0 < ~ < 1;
i = detector number (1 - 3 in this embo-liment);
~r = phase of the output of detector i at time n;
2 5 ~d = phase of the decision at time n.
The detector (402 - 404) having the largest const~ tion corre-lation at the current symbol time is selected (504) as an input to the decision device (405) that quantizes the signal to the 3 0 nearest valid QPSK conste11~tion point. The output of the deci-sion device (405) is fed back to the detectors (402 - 404) for up-~049501 dating the valid constellation points (505). The above describedprocess is repeated for each received sync sequence (201 and 202).
The two different correlation methods discussed above 5 may be used interchangeably between the preferred and alter-nate embodiments. The preferred embodiment correlation method is the rectangular window method while the alternate embodiment discussion teaches the exponentially decaying window method. The exponential window method tracks 10 channel variations by using an exponentially decaying, con-tinuous window. The rectangular window method tracks rh~nnel variations by dividing the time slot into smaller sub-blocks and uses a rectangular window for averaging over these sub-blocks.
1 5 Other alternate embo-liments of the present invention are based on different measurement criteria of the ch~nnel.
A first of these alternate embodiments is illustrated in Fig. 7.
This is a frequency spectrum method based on the identifica-tion of a spectral null with the band. The data modulation is 2 0 assumed to be r~ntlom noise that generates a flat spectra across the çh~nnel frequency band. In a Rayleigh fading rhz~nnel, the whole spectrum across the band is attenuated.
In a delay spread rh~nnel, however, certain regions of the spectrum are attenuated and will show a notch. The short 2 5 term spectra calculation is used to identify the channel condi-tions to activate the equalizer. The implementation of the fre-quency spectrum calculation can be based on a Fast Fourier Transform approach or a filter bank implementation.
Another alternate embodiment is illustrated in Fig. 8.
3 0 This embo-limçnt uses the sync word to identify the rh?,nnel.
The complex correlation of the received data with a sync de-fines the rh~nnel impulse response that can be used to iden-tify the ch~nnel~ This embodiment can be used to perform a single snapshot identification since the sync word is only pre-3 5 sent at the beginning of the TDM frame.
204950~
Still another alternate embodiment of the present inven-tion is illustrated in Fig. 9. This embodiment makes a selec-tion decision based on:
Cde(t) = E{vd(t)ve(t)}
where Cde is the cross-correlation function between the coher-ent detector and the equalizer. If the correlation is above some threshold, then both detectors are performing well (implying 1 0 flat Rayleigh fading) and the coherent detector should be cho-sen. Otherwise, the chAnnel is probably a delay spread çhAnnel and the equalizer should be chosen.
Still another alternate embodiment of the present inven-tion, illustrated in Fig. 10, uses the meA~ square error output 1 5 from both detectors to identify the ~hAnnel. The correlation approach is applied to identify signal quality. The detector with the highest quality is used for the data detection. This embodiment is sensitive to received RF signal level since the level will be reflected in the error signal. A form of normal-2 0 ization or automatic gain control can be used to reduce thelevel depen-iAnce.
The DFE equalizer coefficients can be used to identify the chAnnel in another embodiment of the present invention, illustrated in Fig. 11. This embodiment requires extensive 2 5 pattern mAt(-hing to define the ch~Annel type from a wide range of possihle coefficient values. RF levels will be reflected in the coefficients.
Another embo-limçnt of the present invention, illus-trated in Fig. 12, adjusts the memory coefficient of the equal-3 0 izer to dirrel el-t values in a Rayleigh and delay spread chan-nel. The memory coefficient in the equalizer structure deter-mines the trAçking performance of the eqllAli7er; a value closer to unity is desirable for operation in a Rayleigh chan-nel. A lower value, however, is needed for trA~king the dy-3 5 namic delay spread channel.
20495 ûl Fig. 13 illustrates another alternate embodiment of the present invention based on channel identification directly.
One possible method of identification is to implement a chan-nel estimation based on the LMS or any other type of adaptive 5 algorithm to identify the çh~nnel Fig. 6 shows the present invention as used in a typical application, such as a radiotelephone. The signal is received by the ~ntenn~ (601) and is mixed to baseband by the local oscillator (602) and filtered by the rece*e filter ~603). This 1 0 signal is then sampled and converted to a digital stream of data by the Analog to Digital (A/D) converter (604). The coherent detectors and selection algorithm (100) of the present invention then processes the data as described above.
Although a QPSK detection algorithm is the preferred 1 5 embodiment of the present invention, the present invention will work with any modulation method.In sllmm~ry, the pre-sent invention enables adaptive switching between an equal-izer and a coherent detector or detectors to recover data from a received signal. The switching is accomplished dynamically, 2 0 depentling on whether the receiver is in a delay spread distor-tion enviro~ment, as the signal is received.
Claims (10)
1. A method for adaptive detection/equalization of radio frequency communication channels, over which a plurality of symbols are transmitted, in a communication device having a detection algorithm without an equalizer and a detection algorithm with an equalizer, the detection algorithms generating data for use by the communication device, the method comprising the steps of:
a) measuring a first correlation history for the detection algorithm without the equalizer;
b) measuring a second correlation history for the detection algorithm with the equalizer;
c) if the first correlation history is greater than a predetermined threshold, using the data from the detection algorithm without the equalizer;
d) if the first correlation history is less than or equal to the predetermined threshold and less than the second correlation history, using the data from the detection algorithm with the equalizer; and e) if the first correlation history is less than or equal to the predetermined threshold and greater than or equal to the second correlation history, using the data from the detection algorithm without the equalizer.
a) measuring a first correlation history for the detection algorithm without the equalizer;
b) measuring a second correlation history for the detection algorithm with the equalizer;
c) if the first correlation history is greater than a predetermined threshold, using the data from the detection algorithm without the equalizer;
d) if the first correlation history is less than or equal to the predetermined threshold and less than the second correlation history, using the data from the detection algorithm with the equalizer; and e) if the first correlation history is less than or equal to the predetermined threshold and greater than or equal to the second correlation history, using the data from the detection algorithm without the equalizer.
2. A method for adaptive detection/equalization of radio fre-quency communication channels, over which a plurality of symbols are transmitted, in a communication device having a plurality of detection algorithms, the method comprising the steps of:
a) estimating a symbol-rate sampled channel impulse response, {h(n)};
b) selecting a first detection algorithm in response to the values of {h(n)};
c) building a correlation history, for a predetermined number of symbols, for each of the plurality of detection algo-rithms;
d) selecting a second detection algorithm in response to the value of the correlation history;
e) updating the correlation history; and f) repeating from step d.
a) estimating a symbol-rate sampled channel impulse response, {h(n)};
b) selecting a first detection algorithm in response to the values of {h(n)};
c) building a correlation history, for a predetermined number of symbols, for each of the plurality of detection algo-rithms;
d) selecting a second detection algorithm in response to the value of the correlation history;
e) updating the correlation history; and f) repeating from step d.
3. A method for adaptive detection/equalization of radio frequency communication channels, over which a plurality of symbols are transmitted, in a communication device having a plurality of detection algorithms without an equalizer and a detection algorithm with an equalizer, the detection algorithms generating data for use by the communication device, the method comprising the steps of:
a) measuring a correlation history for each of the detection algorithms without the equalizer;
b) measuring a correlation history for the detection algorithm with the equalizer;
c) if each of the correlations for the detection algorithms without the equalizer is greater than a pre-determined threshold, using the data from a first detection algorithm without the equalizer having the largest correlation history;
d) if each of the correlations for the detection algorithms without the equalizer is less than or equal to the predetermined threshold and less than the correlation history for the detection algorithm with the equalizer, using the data from the detection algorithm with the equalizer; and e) if each of the correlations for the detection algorithms without the equalizer is less than or equal to the predetermined threshold and greater than or equal to the correlation history for the detection algorithm with the equalizer, using the data from the detection algorithm without the equalizer.
a) measuring a correlation history for each of the detection algorithms without the equalizer;
b) measuring a correlation history for the detection algorithm with the equalizer;
c) if each of the correlations for the detection algorithms without the equalizer is greater than a pre-determined threshold, using the data from a first detection algorithm without the equalizer having the largest correlation history;
d) if each of the correlations for the detection algorithms without the equalizer is less than or equal to the predetermined threshold and less than the correlation history for the detection algorithm with the equalizer, using the data from the detection algorithm with the equalizer; and e) if each of the correlations for the detection algorithms without the equalizer is less than or equal to the predetermined threshold and greater than or equal to the correlation history for the detection algorithm with the equalizer, using the data from the detection algorithm without the equalizer.
4. A method for adaptive detection/equalization of radio fre-quency communication channels, over which a plurality of symbols are transmitted, in a communication device having a plurality of detection algorithms without an equalizer and a detection algorithm with an equalizer, the detection algo-rithms generating data for use by the communication device, the method comprising the steps of:
a) measuring a power spectrum in a radio frequency communication channel;
b) if the power spectrum is uniform, using the data from the detection algorithm without the equalizer; and c) if the power spectrum is not uniform, using the data from the detection algorithm with the equalizer.
a) measuring a power spectrum in a radio frequency communication channel;
b) if the power spectrum is uniform, using the data from the detection algorithm without the equalizer; and c) if the power spectrum is not uniform, using the data from the detection algorithm with the equalizer.
5. A method for adaptive detection/equalization of radio fre-quency communication channels, over which a plurality of symbols are transmitted, in a communication device having a plurality of detection algorithms without an equalizer and a detection algorithm with an equalizer, the detection algo-rithms generating data for use by the communication device, the method comprising the steps of:
a) measuring a power spectrum in a plurality of filter-ing means;
b) comparing a first filtering means with a second fil-tering means to determine uniformity of the power spectrum among the plurality of filtering means;
c) if the power spectrum is uniform, using the data from the detection algorithm without the equalizer; and d) if the power spectrum is not uniform, using the data from the detection algorithm with the equalizer.
a) measuring a power spectrum in a plurality of filter-ing means;
b) comparing a first filtering means with a second fil-tering means to determine uniformity of the power spectrum among the plurality of filtering means;
c) if the power spectrum is uniform, using the data from the detection algorithm without the equalizer; and d) if the power spectrum is not uniform, using the data from the detection algorithm with the equalizer.
6. A method for adaptive detection/equalization of radio fre-quency communication channels, over which a plurality of symbols are transmitted, in a communication device, the method comprising the steps of:
a) estimating a symbol-rate sampled channel impulse response, {h(n)};
b) selecting a memory coefficient in response to the values of {h(n)}; and c) using the memory coefficient to vary tracking of an equalizer algorithm.
a) estimating a symbol-rate sampled channel impulse response, {h(n)};
b) selecting a memory coefficient in response to the values of {h(n)}; and c) using the memory coefficient to vary tracking of an equalizer algorithm.
7. A method for adaptive detection/equalization of radio frequency communication channels, over which a plurality of symbols are transmitted, in a communication device having a plurality of detection algorithms without an equalizer and a detection algorithm with an equalizer, the detection algorithms generating data for use by the communication device, the method comprising the steps of:
a) measuring a power spectrum in a plurality of filtering means;
b) comparing a first filtering means with a second filtering means to determine uniformity of the power spectrum among the plurality of filtering means;
c) if the power spectrum is uniform, using the data from the detection algorithm without the equalizer; and d) if the power spectrum is not uniform, using the data from the detection algorithm with the equalizer.
a) measuring a power spectrum in a plurality of filtering means;
b) comparing a first filtering means with a second filtering means to determine uniformity of the power spectrum among the plurality of filtering means;
c) if the power spectrum is uniform, using the data from the detection algorithm without the equalizer; and d) if the power spectrum is not uniform, using the data from the detection algorithm with the equalizer.
8. A method for adaptive detection/equalization of radio frequency communication channels, over which a plurality of symbols are transmitted, in a communication device having at least one equalizer that varies performance in response to a memory coefficient, the method comprising the steps of:
a) performing a correlation to estimate a symbol-rate sampled channel impulse response, {h(n)};
b) selecting a memory coefficient in response to the values of {h(n)}; and c) using the selected memory coefficient to vary tracking of an equalizer algorithm to equalize the radio frequency communication channels.
a) performing a correlation to estimate a symbol-rate sampled channel impulse response, {h(n)};
b) selecting a memory coefficient in response to the values of {h(n)}; and c) using the selected memory coefficient to vary tracking of an equalizer algorithm to equalize the radio frequency communication channels.
9. A method for adaptive detection/equalization of radio frequency communication channels, over which a plurality of symbols are transmitted, in a communication device having a detection algorithm without an equalizer and a detection algorithm with an equalizer, the detection algorithms generating data for use by the communication device, the method comprising the steps of:
a) measuring a first correlation history for the detection algorithm without the equalizer;
b) measuring a second correlation history for the detection algorithm with the equalizer;
c) if the first correlation history is greater than or equal to the second correlation, using the data from the detection algorithm without the equalizer; and d) if the first correlation history is less than the second correlation history, using the data form the detection algorithm with the equalizer.
a) measuring a first correlation history for the detection algorithm without the equalizer;
b) measuring a second correlation history for the detection algorithm with the equalizer;
c) if the first correlation history is greater than or equal to the second correlation, using the data from the detection algorithm without the equalizer; and d) if the first correlation history is less than the second correlation history, using the data form the detection algorithm with the equalizer.
10. A method for adaptive detection/equalization of radio fre-quency communication channels, over which a plurality of symbols are transmitted, in a communication device having an equalization algorithm with an error gradient, the method comprising the steps of:
a) estimating the equalizer error gradient;
b) selecting a memory coefficient in response to the value of the equalizer error gradient; and c) using the memory coefficient to vary tracking of the equalization algorithm.
a) estimating the equalizer error gradient;
b) selecting a memory coefficient in response to the value of the equalizer error gradient; and c) using the memory coefficient to vary tracking of the equalization algorithm.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US07/612,656 US5195106A (en) | 1990-11-14 | 1990-11-14 | Method for channel adaptive detecting/equalize |
| US612,656 | 1990-11-14 |
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| CA2049501A1 CA2049501A1 (en) | 1992-05-15 |
| CA2049501C true CA2049501C (en) | 1995-07-04 |
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| CA002049501A Expired - Fee Related CA2049501C (en) | 1990-11-14 | 1991-08-19 | Method for channel adaptive detecting/equalizing |
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| US (1) | US5195106A (en) |
| CA (1) | CA2049501C (en) |
| GB (1) | GB2251162B (en) |
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| JP3168610B2 (en) * | 1991-06-27 | 2001-05-21 | 日本電気株式会社 | Receiving machine |
| WO1993018593A1 (en) * | 1992-03-02 | 1993-09-16 | Motorola Inc. | Clock recovery method and apparatus in a diversity receiver |
| US5677934A (en) * | 1992-12-30 | 1997-10-14 | Nokia Mobile Phones Limited | Multipath propagation compensation in a TDMA system |
| FR2706100B1 (en) * | 1993-03-03 | 1995-07-21 | Alcatel Mobile Comm France | Method for equalizing a block of data in reception in a time division multiple access communications system and receiver implementing this method. |
| FR2718589B1 (en) * | 1994-04-11 | 1996-05-31 | Alcatel Mobile Comm France | Processing device in reception, in particular for digital radiocommunication system with mobiles. |
| US5659576A (en) * | 1994-05-19 | 1997-08-19 | Hughes Electronics | Balanced processing based on receiver selection |
| US6028901A (en) * | 1994-05-19 | 2000-02-22 | Hughes Electronics Corporation | Receiver selection based on delay spread estimation |
| US5572262A (en) * | 1994-12-29 | 1996-11-05 | Philips Electronics North America Corporation | Receiver based methods and devices for combating co-channel NTSC interference in digital transmission |
| US6075816A (en) * | 1996-11-27 | 2000-06-13 | Lucent Technologies, Inc. | Windowing technique for blind equalization |
| US5802114A (en) * | 1997-06-04 | 1998-09-01 | Ericsson Inc. | Method and apparatus for modulation differentiation |
| US6226336B1 (en) * | 1998-02-20 | 2001-05-01 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus for detecting a frequency synchronization signal |
| US6049574A (en) * | 1998-04-17 | 2000-04-11 | Trustees Of Tufts College | Blind adaptive equalization using cost function that measures dissimilarity between the probability distributions of source and equalized signals |
| US6097767A (en) * | 1998-04-17 | 2000-08-01 | Advanced Micro Devices, Inc. | Apparatus and method for determining an optimum equalizer setting for a signal equalizer in a communication network receiver |
| US6704317B1 (en) | 1998-05-27 | 2004-03-09 | 3Com Corporation | Multi-carrier LAN modem server |
| US6891887B1 (en) | 1998-05-27 | 2005-05-10 | 3Com Corporation | Multi-carrier LAN adapter device using interpolative equalizer |
| US6507585B1 (en) | 1998-05-27 | 2003-01-14 | 3Com Corporation | Multi-carrier LAN adapter device using frequency domain equalizer |
| US6603811B1 (en) | 1998-05-29 | 2003-08-05 | 3Com Corporation | Low complexity frequency domain equalizer having fast re-lock |
| US6377683B1 (en) | 1998-05-29 | 2002-04-23 | 3Com Corporation | Low complexity frequency domain echo canceller for DMT transceivers |
| US6643271B1 (en) | 1999-04-30 | 2003-11-04 | 3Com Corporation | Adjustable gain transmit cancellation in a full-duplex modem data access arrangement (DAA) |
| US6751202B1 (en) | 1999-04-30 | 2004-06-15 | 3Com Corporation | Filtered transmit cancellation in a full-duplex modem data access arrangement (DAA) |
| US6487259B1 (en) | 1999-08-24 | 2002-11-26 | Motorola, Inc. | Partially-parrallel trellis decoder apparatus and method |
| US6480555B1 (en) * | 1999-09-02 | 2002-11-12 | Motorola, Inc. | Method and apparatus for extended range frequency correction burst detection |
| US20040012803A1 (en) * | 2002-07-22 | 2004-01-22 | Umax Data Systems Inc. | System and method of a scanning moudle incorporated with a printer head |
| CN1977505A (en) * | 2004-05-12 | 2007-06-06 | 汤姆森许可公司 | Constellation location dependent step sizes for equalizer error signals |
| WO2005114932A1 (en) * | 2004-05-12 | 2005-12-01 | Thomson Licensing | Noise power estimate based equalizer lock detector |
| US8116254B2 (en) * | 2008-01-31 | 2012-02-14 | Powerwave Technologies, Inc. | Wireless repeater with smart uplink |
| US8326081B1 (en) * | 2009-05-18 | 2012-12-04 | The United States Of America As Represented By The Secretary Of The Navy | Correlation image detector |
| KR101574191B1 (en) * | 2010-01-07 | 2015-12-03 | 삼성전자주식회사 | Apparatus and method for estimating channel in broadband wireless communication system |
| EP2445120B1 (en) | 2010-10-25 | 2016-09-14 | Telefonaktiebolaget LM Ericsson (publ) | Wireless communications system and method |
| US11552831B1 (en) * | 2021-04-01 | 2023-01-10 | Cable Television Laboratories, Inc. | Systems and methods for equalizer correction |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4021738A (en) * | 1976-03-01 | 1977-05-03 | Bell Telephone Laboratories, Incorporated | Adaptive equalizer with fast convergence properties |
| US4805191A (en) * | 1987-11-25 | 1989-02-14 | Motorola, Inc. | Modem with improved timing recovery using equalized data |
| US5065410A (en) * | 1987-12-15 | 1991-11-12 | Nec Corporation | Method and arrangement for setting an amplitude equalization characteristic on an equalizer for use in a modem |
| JP2947752B2 (en) * | 1996-06-21 | 1999-09-13 | 弘高 春田 | Retractable stage with proscenium |
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1990
- 1990-11-14 US US07/612,656 patent/US5195106A/en not_active Expired - Fee Related
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- 1991-08-19 CA CA002049501A patent/CA2049501C/en not_active Expired - Fee Related
- 1991-11-12 GB GB9124020A patent/GB2251162B/en not_active Expired - Lifetime
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| GB9124020D0 (en) | 1992-01-02 |
| GB2251162B (en) | 1995-06-21 |
| US5195106A (en) | 1993-03-16 |
| GB2251162A (en) | 1992-06-24 |
| CA2049501A1 (en) | 1992-05-15 |
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