CA2504438A1 - High-voltage power supply - Google Patents
High-voltage power supply Download PDFInfo
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- CA2504438A1 CA2504438A1 CA002504438A CA2504438A CA2504438A1 CA 2504438 A1 CA2504438 A1 CA 2504438A1 CA 002504438 A CA002504438 A CA 002504438A CA 2504438 A CA2504438 A CA 2504438A CA 2504438 A1 CA2504438 A1 CA 2504438A1
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- voltage
- power supply
- voltage power
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Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M7/25—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only arranged for operation in series, e.g. for multiplication of voltage
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/003—Constructional details, e.g. physical layout, assembly, wiring or busbar connections
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
- H02M7/10—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode arranged for operation in series, e.g. for multiplication of voltage
- H02M7/103—Containing passive elements (capacitively coupled) which are ordered in cascade on one source
- H02M7/106—With physical arrangement details
Abstract
A high-voltage power supply (10) includes: a power scaling section (130) tha t receives an input voltage signal and converts the input voltage signal to a controllable DC voltage; a push-pull converter (140) for converting the controllable DC voltage to a high-frequency wave; and a voltage multiplier (200) receiving the high-frequency wave generated by the push-pull converter (140) and performing successive voltage doubling operations to generate a hi gh- voltage DC output. In one implementation, the voltage multiplier (200) receives a square wave having a frequency of approximately 100kHz and output s an adjustable DC voltage of approximately 0-to-30kV. In one implementation, the high-voltage power supply (10) includes an insulation system (250) for t he voltage multiplier module (200), such an insulation system being formed on n insulating layers and m conducting strips positioned between successive insulating layers.
Description
HIGH-VOLTAGE POWER SUPPLY
RELATED APPLICATIONS
[0001] This application claims priority under 35 U.S.C. ~
1 19(e) of U.S. Provisional Application 60/422,845 filed on November 1, 2002, the entire contents of which are herein incorporated by reference.
FIELD OF THE INVENTION
RELATED APPLICATIONS
[0001] This application claims priority under 35 U.S.C. ~
1 19(e) of U.S. Provisional Application 60/422,845 filed on November 1, 2002, the entire contents of which are herein incorporated by reference.
FIELD OF THE INVENTION
[0002] This invention relates to power supply technology, and more particularly to a high-voltage power supply with improved kV/volume characteristics.
BACKGROUND OF THE INVENTION
BACKGROUND OF THE INVENTION
[0003] A typical voltage supply, e.g., for supplying DC power to a high-voltage load, utilizes power electronics including a transformer to provide voltage step-up and isolation between the input and output and a voltage multiplier stage of suitable configuration for generating the required high-voltage from the transformer output. Normally, high-voltage power supplies for aerospace applications are large in size, typically larger than 0.5 cubic ft., due to such factors as insulation requirements, operating frequency, the large number of voltage doubling stages in the voltage multiplier, which is required for the typical input to the voltage multiplier (e.g., 25 V), and the size of magnetic components (inductor and transformer) in the power electronics. Furthermore, increasing output voltage ripple and voltage drop often occurs under load due to operation characteristics of the successive voltage doubling stages.
SUMMARY OF THE INVENTION
SUMMARY OF THE INVENTION
[0004] One aspect of the present invention is directed to a high-voltage power supply. According to one aspect, the present invention is directed to a continuously variable, high-voltage power supply. A high-voltage power supply in accordance with an embodiment of the present invention comprises: a power scaling section receiving an input voltage signal and converting the input voltage signal to a controllable DC voltage; a push-pull converter for converting the controllable DC voltage to a high-frequency wave; and a voltage multiplier receiving the high-frequency wave generated by the push-pull converter and performing successive voltage doubling operations to generate a high-s voltage DC output. Another aspect of the present invention is directed to a method for providing high-voltage power. A method for providing high-voltage power in accordance with an embodiment of the present invention comprises: receiving an input voltage signal and converting the input voltage signal to a controllable DC voltage at a power scaling section;
converting the controllable DC voltage to a high-frequency wave using a push-pull converter; and receiving the high-frequency wave generated by the push-pull converter and performing successive voltage doubling operations to generate a high-voltage DC output.
BRIEF DESCRIPTION OF THE DRAWINGS
converting the controllable DC voltage to a high-frequency wave using a push-pull converter; and receiving the high-frequency wave generated by the push-pull converter and performing successive voltage doubling operations to generate a high-voltage DC output.
BRIEF DESCRIPTION OF THE DRAWINGS
[0005] A more complete understanding of the present invention will become apparent from the following description taken in conjunction with the accompanying drawings, wherein:
[0006] FIG. 1 is a functional block diagram of a high-voltage power supply system according to an embodiment of the present invention;
[0007] FIG. 2, is a functional block diagram of a power electronics module according to an embodiment of the present invention;
[0008] FIG. 3 is a schematic illustration of a power electronics module of the high-voltage power supply system according to an exemplary implementation of the present invention;
[0009] FIG. 4 is a schematic illustration of the voltage multiplier module of the high-voltage power supply system according to an exemplary implementation of the present invention;
[0010] FIG. 5 is a functional block diagram of a power electronics control module of the high-voltage power supply according to an embodiment of the present invention;
[0011] FIG. 6 is a schematic illustration of the power electronics control module of the high-voltage power supply system according to an exemplary implementation of the present invention;
[0012] FIGS. 7A-7E illustrate a circuit board interconnection configuration for a high-voltage power supply in accordance with an exemplary implementation of the present invention;
[0013] FIG. 8 is a cross-sectional view of a voltage multiplier insulation system according to an implementation of the present invention; and [0014] FIG. 9 is a schematic illustration of an arrangement of voltage-doubling stages in a voltage multiplier of the high-voltage power supply according to an exemplary implementation of the present invention.
DETAILED DESCRIPTION
DETAILED DESCRIPTION
[0015] Other aspects and advantages of the invention will become apparent upon reading the following Detailed Description, with reference to the accompanying drawings.
[0016] FIG. 1 illustrates a functional block diagram of the high-voltage power supply (HVPS) 10 in accordance with principles of the present invention. Generally, the HVPS 10 converts DC (Direct Current) or AC (Alternating Current) voltage to isolated, high-accuracy, continuously adjustable high DC voltage, e.g., in the order of tens of kilovolts (kV). The power supply 1 O configured in accordance with principles of the present invention may have a kV/Volume of as high as 0.6 kV/in3 and a kV/Weight of as high as 30 kV/Ib. The power supply may be naturally cooled (e.g., convection cooling) and operate in a wide range of environments (e.g., earth at sea level to outer space).
[0017] The HVPS 10 shown in FIG. 1 comprises three main modules: a power electronics module 100; a voltage multiplier module 200; and a power electronics control module 300. The power electronics module 100 converts the input AC or DC voltage to an adjustable DC
voltage and then converts the adjustable DC voltage to a high frequency wave, which is a square-wave in a preferred embodiment, of adjustable amplitude. In one implementation of the present invention, the high-frequency wave generated by the power electronics module 100 has a frequency greater than approximately 20 kHz. In one preferred embodiment of the present invention, the high-frequency wave generated by the power electronics module 100 has a frequency of approximately 100 kHz. The power electronics module 100 uses high-frequency operation to reduce the need for bulky and heavy reactive elements. In addition, the high-frequency operation increases the power density of the HVPS 10 and reduces output voltage ripple and voltage drop conditions under load for reasons discussed in detail below. A high frequency transformer of the power electronics module 100 provides isolation between the input and output.
voltage and then converts the adjustable DC voltage to a high frequency wave, which is a square-wave in a preferred embodiment, of adjustable amplitude. In one implementation of the present invention, the high-frequency wave generated by the power electronics module 100 has a frequency greater than approximately 20 kHz. In one preferred embodiment of the present invention, the high-frequency wave generated by the power electronics module 100 has a frequency of approximately 100 kHz. The power electronics module 100 uses high-frequency operation to reduce the need for bulky and heavy reactive elements. In addition, the high-frequency operation increases the power density of the HVPS 10 and reduces output voltage ripple and voltage drop conditions under load for reasons discussed in detail below. A high frequency transformer of the power electronics module 100 provides isolation between the input and output.
[0018] The voltage multiplier module 200 converts the high frequency wave output by the power electronics module 100 to high-voltage. In one implementation of the present invention, the voltage multiplier module 200 is able to output an adjustable voltage of up to greater than approximately 1 kV DC. In a preferred embodiment of the present invention, the voltage multiplier module is able to output an adjustable voltage of up to approximately 30 kV DC. The upper range of the HVPS 10 output can be increased further by using a greater number of voltage doubling stages. An insulation system 250 is provided for with the voltage multiplier module 200. The insulation system 250 (discussed in detail below) allows high voltages in small volume. The control module 300 adjusts the amplitude of the wave output of the power electronics module 100 according to an analog or a digital command signal, in a closed-loop.
[0019] More specifically, the power electronics module 100 receives a power signal, e.g., 28-V, DC, and a variable command signal, e.g., 0-10 V, DC, that represents a continuously variable high-voltage power supply output voltage, e.g., in the range 0-to-30 kV DC. In the power electronics module 100, a so-called "buck-converter" is employed to convert the input voltage to a controllable DC voltage as a function of the desired output voltage. Then, a push-pull converter is used to convert this voltage to a high-frequency wave voltage, e.g., with a square wave having an amplitude of approximately 0-to-1 kV. A closed-loop control system, using a feedback signal between the power electronics module 100 and the power electronics control module 300, ensures that the output voltage of the HVPS 10 follows the command.
[0020] FIG. 2 is a functional block diagram of a power electronics module 100 according to an embodiment of the present invention. As seen in FIG. 2, the power electronics module 100 includes:
a power scaling section 130; a push-pull converter 140; and a power supply portion 120. The power scaling section 130 is a buck-converter, which receives an input power signal, e.g., 28-V, DC, and a scaling control signal representing a desired high-voltage power supply output voltage. The power scaling section 130 includes a switching element Q1, which is gated ON/OFF by a switch gating signal received from the power electronics control module 300 to convert the input power signal to a controllable DC voltage, e.g., in the range of 0-28V.
[0021 ] The push-pull converter 140 converts the output of the power scaling section 130 to a high-frequency wave voltage, e.g., with a peak value of 0-1 kV, using a high-frequency transformer configuration discussed in more detail below. The high-frequency wave generated by the push-pull converter 140 is a square-wave in accordance with an implementation of the present invention, thereby facilitating subsequent voltage multiplication and improving stability characteristics, including lower voltage drop and ripple levels, of the HVPS 10. The push-pull converter 140 provides isolation as well as voltage step-up between the input and output. The high-voltage output by the push-pull converter 140 allows the voltage multiplier module 200 to achieve the required range of output voltages with fewer voltage doubling stages. For example, if the maximum required output voltage of the HVPS 10 is 30 kV DC and the push-pull converter outputs 0-to-1 kV, the voltage multiplier may be implemented with 16 doubling stages. In this implementation, the extra voltage doubling stage is provided to compensate for voltage drop across the diodes at each voltage multiplier stage, to compensate for voltage drop under load, and to ensure that a 30 kV DC output voltage can be obtained over the entire range of the output load, which may cause losses in each stage of the voltage multiplier module. In one implementation of the present invention discussed below, the voltage multiplier stages are divided among two boards (e.g., 8 stages per board).
(0022] The power supply portion 120 receives the input power signal and generates supply voltages (e.g., 5V, 15V) for the various circuitry of the power electronics control module 300. The power supply portion 120 is an optional element, in that the supply voltages may be provided externally.
(0023] FIG. 3 illustrates schematic details of a power electronics module 100 according to an implementation of the present invention, including exemplary implementation details for the power scaling section 130, the push-pull converter 140, and the power supply portion 120. As seen in FIG. 3, the exemplary implementation of the power scaling section 130 is connected (via connections P, N) to receive the input power signal, which is DC power in the exemplary implementation of FIG. 3. The power scaling section 130 includes a switching element Q1, such as IRF 530 MOSFET, the duty cycle of which _7_ is controlled to scale the input voltage in accordance with a switch-gating signal G 1 P received from the power electronics control module 300. The power scaling section 130 shown in FIG. 3 includes resistors R1 and R2 for applying the gating signal G 1 P to the switching element Q1. The power scaling section 130 further includes a filtering arrangement, including inductor L1 and capacitors C7 and C8, for smoothing/filtering the scaled output voltage and an RC configuration, including resistors R4, R5, and capacitor C1 1, for generating a feedback signal VF representing the output of the power scaling section 130, which is fed back to the power electronics control module 300. The resistors R4 and R5_act as a voltage divider to produce the feedback signal VF and capacitor C11 filters the feedback signal. The power scaling section 130 further includes a configuration of capacitors C1, C3 for filtering the input voltage to reduce input voltage noise, and a diode D2, which is a freewheeling diode for the buck converter arrangement of FIG. 3.
[0024] The push-pull converter 140 includes a high-frequency transformer T1, e.g., having a turns ratio of 1:1:36, to provide voltage isolation and step-up between the input and output, and a pair of switches, Q2 and Q3, that are switched in accordance with switch gating signals G2P and G3P received from the power electronics control module 300. The switches Q2 and Q3 may be MOSFET type semiconductor switches, such as IRF 530 switched at a high rate, such as 100 kHz.
[0025] Thus, the switches Q2 and Q3, connected to the primary winding of the transformer T1, provide the transformer with a high-frequency voltage, which is stepped-up by the transformer T1 for output to the voltage multiplier module 200. The push-pull converter 140 further includes an arrangement of resistors R6, R7 for providing the switch gating signal to switch Q2 and resistors R10, R11 for providing the switch gating signal to switch Q3.
[0026] The power supply portion 120 according to the implementation of FIG. 3 includes a first voltage supply section 122 and a _g_ second voltage supply section 124. The first supply section 122 may be implemented as shown in FIG. 3 using a 15V voltage regulator U 1 (e.g., a 7815 integrated circuit). The schematic for the first supply section 122 in FIG. 3 includes capacitor C2 for filtering the input voltage, capacitors C4, C5 for filtering the output voltage of the 15V voltage regulator U 1, and a feedback diode D1 acting as a bypass diode. The second supply section 124 may be implemented using a 5V voltage regulator U2 (e.g., a 7805 integrated circuit). The schematic for the second supply section 124 in FIG. 3 includes capacitors C6 and C9 for input filtering and output filtering, and a bypass diode D4. The power supply module 120 for the exemplary implementation illustrated in FIG. 3 further includes a third supply section 126, which provides an isolated 15 V supply for the switching element Q1 of the power scaling section 130. The third supply section 126 includes an isolated DC to DC converter U3 (e.g., AA5-15 integrated circuit) and capacitors C10, C12, C13 for filtering.
[0027] FIG. 4 illustrates a circuit configuration for an embodiment of the voltage multiplier 200. The voltage multiplier includes a plurality of voltage multiplier stages 210-1, 210-2, ..., 210-n, each comprising a plurality of capacitors and diodes. As seen in FIG. 4, each voltage multiplier stage includes two branches, each with a pair of diodes.
Those skilled in the art will appreciate that the number of stages and the individual values of the components will control the voltage multiplication.
For example, in one configuration the capacitors can have a value of 0.1 micro Farad, and be configured in 16 stages (e.g., divided among two boards) to allow for multiplication to 30 kV DC when the maximum peak value of the wave input to the voltage multiplier module 200 is 1 kV.
Those skilled in the art will appreciate that other ratings of the elements and number of voltage doubling stages will need to be selected based upon the desired output voltage level.
[002] FIG. 5 is a functional block diagram of a power electronics control module 300 according to an embodiment of the _g_ present invention. As seen in FIG. 5, the power electronics control module 300 includes: a clock generator 310 for generating a high-frequency clock signal; a converter gating signal generator 320 for generating a switch gating signal for the switches Q2, Q3 of the push-pull converter 140 of the power electronics module 100; and a scaling section gating signal generator 330 for generating a switch gating signal for the switch Q1 of the power scaling section 130 of the power electronics module 100. The scaling section gating signal generator 330 controls the duty cycle of the switch Q1 of the power scaling section 130 in accordance with a command signal representing the desired output voltage and a feedback VF signal from the power scaling section 130 indicating the scaled voltage magnitude output to the push-pull converter 140.
[0029] FIG. 6 is a schematic illustration of the power electronics control module 300 according to an exemplary impler~nentation of the present invention. In FIG. 6, the clock generator 310 is implemented using a series of NOR gates U 1 A, U 1 B, U 1 C, and U 1 D, e.g., a series of two-input NOR gate CMOS integrated circuits (4001 ), generating gating clock signals Dr2 and Dr3. Gating clock signal Dr2, output by NOR gate U 1 C, is 1 SO degrees out of phase with gating clock signal Dr3, output by NOR gate U1 D, such that switching elements Q2, Q3 of the push-pull converter 140 are driven so that only one will be on at a time, alternately. The clock generator 310 further includes circuitry, including a capacitor C1, resistors R1, R2, R3, and diode D1 for producing the frequency of the pulses at Dr2, Dr3.
[0030] The converter gating signal generator 320 in FIG. 6 may be implemented using a high-speed power MOSFET driver, such as an IR 2125 integrated circuit. A first driver 322 generates a gate switching signal G2P based on clock signal Dr2 and a second driver circuit 324 generates a gate switching signal G3P in accordance with clock signal Dr3 received from the clock generator 310. The scaling section gating signal generator 330 in FIG. 6 includes a control circuit 334 that generates a scaling control signal Dr1 based on an input command voltage Vcmd and the feedback signal VF from the power scaling section 130. In the implementation illustrated in FIG. 6, the control circuit 334 includes a switching mode power supply controller integrated circuit (e.g., a TL 494 integrated circuit) to generate Dr1. The control circuit 334 includes external components, including various resistors and capacitors, that are recommended by the chip manufacturer and are used to define the switching frequency of the PWM operation. A high-speed MOSFET power driver 336 generates a switch gating signal G1 P for the switch Q1 of the power scaling section 130 based on switching control signal Dr1. The power driver 336 may be implemented using a well-known integrated circuit, such as an IR 2125 high-speed power MOSFET driver. The power electronics control module 330 illustrated in FIG. 6 further includes a control gain element 332, which includes an arrangement of resistors 88-R1 1 and a transistor Q1 to produce the control signal for the control circuit 334. The control gain element 332 receives an external command voltage, Cmd, and generates Vcmd, which in this implementation is scaled relative to Cmd. This scaling is performed to facilitate comparison between the command signal and the feedback signal VF as part of the closed-loop control of the output voltage of the power scaling section 130. Control gain element 332 also sets the dead time control function for control circuit 334 (Vdtc input).
[0031] FIGS. 3, 4, and 6 illustrate schematic details according to exemplary implementations of the invention. As will be appreciated by those skilled in the art, the specific elements and values shown can be interchanged with like functioning elements and appropriate combinations to achieve the same functional result (e.g., transistor types can be changed). Further, several elements or functions of groups of elements can be consolidated into single elements (e.g., Application Specific Integrated Circuits (ASICs), hybrid devices, and the like). Additionally, functional portions of the schematic can be supplied from other systems.
For example, the power supply portion 120 can be replaced by suitable external power sources. One skilled in the art will appreciate a variety of circuit configurations can be used to achieve similar functional results, as illustrated by the circuits shown.
[0032] FIGS. 7A-7E (collectively referred to as "FIG. 7") illustrate a circuit board interconnection configuration for a HVPS 10 in accordance with an exemplary implementation of the present invention. In this implementation, the circuitry associated with the power electronics control module 300 is provided on a first circuit board (board 1 ), and the circuitry associated with the power electronics module 100 is provided on a second board (board 2). The circuitry associated with the voltage multiplier module 200 is divided among two additional boards (board 3, board 4). The circuit board of the implementation illustrated in FIG. 7 utilizes a plurality of jumpers, J 1-J6, to provide connections between the various boards and external connections. In the circuit board configuration illustrated in FIG. 7, the control board (board 1 ) and the power electronics board (board 2) are provided below the voltage multiplier boards (boards 3, 4). The spacing between boards 1 and 2 may be considerably less than the spacing between boards 3 and 4, e.g., approximately 0.5 cm spacing as compared to approximately 2.0 cm spacing. The spacing between boards 2 and 3 should at least be as large as the spacing between boards 3 and 4.
[0033] FIG. 8 illustrates an insulation system 250 for providing improved insulation for the circuit boards of the voltage multiplier module 200 in accordance with an implementation of the present invention. The insulation system 250 provides increased breakdown voltage magnitude by controlling the electric field within a multilayer insulation system. The insulation system 250 is constructed from n layers of insulation 252-1, ..., 252-n and m very thin (e.g., less than approximately 1 mm) conducting strips 254-1, ..., 254-m placed between successive insulating layers. Each layer 252-1, ..., 252-n may be composed of a suitable dielectric material, such as a polymeric insulating material. The width of each conducting strip 254-1, ..., 254-m is represented by ICa, the thickness of each insulting later is represented by b, the width of the composite insulation system 250 is represented by A, and the thickness of the composite insulation system is represented by B.
In one implementation of the present invention the multiplier insulation composite is provided on both sides of the voltage multiplier boards (e.g., boards 3 and 4 is FIG. 7).
[0034] The conducting strips 254-1, ..., 254-m in each layer 252 are symmetrically shifted by a constant distance relative to the conductive strips in an adjacent layer, determined by a shifting factor which maximizes the withstand voltage of the insulation. The potential distribution within the composite insulation may be determined from the solution of the Laplace equation for different design parameters, to reduce the electric field. Principles underlying the insulation system 250 are described in detail in the article titled "Design of Field-Controlled Multi-layer Insulation System," IEEE Transactions on Electrical Insulation, Vol.
EI-21, No. 2, April 1986, which is herein incorporated by reference in its entirety. It should be recognized that parameters of the insulation system 250, such as n and m, will depend on the particular application and characteristics of the HVPS 10.
[0035] To improve heat dissipation and enable the position of a greater number of voltage multiplier capacitors in a given area, one implementation of the present invention arranges the capacitors 410 of cascaded voltage doubling stages in a zig-zag pattern relative.to the diodes 420, as illustrated in FIG. 9. By combining the insulation and thermal management techniques set forth herein, spacing for components and clearance between voltage multiplier boards can be reduced, thereby reducing the size of the HVPS 10.
[0036] Having provided a detailed description of embodiments and exemplary implementations of the present invention, characteristics and advantages of such embodiments and exemplary implementations will next be described. Normally, in the high-voltage power supply application, a sine-wave is used as the input to the voltage multiplier. Generating a sine-wave at high-frequency is not straight forward, whereas square-wave can be produced at high frequency without a problem. Furthermore, as mentioned above, a square-wave input to the voltage multiplier module 200 facilitates voltage multiplication and increases stability of the HVPS
under load. Thus, in a preferred embodiment, the HVPS 10 uses a high-frequency square-wave as the input to the voltage multiplier module 10 200.
[0037] Normally, high-voltage power supplies are large in size due to insulation requirements and the necessary number of voltage doubling stages. If it were attempted to use a high voltage square-wave as the input to voltage multiplier of a typical power supply, the high-voltage square-wave present within the stages of voltage multiplier would tend to cause flash-over (spark) between the component terminals which have high potential difference with respect to one another. Therefore, high-voltage power supplies that would attempt to use a high-voltage square-wave cannot be made small in size. The HVPS 10 of this invention uses high-frequency square-wave at high voltage levels in a small board size.
[0038] One advantage of using high-frequency operation is reduction of the magnetic components, i.e., isolation/step-up transformer in the push-pull converter and inductor in the buck converter. Another reason for using high-frequency operation is to reduce the output voltage ripple and voltage drop under load, as will be apparent from the following discussion.
[0039] The output voltage ripple is given by the following relation:
8V = I ~~~+1) __ q ( 1 + ~ +...+ ~ ) (equation 1 ) fC 4 2 Cl C2 C, where n is the number of voltage doubter stages in the voltage multiplier and q is the charge transferred from the capacitors to the load per period.
With Cl = C2 = ... = Cn , the output voltage ripple becomes:
~~ - q n(n + 1) - I n(h + 1) (equation 2) C 4 fC 4 As seen, as f is increased, ~T~ will decrease. This is one advantage of high-frequency operation.
[0040] The output voltage drop under load ( OVo ) due to current drawn from the capacitors and transferred to the load is given by:
OT~o = fC (2 3 +2 2 +...- 3) (equation 3) [0041] The above relation can be simplified for h)5 as follows:
OVo - ~C, 23 for n)5 (equation 4) or ~R'27~3 ~~O - C 3 for n)~5 (equation 5) [0042] As seen, as f is increased, OVo will decrease and as a result the maximum output voltage will increase. The maximum output voltage is given by the following equation:
~o,max =~(2~max.,noload)-~~o (equation 6) [0043] In the HVPS 10 of the present invention, the potential for flash-over caused by using high-frequency square-wave is addressed through implementation of a printed circuit board and insulation design.
For insulation, a field-controlled dielectric material utilizing the bulk of the insulating material rather than its surface may be used. The bulk is known to have a dielectric strength of at least 10 times that of the surface.
Therefore, the size of the board carrying the voltage multiplier can be dramatically reduced. With special design of the dielectric layers, the electric field is forced to penetrate into the material where the dielectric strength is high, instead of being set up on the surface.
[0044] The power electronics control module 300 has two main tasks. The first main task is to control the buck converter to produce scaled voltage from the source voltage based on a command signal, e.g., an analog command signal between 0 and 10 V, DC. A closed-loop control adjusts the magnitude of the buck converter output voltage by adjusting the duty cycle of the MOSFET switch to control the high-voltage output of the power supply between 0 and 30 kV based on the command signal. The second main task of the power electronics control module 300 is to produce the 'gating pulses for the push-pull converter MOSFET
switches. In one implementation, these switches are operated at 50%
duty ratio and 100 kHz switching frequency. In a preferred embodiment, analog control has been selected for the control of the buck converter to take advantage of its high-speed characteristics (e.g., achieving a stable output voltage within milliseconds of the input command). To achieve reference tracking at the switching rate of 100 kHz using digital control, a large and expensive microcontroller or DSP board is needed. Analog control guarantees very good results at a reasonable size and cost.
[0045] ~ The HVPS 10 of the present invention may be characterized by small size and weight (size: 5 cm x 10 cm x 15 cm, weight: < 500 grams). As a result of using an insulation system 250 in the voltage multiplier module 200, the size and clearance of the printed circuit boards housing the voltage doubter stages may be dramatically reduced. Existing high-voltage power supplies using voltage multipliers formed of capacitors and diodes are large in size, as the limited insulation strength and conventional insulation design does not allow small board size and clearances. Furthermore, due to operating at a high switching frequency, the sizes and weights of the magnetic components (buck converter inductor and push-pull converter transformer) have been dramatically reduced. A third factor contributing to the reduction in size and weight of the high-voltage power supply is the fact that the power supply may be naturally cooled and does not need any additional cooling mechanism (e.g., a fan) beside a heat sink used for the switches elements.
[0046] As discussed above, the HVPS can be installed in a variety of environments, for example, MIL-E-5400 requirement,,which defines environmental conditions under which the equipment must operate (including ambient temperature and pressure ranges that may be encountered by the equipment), or an unpressurized environment with ambient temperature up to 70 degree C. The operating altitude may vary from sea level to 16,000m.
a power scaling section 130; a push-pull converter 140; and a power supply portion 120. The power scaling section 130 is a buck-converter, which receives an input power signal, e.g., 28-V, DC, and a scaling control signal representing a desired high-voltage power supply output voltage. The power scaling section 130 includes a switching element Q1, which is gated ON/OFF by a switch gating signal received from the power electronics control module 300 to convert the input power signal to a controllable DC voltage, e.g., in the range of 0-28V.
[0021 ] The push-pull converter 140 converts the output of the power scaling section 130 to a high-frequency wave voltage, e.g., with a peak value of 0-1 kV, using a high-frequency transformer configuration discussed in more detail below. The high-frequency wave generated by the push-pull converter 140 is a square-wave in accordance with an implementation of the present invention, thereby facilitating subsequent voltage multiplication and improving stability characteristics, including lower voltage drop and ripple levels, of the HVPS 10. The push-pull converter 140 provides isolation as well as voltage step-up between the input and output. The high-voltage output by the push-pull converter 140 allows the voltage multiplier module 200 to achieve the required range of output voltages with fewer voltage doubling stages. For example, if the maximum required output voltage of the HVPS 10 is 30 kV DC and the push-pull converter outputs 0-to-1 kV, the voltage multiplier may be implemented with 16 doubling stages. In this implementation, the extra voltage doubling stage is provided to compensate for voltage drop across the diodes at each voltage multiplier stage, to compensate for voltage drop under load, and to ensure that a 30 kV DC output voltage can be obtained over the entire range of the output load, which may cause losses in each stage of the voltage multiplier module. In one implementation of the present invention discussed below, the voltage multiplier stages are divided among two boards (e.g., 8 stages per board).
(0022] The power supply portion 120 receives the input power signal and generates supply voltages (e.g., 5V, 15V) for the various circuitry of the power electronics control module 300. The power supply portion 120 is an optional element, in that the supply voltages may be provided externally.
(0023] FIG. 3 illustrates schematic details of a power electronics module 100 according to an implementation of the present invention, including exemplary implementation details for the power scaling section 130, the push-pull converter 140, and the power supply portion 120. As seen in FIG. 3, the exemplary implementation of the power scaling section 130 is connected (via connections P, N) to receive the input power signal, which is DC power in the exemplary implementation of FIG. 3. The power scaling section 130 includes a switching element Q1, such as IRF 530 MOSFET, the duty cycle of which _7_ is controlled to scale the input voltage in accordance with a switch-gating signal G 1 P received from the power electronics control module 300. The power scaling section 130 shown in FIG. 3 includes resistors R1 and R2 for applying the gating signal G 1 P to the switching element Q1. The power scaling section 130 further includes a filtering arrangement, including inductor L1 and capacitors C7 and C8, for smoothing/filtering the scaled output voltage and an RC configuration, including resistors R4, R5, and capacitor C1 1, for generating a feedback signal VF representing the output of the power scaling section 130, which is fed back to the power electronics control module 300. The resistors R4 and R5_act as a voltage divider to produce the feedback signal VF and capacitor C11 filters the feedback signal. The power scaling section 130 further includes a configuration of capacitors C1, C3 for filtering the input voltage to reduce input voltage noise, and a diode D2, which is a freewheeling diode for the buck converter arrangement of FIG. 3.
[0024] The push-pull converter 140 includes a high-frequency transformer T1, e.g., having a turns ratio of 1:1:36, to provide voltage isolation and step-up between the input and output, and a pair of switches, Q2 and Q3, that are switched in accordance with switch gating signals G2P and G3P received from the power electronics control module 300. The switches Q2 and Q3 may be MOSFET type semiconductor switches, such as IRF 530 switched at a high rate, such as 100 kHz.
[0025] Thus, the switches Q2 and Q3, connected to the primary winding of the transformer T1, provide the transformer with a high-frequency voltage, which is stepped-up by the transformer T1 for output to the voltage multiplier module 200. The push-pull converter 140 further includes an arrangement of resistors R6, R7 for providing the switch gating signal to switch Q2 and resistors R10, R11 for providing the switch gating signal to switch Q3.
[0026] The power supply portion 120 according to the implementation of FIG. 3 includes a first voltage supply section 122 and a _g_ second voltage supply section 124. The first supply section 122 may be implemented as shown in FIG. 3 using a 15V voltage regulator U 1 (e.g., a 7815 integrated circuit). The schematic for the first supply section 122 in FIG. 3 includes capacitor C2 for filtering the input voltage, capacitors C4, C5 for filtering the output voltage of the 15V voltage regulator U 1, and a feedback diode D1 acting as a bypass diode. The second supply section 124 may be implemented using a 5V voltage regulator U2 (e.g., a 7805 integrated circuit). The schematic for the second supply section 124 in FIG. 3 includes capacitors C6 and C9 for input filtering and output filtering, and a bypass diode D4. The power supply module 120 for the exemplary implementation illustrated in FIG. 3 further includes a third supply section 126, which provides an isolated 15 V supply for the switching element Q1 of the power scaling section 130. The third supply section 126 includes an isolated DC to DC converter U3 (e.g., AA5-15 integrated circuit) and capacitors C10, C12, C13 for filtering.
[0027] FIG. 4 illustrates a circuit configuration for an embodiment of the voltage multiplier 200. The voltage multiplier includes a plurality of voltage multiplier stages 210-1, 210-2, ..., 210-n, each comprising a plurality of capacitors and diodes. As seen in FIG. 4, each voltage multiplier stage includes two branches, each with a pair of diodes.
Those skilled in the art will appreciate that the number of stages and the individual values of the components will control the voltage multiplication.
For example, in one configuration the capacitors can have a value of 0.1 micro Farad, and be configured in 16 stages (e.g., divided among two boards) to allow for multiplication to 30 kV DC when the maximum peak value of the wave input to the voltage multiplier module 200 is 1 kV.
Those skilled in the art will appreciate that other ratings of the elements and number of voltage doubling stages will need to be selected based upon the desired output voltage level.
[002] FIG. 5 is a functional block diagram of a power electronics control module 300 according to an embodiment of the _g_ present invention. As seen in FIG. 5, the power electronics control module 300 includes: a clock generator 310 for generating a high-frequency clock signal; a converter gating signal generator 320 for generating a switch gating signal for the switches Q2, Q3 of the push-pull converter 140 of the power electronics module 100; and a scaling section gating signal generator 330 for generating a switch gating signal for the switch Q1 of the power scaling section 130 of the power electronics module 100. The scaling section gating signal generator 330 controls the duty cycle of the switch Q1 of the power scaling section 130 in accordance with a command signal representing the desired output voltage and a feedback VF signal from the power scaling section 130 indicating the scaled voltage magnitude output to the push-pull converter 140.
[0029] FIG. 6 is a schematic illustration of the power electronics control module 300 according to an exemplary impler~nentation of the present invention. In FIG. 6, the clock generator 310 is implemented using a series of NOR gates U 1 A, U 1 B, U 1 C, and U 1 D, e.g., a series of two-input NOR gate CMOS integrated circuits (4001 ), generating gating clock signals Dr2 and Dr3. Gating clock signal Dr2, output by NOR gate U 1 C, is 1 SO degrees out of phase with gating clock signal Dr3, output by NOR gate U1 D, such that switching elements Q2, Q3 of the push-pull converter 140 are driven so that only one will be on at a time, alternately. The clock generator 310 further includes circuitry, including a capacitor C1, resistors R1, R2, R3, and diode D1 for producing the frequency of the pulses at Dr2, Dr3.
[0030] The converter gating signal generator 320 in FIG. 6 may be implemented using a high-speed power MOSFET driver, such as an IR 2125 integrated circuit. A first driver 322 generates a gate switching signal G2P based on clock signal Dr2 and a second driver circuit 324 generates a gate switching signal G3P in accordance with clock signal Dr3 received from the clock generator 310. The scaling section gating signal generator 330 in FIG. 6 includes a control circuit 334 that generates a scaling control signal Dr1 based on an input command voltage Vcmd and the feedback signal VF from the power scaling section 130. In the implementation illustrated in FIG. 6, the control circuit 334 includes a switching mode power supply controller integrated circuit (e.g., a TL 494 integrated circuit) to generate Dr1. The control circuit 334 includes external components, including various resistors and capacitors, that are recommended by the chip manufacturer and are used to define the switching frequency of the PWM operation. A high-speed MOSFET power driver 336 generates a switch gating signal G1 P for the switch Q1 of the power scaling section 130 based on switching control signal Dr1. The power driver 336 may be implemented using a well-known integrated circuit, such as an IR 2125 high-speed power MOSFET driver. The power electronics control module 330 illustrated in FIG. 6 further includes a control gain element 332, which includes an arrangement of resistors 88-R1 1 and a transistor Q1 to produce the control signal for the control circuit 334. The control gain element 332 receives an external command voltage, Cmd, and generates Vcmd, which in this implementation is scaled relative to Cmd. This scaling is performed to facilitate comparison between the command signal and the feedback signal VF as part of the closed-loop control of the output voltage of the power scaling section 130. Control gain element 332 also sets the dead time control function for control circuit 334 (Vdtc input).
[0031] FIGS. 3, 4, and 6 illustrate schematic details according to exemplary implementations of the invention. As will be appreciated by those skilled in the art, the specific elements and values shown can be interchanged with like functioning elements and appropriate combinations to achieve the same functional result (e.g., transistor types can be changed). Further, several elements or functions of groups of elements can be consolidated into single elements (e.g., Application Specific Integrated Circuits (ASICs), hybrid devices, and the like). Additionally, functional portions of the schematic can be supplied from other systems.
For example, the power supply portion 120 can be replaced by suitable external power sources. One skilled in the art will appreciate a variety of circuit configurations can be used to achieve similar functional results, as illustrated by the circuits shown.
[0032] FIGS. 7A-7E (collectively referred to as "FIG. 7") illustrate a circuit board interconnection configuration for a HVPS 10 in accordance with an exemplary implementation of the present invention. In this implementation, the circuitry associated with the power electronics control module 300 is provided on a first circuit board (board 1 ), and the circuitry associated with the power electronics module 100 is provided on a second board (board 2). The circuitry associated with the voltage multiplier module 200 is divided among two additional boards (board 3, board 4). The circuit board of the implementation illustrated in FIG. 7 utilizes a plurality of jumpers, J 1-J6, to provide connections between the various boards and external connections. In the circuit board configuration illustrated in FIG. 7, the control board (board 1 ) and the power electronics board (board 2) are provided below the voltage multiplier boards (boards 3, 4). The spacing between boards 1 and 2 may be considerably less than the spacing between boards 3 and 4, e.g., approximately 0.5 cm spacing as compared to approximately 2.0 cm spacing. The spacing between boards 2 and 3 should at least be as large as the spacing between boards 3 and 4.
[0033] FIG. 8 illustrates an insulation system 250 for providing improved insulation for the circuit boards of the voltage multiplier module 200 in accordance with an implementation of the present invention. The insulation system 250 provides increased breakdown voltage magnitude by controlling the electric field within a multilayer insulation system. The insulation system 250 is constructed from n layers of insulation 252-1, ..., 252-n and m very thin (e.g., less than approximately 1 mm) conducting strips 254-1, ..., 254-m placed between successive insulating layers. Each layer 252-1, ..., 252-n may be composed of a suitable dielectric material, such as a polymeric insulating material. The width of each conducting strip 254-1, ..., 254-m is represented by ICa, the thickness of each insulting later is represented by b, the width of the composite insulation system 250 is represented by A, and the thickness of the composite insulation system is represented by B.
In one implementation of the present invention the multiplier insulation composite is provided on both sides of the voltage multiplier boards (e.g., boards 3 and 4 is FIG. 7).
[0034] The conducting strips 254-1, ..., 254-m in each layer 252 are symmetrically shifted by a constant distance relative to the conductive strips in an adjacent layer, determined by a shifting factor which maximizes the withstand voltage of the insulation. The potential distribution within the composite insulation may be determined from the solution of the Laplace equation for different design parameters, to reduce the electric field. Principles underlying the insulation system 250 are described in detail in the article titled "Design of Field-Controlled Multi-layer Insulation System," IEEE Transactions on Electrical Insulation, Vol.
EI-21, No. 2, April 1986, which is herein incorporated by reference in its entirety. It should be recognized that parameters of the insulation system 250, such as n and m, will depend on the particular application and characteristics of the HVPS 10.
[0035] To improve heat dissipation and enable the position of a greater number of voltage multiplier capacitors in a given area, one implementation of the present invention arranges the capacitors 410 of cascaded voltage doubling stages in a zig-zag pattern relative.to the diodes 420, as illustrated in FIG. 9. By combining the insulation and thermal management techniques set forth herein, spacing for components and clearance between voltage multiplier boards can be reduced, thereby reducing the size of the HVPS 10.
[0036] Having provided a detailed description of embodiments and exemplary implementations of the present invention, characteristics and advantages of such embodiments and exemplary implementations will next be described. Normally, in the high-voltage power supply application, a sine-wave is used as the input to the voltage multiplier. Generating a sine-wave at high-frequency is not straight forward, whereas square-wave can be produced at high frequency without a problem. Furthermore, as mentioned above, a square-wave input to the voltage multiplier module 200 facilitates voltage multiplication and increases stability of the HVPS
under load. Thus, in a preferred embodiment, the HVPS 10 uses a high-frequency square-wave as the input to the voltage multiplier module 10 200.
[0037] Normally, high-voltage power supplies are large in size due to insulation requirements and the necessary number of voltage doubling stages. If it were attempted to use a high voltage square-wave as the input to voltage multiplier of a typical power supply, the high-voltage square-wave present within the stages of voltage multiplier would tend to cause flash-over (spark) between the component terminals which have high potential difference with respect to one another. Therefore, high-voltage power supplies that would attempt to use a high-voltage square-wave cannot be made small in size. The HVPS 10 of this invention uses high-frequency square-wave at high voltage levels in a small board size.
[0038] One advantage of using high-frequency operation is reduction of the magnetic components, i.e., isolation/step-up transformer in the push-pull converter and inductor in the buck converter. Another reason for using high-frequency operation is to reduce the output voltage ripple and voltage drop under load, as will be apparent from the following discussion.
[0039] The output voltage ripple is given by the following relation:
8V = I ~~~+1) __ q ( 1 + ~ +...+ ~ ) (equation 1 ) fC 4 2 Cl C2 C, where n is the number of voltage doubter stages in the voltage multiplier and q is the charge transferred from the capacitors to the load per period.
With Cl = C2 = ... = Cn , the output voltage ripple becomes:
~~ - q n(n + 1) - I n(h + 1) (equation 2) C 4 fC 4 As seen, as f is increased, ~T~ will decrease. This is one advantage of high-frequency operation.
[0040] The output voltage drop under load ( OVo ) due to current drawn from the capacitors and transferred to the load is given by:
OT~o = fC (2 3 +2 2 +...- 3) (equation 3) [0041] The above relation can be simplified for h)5 as follows:
OVo - ~C, 23 for n)5 (equation 4) or ~R'27~3 ~~O - C 3 for n)~5 (equation 5) [0042] As seen, as f is increased, OVo will decrease and as a result the maximum output voltage will increase. The maximum output voltage is given by the following equation:
~o,max =~(2~max.,noload)-~~o (equation 6) [0043] In the HVPS 10 of the present invention, the potential for flash-over caused by using high-frequency square-wave is addressed through implementation of a printed circuit board and insulation design.
For insulation, a field-controlled dielectric material utilizing the bulk of the insulating material rather than its surface may be used. The bulk is known to have a dielectric strength of at least 10 times that of the surface.
Therefore, the size of the board carrying the voltage multiplier can be dramatically reduced. With special design of the dielectric layers, the electric field is forced to penetrate into the material where the dielectric strength is high, instead of being set up on the surface.
[0044] The power electronics control module 300 has two main tasks. The first main task is to control the buck converter to produce scaled voltage from the source voltage based on a command signal, e.g., an analog command signal between 0 and 10 V, DC. A closed-loop control adjusts the magnitude of the buck converter output voltage by adjusting the duty cycle of the MOSFET switch to control the high-voltage output of the power supply between 0 and 30 kV based on the command signal. The second main task of the power electronics control module 300 is to produce the 'gating pulses for the push-pull converter MOSFET
switches. In one implementation, these switches are operated at 50%
duty ratio and 100 kHz switching frequency. In a preferred embodiment, analog control has been selected for the control of the buck converter to take advantage of its high-speed characteristics (e.g., achieving a stable output voltage within milliseconds of the input command). To achieve reference tracking at the switching rate of 100 kHz using digital control, a large and expensive microcontroller or DSP board is needed. Analog control guarantees very good results at a reasonable size and cost.
[0045] ~ The HVPS 10 of the present invention may be characterized by small size and weight (size: 5 cm x 10 cm x 15 cm, weight: < 500 grams). As a result of using an insulation system 250 in the voltage multiplier module 200, the size and clearance of the printed circuit boards housing the voltage doubter stages may be dramatically reduced. Existing high-voltage power supplies using voltage multipliers formed of capacitors and diodes are large in size, as the limited insulation strength and conventional insulation design does not allow small board size and clearances. Furthermore, due to operating at a high switching frequency, the sizes and weights of the magnetic components (buck converter inductor and push-pull converter transformer) have been dramatically reduced. A third factor contributing to the reduction in size and weight of the high-voltage power supply is the fact that the power supply may be naturally cooled and does not need any additional cooling mechanism (e.g., a fan) beside a heat sink used for the switches elements.
[0046] As discussed above, the HVPS can be installed in a variety of environments, for example, MIL-E-5400 requirement,,which defines environmental conditions under which the equipment must operate (including ambient temperature and pressure ranges that may be encountered by the equipment), or an unpressurized environment with ambient temperature up to 70 degree C. The operating altitude may vary from sea level to 16,000m.
Claims (15)
1. A high-voltage power supply (10), comprising:
a power scaling section (130) receiving an input voltage signal and converting said input voltage signal to a controllable DC voltage;
a push-pull converter (140) for converting said controllable DC voltage to a high-frequency wave; and a voltage multiplier (200) receiving said high-frequency wave generated by said push-pull (140) converter and performing successive voltage doubling operations to generate a high-voltage DC output.
a power scaling section (130) receiving an input voltage signal and converting said input voltage signal to a controllable DC voltage;
a push-pull converter (140) for converting said controllable DC voltage to a high-frequency wave; and a voltage multiplier (200) receiving said high-frequency wave generated by said push-pull (140) converter and performing successive voltage doubling operations to generate a high-voltage DC output.
2. The high-voltage power supply (10) of claim 1, further comprising:
a control module (300) for controlling said power scaling section (130) and said push-pull converter (140).
a control module (300) for controlling said power scaling section (130) and said push-pull converter (140).
3. The high-voltage power supply (10) according to claim 2, wherein said power scaling section (130) includes a switching element (Q1), a duty cycle of which controls the amplitude of said controllable DC voltage, and said control module (300) outputs a gate switching signal to said switching element (Q1) of said power scaling section (130) as a function of a desired output voltage of the high-voltage power supply (10).
4. The high-voltage power supply (10) according to claim 3, wherein said control module (300) receives a feedback signal based on the output of said power scaling section (130) to adjust said gate switching signal.
5. The high-voltage power supply (10) according to claim 1, wherein said high-frequency wave is a square wave.
6. The high-voltage power supply (10) according to claim 1, wherein the frequency of said high-frequency wave is approximately 100 kHz.
7. The high-voltage power supply (10) according to claim 1, wherein said controllable DC voltage is in the range of approximately 0-to28 kV.
8. The high-voltage power supply (10) according to claim 1, wherein said power supply generates an output voltage of in the range of approximately 0-to-30 kV, DC.
9. The high-voltage power supply (10) according to claim 2, wherein said control module (300) is an analog controller.
10. The high-voltage power supply (10) according to claim 1, wherein said voltage multiplier (200) includes a plurality of voltage doubler stages (210-1,...,200-n) on a circuit board and said high-voltage power supply (10) further comprises an insulation system (250) associated with said circuit board.
11. The high-voltage power supply according to claim 10, wherein said insulation system (250) is a multi-layer system of n layers of insulation and m conducting strips positioned between successive insulating layers.
12. The high-voltage power supply according to claim 10, wherein said insulation system (250) is a field-controlled multi-layer insulation system.
13. The high-voltage power supply (10) according to claim 10, wherein said plurality of voltage doublet stages (210-1,...,210-n) are divided among multiple circuit boards, separate from said power scaling section (130) and said push-pull converter (140).
14. The high-voltage power supply (10) according to claim 10, wherein said plurality of voltage doubter stages (210-1,...210-n) include capacitors (410) arranged in a zig-zag pattern.
15. A method for providing high-voltage power, comprising:
receiving an input voltage signal and scaling said input voltage signal to a controllable DC voltage;
converting said controllable DC voltage to a high-frequency wave; and performing voltage multiplication on said high-frequency wave generated by said converting step to generate a high-voltage DC output.
receiving an input voltage signal and scaling said input voltage signal to a controllable DC voltage;
converting said controllable DC voltage to a high-frequency wave; and performing voltage multiplication on said high-frequency wave generated by said converting step to generate a high-voltage DC output.
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| US10/630,684 US7477529B2 (en) | 2002-11-01 | 2003-07-31 | High-voltage power supply |
| PCT/US2003/034219 WO2004042907A1 (en) | 2002-11-01 | 2003-10-28 | High-voltage power supply |
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Families Citing this family (44)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR100635282B1 (en) * | 2005-08-11 | 2006-10-19 | 삼성전자주식회사 | Apparatus for generating high voltage by digital control and method thereof |
| JP4719567B2 (en) | 2005-12-21 | 2011-07-06 | 日立オートモティブシステムズ株式会社 | Bidirectional DC-DC converter and control method thereof |
| US8861228B2 (en) * | 2009-12-07 | 2014-10-14 | Durr Systems Gmbh | High voltage controller with improved monitoring and diagnostics |
| WO2011077583A1 (en) * | 2009-12-26 | 2011-06-30 | キヤノン株式会社 | High-voltage power supply |
| US9278846B2 (en) | 2010-09-18 | 2016-03-08 | Fairchild Semiconductor Corporation | Micromachined monolithic 6-axis inertial sensor |
| WO2012037501A2 (en) | 2010-09-18 | 2012-03-22 | Cenk Acar | Flexure bearing to reduce quadrature for resonating micromachined devices |
| KR101352827B1 (en) | 2010-09-18 | 2014-01-17 | 페어차일드 세미컨덕터 코포레이션 | Micromachined 3-axis accelerometer with a single proof-mass |
| US9278845B2 (en) | 2010-09-18 | 2016-03-08 | Fairchild Semiconductor Corporation | MEMS multi-axis gyroscope Z-axis electrode structure |
| WO2012037536A2 (en) | 2010-09-18 | 2012-03-22 | Fairchild Semiconductor Corporation | Packaging to reduce stress on microelectromechanical systems |
| EP2616389B1 (en) | 2010-09-18 | 2017-04-05 | Fairchild Semiconductor Corporation | Multi-die mems package |
| US10065851B2 (en) | 2010-09-20 | 2018-09-04 | Fairchild Semiconductor Corporation | Microelectromechanical pressure sensor including reference capacitor |
| GB2509009B (en) * | 2011-08-30 | 2016-03-09 | Ge Aviat Systems Ltd | Power distribution in aircraft |
| US9062972B2 (en) | 2012-01-31 | 2015-06-23 | Fairchild Semiconductor Corporation | MEMS multi-axis accelerometer electrode structure |
| US9488693B2 (en) | 2012-04-04 | 2016-11-08 | Fairchild Semiconductor Corporation | Self test of MEMS accelerometer with ASICS integrated capacitors |
| EP2648334B1 (en) | 2012-04-05 | 2020-06-10 | Fairchild Semiconductor Corporation | Mems device front-end charge amplifier |
| EP2647952B1 (en) | 2012-04-05 | 2017-11-15 | Fairchild Semiconductor Corporation | Mems device automatic-gain control loop for mechanical amplitude drive |
| EP2647955B8 (en) | 2012-04-05 | 2018-12-19 | Fairchild Semiconductor Corporation | MEMS device quadrature phase shift cancellation |
| US9625272B2 (en) | 2012-04-12 | 2017-04-18 | Fairchild Semiconductor Corporation | MEMS quadrature cancellation and signal demodulation |
| KR101999745B1 (en) * | 2012-04-12 | 2019-10-01 | 페어차일드 세미컨덕터 코포레이션 | Micro-electro-mechanical-system(mems) driver |
| CN102624244B (en) * | 2012-04-21 | 2014-04-16 | 桂林理工大学 | Implementation method for high-isolation adjustable direct current regulated power supply |
| DE102013014881B4 (en) | 2012-09-12 | 2023-05-04 | Fairchild Semiconductor Corporation | Enhanced silicon via with multi-material fill |
| US9759564B2 (en) | 2013-03-15 | 2017-09-12 | Fairchild Semiconductor Corporation | Temperature and power supply calibration |
| WO2015073921A1 (en) | 2013-11-14 | 2015-05-21 | Eagle Harbor Technologies, Inc. | This disclosure relates generally to a high voltage nanosecond pulser. |
| US11539352B2 (en) | 2013-11-14 | 2022-12-27 | Eagle Harbor Technologies, Inc. | Transformer resonant converter |
| US10892140B2 (en) | 2018-07-27 | 2021-01-12 | Eagle Harbor Technologies, Inc. | Nanosecond pulser bias compensation |
| US10020800B2 (en) | 2013-11-14 | 2018-07-10 | Eagle Harbor Technologies, Inc. | High voltage nanosecond pulser with variable pulse width and pulse repetition frequency |
| US10978955B2 (en) | 2014-02-28 | 2021-04-13 | Eagle Harbor Technologies, Inc. | Nanosecond pulser bias compensation |
| US10483089B2 (en) | 2014-02-28 | 2019-11-19 | Eagle Harbor Technologies, Inc. | High voltage resistive output stage circuit |
| US9835647B2 (en) | 2014-03-18 | 2017-12-05 | Fairchild Semiconductor Corporation | Apparatus and method for extending analog front end sense range of a high-Q MEMS sensor |
| US11004660B2 (en) | 2018-11-30 | 2021-05-11 | Eagle Harbor Technologies, Inc. | Variable output impedance RF generator |
| US11430635B2 (en) | 2018-07-27 | 2022-08-30 | Eagle Harbor Technologies, Inc. | Precise plasma control system |
| EP4266579A3 (en) | 2017-02-07 | 2023-12-27 | Eagle Harbor Technologies, Inc. | Transformer resonant converter |
| WO2019040949A1 (en) | 2017-08-25 | 2019-02-28 | Eagle Harbor Technologies, Inc. | Arbitarary waveform generation using nanosecond pulses |
| CN109995240A (en) | 2018-01-02 | 2019-07-09 | 通用电气公司 | Power adapter and power conversion method |
| US11145455B2 (en) | 2018-07-17 | 2021-10-12 | General Electric Company | Transformer and an associated method thereof |
| US11532457B2 (en) | 2018-07-27 | 2022-12-20 | Eagle Harbor Technologies, Inc. | Precise plasma control system |
| US11222767B2 (en) | 2018-07-27 | 2022-01-11 | Eagle Harbor Technologies, Inc. | Nanosecond pulser bias compensation |
| US11302518B2 (en) | 2018-07-27 | 2022-04-12 | Eagle Harbor Technologies, Inc. | Efficient energy recovery in a nanosecond pulser circuit |
| CN112805920A (en) | 2018-08-10 | 2021-05-14 | 鹰港科技有限公司 | Plasma sheath control for RF plasma reactor |
| WO2020146436A1 (en) * | 2019-01-08 | 2020-07-16 | Eagle Harbor Technologies, Inc. | Efficient energy recovery in a nanosecond pulser circuit |
| TWI778449B (en) | 2019-11-15 | 2022-09-21 | 美商鷹港科技股份有限公司 | High voltage pulsing circuit |
| KR20230150396A (en) | 2019-12-24 | 2023-10-30 | 이글 하버 테크놀로지스, 인코포레이티드 | Nanosecond pulser rf isolation for plasma systems |
| CN113625143B (en) * | 2021-07-01 | 2024-02-06 | 深圳供电局有限公司 | Ultralow frequency cosine square wave generating device and driving method thereof |
| RU209671U1 (en) * | 2021-12-13 | 2022-03-17 | Федеральное государственное бюджетное учреждение науки Федеральный исследовательский центр "Кольский научный центр Российской академии наук" (ФИЦ КНЦ РАН) | HIGH VOLTAGE STABILIZED POWER SUPPLY |
Family Cites Families (29)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US20653A (en) * | 1858-06-22 | Improved corn-husker | ||
| US3721824A (en) * | 1967-04-10 | 1973-03-20 | Packard Instrument Co Inc | Apparatus and method for adjusting gain of photomultiplier type scintillation detectors |
| US3573595A (en) * | 1969-05-28 | 1971-04-06 | Venus Scient Inc | Constant current feedback regulator with adjustable impedance for maintaining constant current |
| DE1929520B2 (en) | 1969-06-11 | 1971-08-15 | HIGH VOLTAGE DEVICE FOR IMAGE CONVERTER TUBES SMALLER DIAMETER | |
| US3840797A (en) * | 1970-12-28 | 1974-10-08 | Us Navy | Related power supply |
| GB1425506A (en) * | 1972-03-03 | 1976-02-18 | Redelec | High-voltage dc generator in an electrostatic apparatus |
| US3731179A (en) * | 1972-09-22 | 1973-05-01 | Gen Electric | Adjustable high voltage power supply with output polarity switching |
| IT1006799B (en) * | 1973-02-01 | 1976-10-20 | Air Ind | WATERFALL MULTIPLIER AND BUILT-IN ELECTROSTATIC SPRAY GUN |
| US3819942A (en) * | 1973-05-07 | 1974-06-25 | Savin Business Machines Corp | Regulated power supply for corona charging unit |
| US4012687A (en) * | 1975-08-07 | 1977-03-15 | E. I. Du Pont De Nemours And Company | Driver amplifier for adjustable high voltage power supply |
| US4060726A (en) * | 1976-10-18 | 1977-11-29 | Beckman Instruments, Inc. | Gamma counter calibration system |
| US4151412A (en) * | 1976-10-22 | 1979-04-24 | Beckman Instruments, Inc. | Method and apparatus for automatic spectrum scanning in a proportional counter |
| US4112354A (en) * | 1976-11-09 | 1978-09-05 | General Cable Corporation | Mobile bridge test apparatus and method utilizing a sub-power frequency test signal for cable system evaluation |
| US4251857A (en) * | 1979-02-21 | 1981-02-17 | Sperry Corporation | Loss compensation regulation for an inverter power supply |
| US4517472A (en) * | 1983-07-06 | 1985-05-14 | The United States Of America As Represented By The Administrator Of The National Aeronautics And Space Administration | High voltage power supply |
| US4723079A (en) * | 1986-03-27 | 1988-02-02 | Peter Norton | Vehicle power supply with regulated voltage and adjustable voltage outputs |
| US4893227A (en) * | 1988-07-08 | 1990-01-09 | Venus Scientific, Inc. | Push pull resonant flyback switchmode power supply converter |
| FR2663169A1 (en) * | 1990-06-08 | 1991-12-13 | Alcatel Espace | DEVICE FOR REGULATING A PARAMETER BY A BIDIRECTIONAL CURRENT STRUCTURE. |
| ATE138480T1 (en) * | 1990-12-17 | 1996-06-15 | Patented Devices Pty Ltd | MONITORING PARTIAL DISCHARGES |
| US5414224A (en) * | 1991-04-01 | 1995-05-09 | Filial Vsesojuznogo Nauchno Issledovatelskogo Instituta | Multilayer printed circuit board and method of manufacturing same |
| IT1247356B (en) * | 1991-06-17 | 1994-12-12 | Plessey Spa | CONTINUOUS HIGH VOLTAGE GENERATION EQUIPMENT ESTABLISHED IN PARTICULAR FOR USE IN COMBINATION WITH AN ANTI-POLLUTION EXHAUST |
| US5218194A (en) * | 1991-08-19 | 1993-06-08 | Varo Inc. | Advanced high voltage power supply for night vision image intensifer |
| US5578112A (en) * | 1995-06-01 | 1996-11-26 | 999520 Ontario Limited | Modular and low power ionizer |
| JPH10134989A (en) | 1996-10-28 | 1998-05-22 | Origin Electric Co Ltd | Dc high voltage generation device for tubular bulb |
| WO1998033267A2 (en) * | 1997-01-24 | 1998-07-30 | Fische, Llc | High efficiency power converter |
| EP0944162B1 (en) * | 1998-03-19 | 2009-02-18 | Alcatel Lucent | Auto-synchronized DC/DC converter and method of operating same |
| US6157021A (en) * | 1998-11-02 | 2000-12-05 | The United States Of America As Represented By The Secretary Of The Navy | Active regulator for image intensifier power supply |
| US7712687B2 (en) | 1999-08-18 | 2010-05-11 | The Procter & Gamble Company | Electrostatic spray device |
| US6370039B1 (en) * | 1999-11-19 | 2002-04-09 | Iwatt | Isolated power converter having primary feedback control |
-
2003
- 2003-07-31 US US10/630,684 patent/US7477529B2/en not_active Expired - Fee Related
- 2003-10-28 KR KR1020057007814A patent/KR20050067223A/en not_active Application Discontinuation
- 2003-10-28 WO PCT/US2003/034219 patent/WO2004042907A1/en active Application Filing
- 2003-10-28 AU AU2003285065A patent/AU2003285065A1/en not_active Abandoned
- 2003-10-28 CA CA002504438A patent/CA2504438A1/en not_active Abandoned
- 2003-10-28 EP EP03779380A patent/EP1563596A1/en not_active Withdrawn
- 2003-10-28 JP JP2005502194A patent/JP2006505239A/en not_active Withdrawn
- 2003-10-31 TW TW092130489A patent/TW200423514A/en unknown
Also Published As
| Publication number | Publication date |
|---|---|
| EP1563596A1 (en) | 2005-08-17 |
| WO2004042907A1 (en) | 2004-05-21 |
| TW200423514A (en) | 2004-11-01 |
| KR20050067223A (en) | 2005-06-30 |
| US7477529B2 (en) | 2009-01-13 |
| AU2003285065A1 (en) | 2004-06-07 |
| US20040085784A1 (en) | 2004-05-06 |
| JP2006505239A (en) | 2006-02-09 |
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