CA2652407A1 - Electronic biosensor arrangement - Google Patents
Electronic biosensor arrangement Download PDFInfo
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- CA2652407A1 CA2652407A1 CA002652407A CA2652407A CA2652407A1 CA 2652407 A1 CA2652407 A1 CA 2652407A1 CA 002652407 A CA002652407 A CA 002652407A CA 2652407 A CA2652407 A CA 2652407A CA 2652407 A1 CA2652407 A1 CA 2652407A1
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- sensor
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- differential amplifier
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01N—INVESTIGATING OR ANALYSING MATERIALS BY DETERMINING THEIR CHEMICAL OR PHYSICAL PROPERTIES
- G01N27/00—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means
- G01N27/26—Investigating or analysing materials by the use of electric, electrochemical, or magnetic means by investigating electrochemical variables; by using electrolysis or electrophoresis
- G01N27/28—Electrolytic cell components
- G01N27/30—Electrodes, e.g. test electrodes; Half-cells
- G01N27/327—Biochemical electrodes, e.g. electrical or mechanical details for in vitro measurements
- G01N27/3275—Sensing specific biomolecules, e.g. nucleic acid strands, based on an electrode surface reaction
- G01N27/3276—Sensing specific biomolecules, e.g. nucleic acid strands, based on an electrode surface reaction being a hybridisation with immobilised receptors
-
- C—CHEMISTRY; METALLURGY
- C12—BIOCHEMISTRY; BEER; SPIRITS; WINE; VINEGAR; MICROBIOLOGY; ENZYMOLOGY; MUTATION OR GENETIC ENGINEERING
- C12Q—MEASURING OR TESTING PROCESSES INVOLVING ENZYMES, NUCLEIC ACIDS OR MICROORGANISMS; COMPOSITIONS OR TEST PAPERS THEREFOR; PROCESSES OF PREPARING SUCH COMPOSITIONS; CONDITION-RESPONSIVE CONTROL IN MICROBIOLOGICAL OR ENZYMOLOGICAL PROCESSES
- C12Q1/00—Measuring or testing processes involving enzymes, nucleic acids or microorganisms; Compositions therefor; Processes of preparing such compositions
- C12Q1/68—Measuring or testing processes involving enzymes, nucleic acids or microorganisms; Compositions therefor; Processes of preparing such compositions involving nucleic acids
- C12Q1/6813—Hybridisation assays
- C12Q1/6816—Hybridisation assays characterised by the detection means
Abstract
Electronic biosensor arrangement (1) comprising a receptacle region (2) for biological material, to which is assigned a sensor electrode arrangement (3) with sensor electrodes (3a, 3b) intermeshing in comblike fashion, to which can be connected a measuring circuit (5) for measuring an electrical measurement quantity influenced by biological material at the sensor electrodes (3a, 3b), wherein the sensor electrodes (3a, 3b) form a multiplicity of sensor capacitors (Csensor,1 to Csensor,N) which are assigned electronic switching means (S2,1 to S2,N) - driven by a control logic (8) - for connection to earth or to a voltage source carrying a measurement voltage (Vdrive), comprising a reference capacitor (Cref), which is likewise assigned a switching means (S1) - driven by the control logic (8) - for optional connection to earth or to the measurement voltage (Vdrive), wherein the capacitors (Cref, Csensor,1 - Csensor,N) are on the other hand combined at a node (A), which is connected to an input (-) of a differential amplifier (10) and to which are furthermore connected measuring capacitors (C0 to C5) having binary weighted capacitances (Cmin to 32Cmin) for forming an SAR comparison unit, which, on the other hand, via switching means (S3,0 to S3,5) driven by the control logic (8), are able to be connected selectively to a voltage source (Varray) or to earth for the purpose of charge difference formation for the SAR conversion.
Description
Electronic biosensor arrangement The invention relates to an electronic biosensor arrange-ment, comprising a receiving region for biological material, a sensor-electrode arrangement with sensor electrodes which inter-mesh in a comb-like manner being associated therewith, to which sensor electrodes a measuring circuit is connectable for measur-ing an electrical measurement parameter on the sensor electrodes and influenced by biological material.
Such a biosensor arrangement is known, e.g., from the art-icle written by E. Laureyn et al., "Nanoscaled interdigitated titanium electrodes for impedimetric biosensing", Sensors and Actuators B, vol. 68 (2000), pp 360-370. Here, the object is to detect affinity-based interactions between complementary mo-lecules, wherein a binding of target molecules to selective probe coatings is caused in the electric properties in the re-gion of the electrodes intermeshing in a comb-like manner. These changes may be detected as an impedance shift, thus obtaining a direct electric signal referred to the affinity binding. In this manner, e.g., the immobilization of the glucose oxidase can be monitored using impedance spectroscopy.
For biological purposes it is often sufficient to distin-guish simply between the states of binding and non-binding. Usu-ally, the reactions, which occur when, e.g., a DNA or a different specific material (depending on the application) binds to receptors on the electrodes, are slow, taking several seconds, as a,rule. It involves relatively great efforts to mon-itor such biological reaction processes by means of the known biosensor arrangement, wherein the expenditures on equipment re-quired for monitoring a plurality of such reaction proce.sses are also high.
Studies have shown that there will be a 20% magnitude change in the dielectric constant of a primarily aqueous solution, which is present between the electrodes, during a reaction pro-cess, e.g. when a DNA or a different biological material, de-pending on the application, is being bound to receptors on the electrodes, with water (having a relatively high dielectric con-stant) being displaced by reactants with a low dielectric con-stant; this means that the capacity between two electrodes of such a capacitive biosensor formed by a pair of electrodes may be reduced to be as low as about 80% of the initial capacity value. Therefore, such capacity values and changes in the capa-city values should be detected, it being desirable to monitor a large number of reaction processes and, thus, biosensors at the same time, and the efforts involved in measurement need never-theless be kept low.
Thus, it is an object of the invention to propose a bio-sensor arrangement as initially described, by means of which a large number of biological reaction processes can be monitored at the same time, with e.g. 10,000 simultaneous processes being definitely conceivable, and wherein the expenditure on equipment necessary shall be kept low.
In this context, the invention is based on the finding that the usually used on-chip-analyzing systems for biochemistry of-fer the uncomplicated possibility of realizing individual bio-sensors in the mentioned large number on one chip and of reading these biosensors at a correspondingly high rate as regards a ca-pacity decrease, wherein, basically, a principle known per se from other measurement applications, i.e. the SAR technique (SAR
- successive approximation registers), may furthermore be used for capacity measurement so as to quickly achieve a measurement result in digital form which is sufficiently accurate. This SAR
technique is known, e.g., for use with acceleration sensors and pressure sensors, and it is based on measurements of capacity differences, cf., e.g., the article written by Joseph T. Kung et al., "Digital Cancellation of Noise and Offset for Capacitive Sensors", Transactions on Instrumentation and Measurement, vol 42, no 5, October 1993, pp 939-942. This SAR technique known per se may be advantageously used within the scope of the present biosensor arrangement in a modified form with numerous bio-sensors.
Accordingly, the invention provides for an electric bio-sensor arrangement as defined in the enclosed claim 1. Advant-ageous embodiments and further developments are set out in the dependent claims.
With the inventive biosensor arrangement, the individual sensor capacitors formed by the respective sensor electrodes, i.e. biosensors, are one by one connected to the measurement voltage and "read", wherein a SAR approximation is carried out for each reading, i.e. for each measurement-value detection, wherein, at a correspondingly higher rate, the individual meas-urement capacitors, starting with the measurement capacitor of the highest value in the output of the digital measurement res-ults in correspondence with the bit of the highest value, will be connected and will be switched off as a function of the re-spective result of difference, or will remain connected when the measurement capacitor of the next-lowest value will subsequently be connected. This successive approximation finally provides a digital signal which indicates the capacity difference between the reference capacitor and the respective sensor capacitor at a predefined resolution, in correspondence with the smallest meas-urement capacitor. Thus, the capacity value of the respective sensor capacitor will be obtained by subtracting said capacity difference from the capacity value of the reference capacitor.
However, for further data processing, already the value of capa-city difference is often sufficient which will be obtained imme-diately in digital form and may thus be immediately further processed.
While the one input of the differential amplifier, in par-ticular the inverting input (-) is supplied with the net voltage given by the total charges on the respective capacitors, the other input, i.e. the non-inverting input (+), may be preferably connected to earth, yet with an offset voltage possibly occur-ring. To provide remedy in this context, it is advantageous if the differential amplifier has a feedback branch extending from its output to the one input, wherein a switch controlled by the control logic is arranged in said feedback branch for dlosing and opening the latter, and wherein the differential amplifier acts as a comparator during the SAR approximation phase when the feedback branch is open, whereas when the feedback branch is closed, the differential amplifier, during an initializing phase feedbacks an amplifier-offset voltage to the node for the sub-sequent compensation. Here, for reasons of a safe measurement, it is advantageously further provided that after the initializ-ing phase the switch provided in the feedback branch will be driven by the control logic either simultaneously with the switching of the switching means assigned to the reference capa-citor or directly before said switching means will be switched so as to connect the reference capacitor to earth rather than to measurement voltage.
Such a biosensor arrangement is known, e.g., from the art-icle written by E. Laureyn et al., "Nanoscaled interdigitated titanium electrodes for impedimetric biosensing", Sensors and Actuators B, vol. 68 (2000), pp 360-370. Here, the object is to detect affinity-based interactions between complementary mo-lecules, wherein a binding of target molecules to selective probe coatings is caused in the electric properties in the re-gion of the electrodes intermeshing in a comb-like manner. These changes may be detected as an impedance shift, thus obtaining a direct electric signal referred to the affinity binding. In this manner, e.g., the immobilization of the glucose oxidase can be monitored using impedance spectroscopy.
For biological purposes it is often sufficient to distin-guish simply between the states of binding and non-binding. Usu-ally, the reactions, which occur when, e.g., a DNA or a different specific material (depending on the application) binds to receptors on the electrodes, are slow, taking several seconds, as a,rule. It involves relatively great efforts to mon-itor such biological reaction processes by means of the known biosensor arrangement, wherein the expenditures on equipment re-quired for monitoring a plurality of such reaction proce.sses are also high.
Studies have shown that there will be a 20% magnitude change in the dielectric constant of a primarily aqueous solution, which is present between the electrodes, during a reaction pro-cess, e.g. when a DNA or a different biological material, de-pending on the application, is being bound to receptors on the electrodes, with water (having a relatively high dielectric con-stant) being displaced by reactants with a low dielectric con-stant; this means that the capacity between two electrodes of such a capacitive biosensor formed by a pair of electrodes may be reduced to be as low as about 80% of the initial capacity value. Therefore, such capacity values and changes in the capa-city values should be detected, it being desirable to monitor a large number of reaction processes and, thus, biosensors at the same time, and the efforts involved in measurement need never-theless be kept low.
Thus, it is an object of the invention to propose a bio-sensor arrangement as initially described, by means of which a large number of biological reaction processes can be monitored at the same time, with e.g. 10,000 simultaneous processes being definitely conceivable, and wherein the expenditure on equipment necessary shall be kept low.
In this context, the invention is based on the finding that the usually used on-chip-analyzing systems for biochemistry of-fer the uncomplicated possibility of realizing individual bio-sensors in the mentioned large number on one chip and of reading these biosensors at a correspondingly high rate as regards a ca-pacity decrease, wherein, basically, a principle known per se from other measurement applications, i.e. the SAR technique (SAR
- successive approximation registers), may furthermore be used for capacity measurement so as to quickly achieve a measurement result in digital form which is sufficiently accurate. This SAR
technique is known, e.g., for use with acceleration sensors and pressure sensors, and it is based on measurements of capacity differences, cf., e.g., the article written by Joseph T. Kung et al., "Digital Cancellation of Noise and Offset for Capacitive Sensors", Transactions on Instrumentation and Measurement, vol 42, no 5, October 1993, pp 939-942. This SAR technique known per se may be advantageously used within the scope of the present biosensor arrangement in a modified form with numerous bio-sensors.
Accordingly, the invention provides for an electric bio-sensor arrangement as defined in the enclosed claim 1. Advant-ageous embodiments and further developments are set out in the dependent claims.
With the inventive biosensor arrangement, the individual sensor capacitors formed by the respective sensor electrodes, i.e. biosensors, are one by one connected to the measurement voltage and "read", wherein a SAR approximation is carried out for each reading, i.e. for each measurement-value detection, wherein, at a correspondingly higher rate, the individual meas-urement capacitors, starting with the measurement capacitor of the highest value in the output of the digital measurement res-ults in correspondence with the bit of the highest value, will be connected and will be switched off as a function of the re-spective result of difference, or will remain connected when the measurement capacitor of the next-lowest value will subsequently be connected. This successive approximation finally provides a digital signal which indicates the capacity difference between the reference capacitor and the respective sensor capacitor at a predefined resolution, in correspondence with the smallest meas-urement capacitor. Thus, the capacity value of the respective sensor capacitor will be obtained by subtracting said capacity difference from the capacity value of the reference capacitor.
However, for further data processing, already the value of capa-city difference is often sufficient which will be obtained imme-diately in digital form and may thus be immediately further processed.
While the one input of the differential amplifier, in par-ticular the inverting input (-) is supplied with the net voltage given by the total charges on the respective capacitors, the other input, i.e. the non-inverting input (+), may be preferably connected to earth, yet with an offset voltage possibly occur-ring. To provide remedy in this context, it is advantageous if the differential amplifier has a feedback branch extending from its output to the one input, wherein a switch controlled by the control logic is arranged in said feedback branch for dlosing and opening the latter, and wherein the differential amplifier acts as a comparator during the SAR approximation phase when the feedback branch is open, whereas when the feedback branch is closed, the differential amplifier, during an initializing phase feedbacks an amplifier-offset voltage to the node for the sub-sequent compensation. Here, for reasons of a safe measurement, it is advantageously further provided that after the initializ-ing phase the switch provided in the feedback branch will be driven by the control logic either simultaneously with the switching of the switching means assigned to the reference capa-citor or directly before said switching means will be switched so as to connect the reference capacitor to earth rather than to measurement voltage.
Likewise, it is also of advantage if a resistance is connec-ted upstream of the one input of the differential amplifier for frequency damping in connection with an input capacity of the differential amplifier during the initializing phase.
Similarly to the sensor capacitors, the reference capacitor may be formed on the common chip, yet, the reference capacitor will be kept free of biological material, i.e. no reaction pro-cess will take place in its region so that there will be no change in its capacity value either. As to a structure which is particularly simple then as well as to a simple detection of measurement results, it is beneficial if the capacity value of the reference capacitor is selected so as to equate to the highest-possible capacity value of the individual sensor capa-citors, wherein the capacity values of the sensor capacitors will be reduced by reactions occurring in the biological materi-al.
Besides, also the measurement capacitors may be realized on the same chip.
In the following, the invention will be explained in more detail by way of the preferred exemplary embodiment illustrated in the drawings, yet without being restricted thereto. In de-tail:
Fig. 1 schematically shows an electronic biosensor arrange-ment according to the invention, including a following data-pro-cessing unit;
Fig. 2A shows a schematic circuit diagram of the individual biosensors and sensor capacitors with the measuring circuit as-signed thereto;
Fig. 2B shows a diagram of some of the voltage signals oc-curring in the circuit operation according to Fig. 2;
Fig. 3A shows a schematic circuit diagram of the control lo-gic of the measuring circuit according to Fig. 2A;
Fig. 3B shows different pulse signals of said control logic;
and Fig. 3C shows, in a schematic representation, a number of biosensors comprised of sensor electrodes, including switching means assigned thereto, wherein also the reference capacitor comprised of comparable electrodes is shown with switching means assigned thereto.
In Fig. 1, the basic principle of the present electronic bi-osensor arrangement 1 is schematically illustrated, wherein a sensor-electrode arrangement 3 is provided on a common chip 4 in a receiving region 2 (only very schematically indicated) for re-ceiving biological material; outside of the receiving region 2 for the biological material, the sensor-electrode arrangement 3 furthermore provides a reference capacitor Cref and measurement capacitors Co to C5. In the receiving region 2 for the biological material a number (e.g. 10,000) of sensor capacitors Csensor,i, wherein i= 1, 2, 3... N, is defined.
Here, the individual sensor capacitors Csensor,i as well as the reference capacitor Cref (and likewise the measuring capacitors Co to C5) are formed by electrodes intermeshing in a comb-like man-ner, as can be schematically seen from Fig. 3C. These capacitors Cref and/or Csensor,i are assigned to switching means S1, S2,1, S2,2, ... SZ,i ... SZ,N, and S3,0, S3,1 . . . S3,5, respectively, which may be seen from Figs. 2A and 3C, and which are specifically formed by electronic switching means, e.g., directly in the re-gion of the semiconductor chip 4, with a bus leading to the in-dividual switching means S1 to S3 of the measuring circuit 5 for control purposes. The switching means S1r S2,i and S3,0 to S3,5 are driven by a control logic 8 of the measuring circuit 5, wherein the switching means S1 and S2, depending on the respective meas-urement phase, in operation, establish an electric connection to a measurement voltage Vdrive or to earth, but they may optionally also remain floating (with the latter case relating to the switches Sz,i for the biosensors, i.e. for the sensor capacitors Csensor,i) = The switching means S2,i (wherein i = 1 ... N) and S1 are driven via a control bus 6 and a control line 7, cf. Fig. 2A. A
further, control bus 9 of the control logic 8 leads to the switching means S3,0 to S3,51 which selectively connect the binary-weighted measurement capacitors Co to C5 to a voltage source Varray or to earth. The other side of the measurement capacitors Co to C5 leads to a node A, to which also the reference capacitor Cref as well as the sensor capacitors Csensor,i are connected. This node A is connected to the one input, i.e. the inverted input (-), of a differential amplifier 10 via a resistance R,omp, with the oth-er, non-inverted input (+) thereof being connected to earth, wherein Fig. 2A additionally shows in a schematic fashion that an offset voltage Vos may possibly occur during operation. To be able to compensate for this offset voltage Vos after an initial-izing phase I(cf. Fig. 3B), the differential amplifier 10 has a feedback branch 11 including a switch So. This electronic switch So is again driven by the control logic 8 via a control line 12, this taking place in said initializing phase I prior to the measurement and approximation phase proper, which constitutes a phase of successive approximation, wherein this measurement phase M for one of the sensor capacitors Csensor,i as well as the initializing phase I are illustrated in Fig. 3B.
For the sake of completeness, in the circuit according to Fig. 2A, also a parasitic capacity present in chip 4 (cf. Fig.
1) is illustrated by a capacitor (CParasit) = Furthermore, a re-gister 13 is shown which is provided for storing the bit order per sensor capacitor Csensor,if which forms the measurement result, wherein said measurement results will then be received by a data-processing unit 14 (cf. Fig. 1) for further processing.
The register 13 is indexed for each sensor capacitor, cf.
the pulse signal "ready" in Figs. 2A and 2B, wherein said pulses are used as clock pulses for the register 13. At an input D of the register 13, the switch signals are supplied which are provided for the switching means S3,0 to S35 5 and depend on the comparison of potentials of the measurement capacitors Co to C5 with the respective sensor capacitor Csensor,if wherein this depend-ency manifests itself in the output signal (comp in) of the dif-ferential amplifier 10, said output signal being supplied at the respectively designated input comp_in of the control logic 8.
Furthermore, Fig. 2B illustrates a control signal Vb applied to the switching mearis Sl, and a control signal V6 applied to the switching means S2;i, as well as a clock signal Vclk. Here, it fol-lows that the switching means S2,; may also remain floating, with the exception of the specific switching means for the currently measured sensor capacitor Csensor,i, as will be explained in more detail below.
Fig. 3A shows the control bus 6 with individual lines for individual switch signals Vsensorl,2,3... , for the individual sensor capacitors Csensor,l ..., in more detail. These switch signals, as well as the switch signal VB supplied for the switching means.Sl for the reference capacitor Cref via the control line 7, and a further switch signal for the individual bit stages bit5, bit4, bit3, bit2, bitl and bitO in correspondence to the measurement capacitors C5, C91 ... C1, Co, are generated by an initializing and sensor-selecting circuit 15 within the control logic 8. The individual bit stages bitO to bit5 are supplied with the output signal of the differential amplifier 10, i.e. the signal comp_iri, so as to logically combine the same with the pulse sig-nal VA, cf. also Fig. 3B, with the result thereof being used for generating the control signals (on the bus 9, indicated by ar-rows) for the switching means S3,5, S3,4 ... S3,0 assigned to the measurement capacitors C5, CQ... Co. In Fig. 3A, these control signals are shown directly below the individual outputs to the individual switching means S3,5 etc.; here, the horizontal dotted line also indicates that the control signal may also remain high, as a function of the signal output and the output of the differential amplifier, i.e. as a function of the result of the comparison currently being made, as will be explained in further detail below.
Fig. 3B shows in combination the individual pulse signals Vc1k (clocked signal), VB (control signal for the switching means S1 on the control line 7, as well as to the switch So on the con-trol line 12), VA (pulse signal to the bit stages bit5 to bitO) as well as the individual switch signals S3,i, wherein, as regards the initializing phase I preceding the measurement phase M prop-er, it is illustrated that the reference capacitor Cref is applied to the measurement voltage Vdrive according to the signal VB, and also that the feedback branch 11 of the differential amplifier is closed by the switch So, whereas the switching means S3,0 to S3,5 assigned to the measurement capacitors Co to C5 are connected to -earth.
In detail, as regards the pulse signals described, it shall be additionally mentioned that the clock signal V,lk is assumed to trigger all respectiveprocesses with its rising edges. However, the sampling of the output signal comp in of the differential amplifier 10 is effected in case of a rising edge of the inver-ted clock signal, i.e. after the respective bit has had a half period of time for charging. As mentioned above, the outputs of the bit stages bit5 to bitO constitute the control signals for the switching means S3,s, ... S3,0, which are also referred to as SAR switches (SAR - successive approximation register). As men-tioned above, in Fig. 3A, these switch,signals are shown below the bit stages bit5 to bitO, wherein it is also illustrated that they rise successively. The possibility of keeping the signal in the high state, as is illustrated in dotted lines, will result if the respective approximation sum at the input of the compar-ator, i.e. at the input (-) of the differential amplifier 10, is negative.
The signal VA triggers the measurement and approximation phase proper. When the reference capacitor Cref has been charged, with all measurement capacitors C5 to Co having to be connected to earth during said charging time (initializing phase I), this signal VA, when rising, triggers the output of the first array bit, the MSB bit, in the present case the bit no. 5. Here, it should be mentioned that a design with six bits serves as an ex-ample and reflects a good compromise between the number of weighted capacitors Co, C1, ... and the limit of detection. Cer-tainly, also more (or less) measurement capacitors, e.g. 12 measurement capacitors C, may be provided.
During the initializing phase I the signal VB is high prior to the meas.urement proper and urges the switching means S1 for the reference capacitor CLef into its closed state. Likewise, the signal V. causes the switch So to close in the feedback branch 11 of the differential amplifier 10 so as to ensure a uniform feed-back during this phase. The number of clock periods during which the the signal VB is high depends on the decay time necessary after the approximation phase.
The output signal compin of the differential amplifier 10 is of interest when the latter acts as a comparator, i.e. during the measurement or approximation phase proper. The signal comp_in will be positive if the total sum of measurement capa-cities switched on is higher at the respective point of time than the absolute capacity difference.between the sensor capa-citor Csensor,i and. the reference capacitor Cref. However, the signal comp_in will be negative if several measurement capacitors have to be switched on. It is assumed that this output signal compin of the differential amplifier will be rising quickly enough after a respective bit (on the bus 9) has been switched on. This is why attention has to be paid to a quick mode of operation of the differential amplifier 10.
The circuit described allows for sampling rates of up to 15 mega samples per second to be reached at low resolutions, wherein the circuit with the SAR converter 16, which has been realized by the measurement capacitors Co to C5 as well as by a starting capacitor Cmin including the control logic 8 assigned thereto and the differential amplifier or comparator 10, may be implemented with an extraordinary compact circuit with low power input.
During operation, the sensor capacitors Csensor,l == - Csensor,n are multiplexed, wherein, for determining one of these sensor capa-citors, Csensor,ir the measurement capacitors Cn, with n = 0 to 5, are connected one after the other so as to perform SAR al-gorithm. All these capacitors are connected to the node A which is located above the resistance Rcomp at the (-)- input of the differential amplifier 10 and has a high impedance when the feedback branch 11 is open, i.e. when the switch S4 is opened. A
single sensor capacitor is selected from a number of sensor ca-pacitors for each measurement circuit, and those which are cur-rently not being measured remain floating.
If the biosensor arrangement is activated during operation, the whole digital logic plus switch-driving means will be ini-tialized; at this point of time there will be no measurement yet, and there will be no information available at the output.
In this situation the individual switches have the following states:
So open S1 connected to earth S2,a11 floating S3, all connected to earth Then, 'the measurement of the first sensor capacitor, e.g. Csensor,li will be started. As mentioned above, two phases each are provided, i.e. the initializing phase I during which the refer-ence capacitor Crefhaving a fixed capacity value is charged, and the measurement phase M proper during which the respective sensor capacitor Csensor,i is being switched on; thereafter, the measurement capacitors C5 to Co will be connected in the SAR ar-ray. The positions of the switches for the initializing phase during measurement, e.g. of the sensor capacitor Csensor, i ~ are as follows:
So closed S1 connected to Vdrive S2,1 connected to earth Sz,,,Iili floating S3,x connected to earth The addition "x" indicates that in each case all switches are involved, whereas the addition "/{1}" indicates that all switches S2 are floating, except the switch for the first sensor capacitor Csensor, l=
During the measurement phase M proper the states of the in-dividual switches are as follows:
So open S1 connected to earth S2,1 connected to Vdrive S2,%/i11 floating S3,n cn= 5,4, ... 0) these switches S3,n are actuated one after the other and optionally kept connected, depending on the output comp_in of the comparator, as already mentioned above, cf. also the switch or control signals below the control logic 8 in Fig.
3A.
As mentioned above, during the initializing phase I the feedback switch So is closed and the reference capacitor Cref is connected to the measurement voltage Vdrivei while at least the first sensor capacitor Csensor,i to be measured is connected to earth, as are the measurement capacitors C,,. The differential amplifier 10 virtually forces mass potential at its (-)- input.
During this phase the resistance RcomP serves for achieving a phantom-zero frequency compensation in combination with an input capacity. During the measurement phase or approximation phase M
following thereupon the.resistance R,omp will have practically no effect since the current flowing through the same will be negli-gible.
During this approximation or measurement phase M the ampli-fication is high at the operation frequency of the differential amplifier 10 so as to amplify the smallest input voltages during the last step of the approximation to a logic "high" or "low".
The smaller the sensor capacitors to be measured, the higher the amplification of the amplifier or comparator 10 has to be.
During this measurement phase M one sensor capacitor after the other is connected to the measurement voltage Vdrivei wherein the voltage at the node A will then become proportional to the capacity difference. The algorithm of the successive approxima-tion is used for converging to a digital measurement signal rep-resenting said capacity difference. To this end, as mentioned above, the measurement capacitors are connected to the voltage Varra,,, starting with the capacitor C5 of the highest capacity value (32 Cmin), and are kept connected, if occasion arises. This depends on the resulting sign of the voltage in the node A.
In the following, this will be explained in more detail by way of a specific example: it is assumed that all sensor capa-citors have a nominal value (without biological reaction) of about 10pF. Furthermore, it is assumed that the first sensor ca-pacitor Csensor,if influenced by a biological reaction, has de-creased to a value of 8.58pF. This change in the capacity is substantial enough to be representative for the process and may thus be detected.
The reference capacitor Cref is a capacitor on which no reac-tion takes place; its capacity value is lOpF. Correspondingly to a binary order, the measurement capacitors Co to C5 have the ca-pacity values 50fF, 100fF, 200fF, 400fF, 800fF and 1.6pF. During measurement, the difference of the capacity of the sensor capa-citor Csensor,l and the reference capacitor Cref will be stored at the node A. The charge will thus be equivalent to a capacity of 8.58pF - lOpF =-1.42pF. In the following, the measurement capa-citors CS to Co will one after the other be connected to the voltage VarraY, as described above, so as to find out whether.the equivalent charge is positive or negative. In the present ex-ample, this results in the following table of values:
C5,..0 node A equivalent compin. binary output.
remaining capacity 1.6pF -1.42pF +0.18pF negative 0 800fF -1.42pF -0.62pF positive 1 400fF -0.62pF -0.22pF positive 1 200fF -0.22pF -0.02pF positive 1 100fF -0.02pF +0.08pF negative 0 50fF -0.02pF +0.03pF negative 0 The following result will be obtained:
Similarly to the sensor capacitors, the reference capacitor may be formed on the common chip, yet, the reference capacitor will be kept free of biological material, i.e. no reaction pro-cess will take place in its region so that there will be no change in its capacity value either. As to a structure which is particularly simple then as well as to a simple detection of measurement results, it is beneficial if the capacity value of the reference capacitor is selected so as to equate to the highest-possible capacity value of the individual sensor capa-citors, wherein the capacity values of the sensor capacitors will be reduced by reactions occurring in the biological materi-al.
Besides, also the measurement capacitors may be realized on the same chip.
In the following, the invention will be explained in more detail by way of the preferred exemplary embodiment illustrated in the drawings, yet without being restricted thereto. In de-tail:
Fig. 1 schematically shows an electronic biosensor arrange-ment according to the invention, including a following data-pro-cessing unit;
Fig. 2A shows a schematic circuit diagram of the individual biosensors and sensor capacitors with the measuring circuit as-signed thereto;
Fig. 2B shows a diagram of some of the voltage signals oc-curring in the circuit operation according to Fig. 2;
Fig. 3A shows a schematic circuit diagram of the control lo-gic of the measuring circuit according to Fig. 2A;
Fig. 3B shows different pulse signals of said control logic;
and Fig. 3C shows, in a schematic representation, a number of biosensors comprised of sensor electrodes, including switching means assigned thereto, wherein also the reference capacitor comprised of comparable electrodes is shown with switching means assigned thereto.
In Fig. 1, the basic principle of the present electronic bi-osensor arrangement 1 is schematically illustrated, wherein a sensor-electrode arrangement 3 is provided on a common chip 4 in a receiving region 2 (only very schematically indicated) for re-ceiving biological material; outside of the receiving region 2 for the biological material, the sensor-electrode arrangement 3 furthermore provides a reference capacitor Cref and measurement capacitors Co to C5. In the receiving region 2 for the biological material a number (e.g. 10,000) of sensor capacitors Csensor,i, wherein i= 1, 2, 3... N, is defined.
Here, the individual sensor capacitors Csensor,i as well as the reference capacitor Cref (and likewise the measuring capacitors Co to C5) are formed by electrodes intermeshing in a comb-like man-ner, as can be schematically seen from Fig. 3C. These capacitors Cref and/or Csensor,i are assigned to switching means S1, S2,1, S2,2, ... SZ,i ... SZ,N, and S3,0, S3,1 . . . S3,5, respectively, which may be seen from Figs. 2A and 3C, and which are specifically formed by electronic switching means, e.g., directly in the re-gion of the semiconductor chip 4, with a bus leading to the in-dividual switching means S1 to S3 of the measuring circuit 5 for control purposes. The switching means S1r S2,i and S3,0 to S3,5 are driven by a control logic 8 of the measuring circuit 5, wherein the switching means S1 and S2, depending on the respective meas-urement phase, in operation, establish an electric connection to a measurement voltage Vdrive or to earth, but they may optionally also remain floating (with the latter case relating to the switches Sz,i for the biosensors, i.e. for the sensor capacitors Csensor,i) = The switching means S2,i (wherein i = 1 ... N) and S1 are driven via a control bus 6 and a control line 7, cf. Fig. 2A. A
further, control bus 9 of the control logic 8 leads to the switching means S3,0 to S3,51 which selectively connect the binary-weighted measurement capacitors Co to C5 to a voltage source Varray or to earth. The other side of the measurement capacitors Co to C5 leads to a node A, to which also the reference capacitor Cref as well as the sensor capacitors Csensor,i are connected. This node A is connected to the one input, i.e. the inverted input (-), of a differential amplifier 10 via a resistance R,omp, with the oth-er, non-inverted input (+) thereof being connected to earth, wherein Fig. 2A additionally shows in a schematic fashion that an offset voltage Vos may possibly occur during operation. To be able to compensate for this offset voltage Vos after an initial-izing phase I(cf. Fig. 3B), the differential amplifier 10 has a feedback branch 11 including a switch So. This electronic switch So is again driven by the control logic 8 via a control line 12, this taking place in said initializing phase I prior to the measurement and approximation phase proper, which constitutes a phase of successive approximation, wherein this measurement phase M for one of the sensor capacitors Csensor,i as well as the initializing phase I are illustrated in Fig. 3B.
For the sake of completeness, in the circuit according to Fig. 2A, also a parasitic capacity present in chip 4 (cf. Fig.
1) is illustrated by a capacitor (CParasit) = Furthermore, a re-gister 13 is shown which is provided for storing the bit order per sensor capacitor Csensor,if which forms the measurement result, wherein said measurement results will then be received by a data-processing unit 14 (cf. Fig. 1) for further processing.
The register 13 is indexed for each sensor capacitor, cf.
the pulse signal "ready" in Figs. 2A and 2B, wherein said pulses are used as clock pulses for the register 13. At an input D of the register 13, the switch signals are supplied which are provided for the switching means S3,0 to S35 5 and depend on the comparison of potentials of the measurement capacitors Co to C5 with the respective sensor capacitor Csensor,if wherein this depend-ency manifests itself in the output signal (comp in) of the dif-ferential amplifier 10, said output signal being supplied at the respectively designated input comp_in of the control logic 8.
Furthermore, Fig. 2B illustrates a control signal Vb applied to the switching mearis Sl, and a control signal V6 applied to the switching means S2;i, as well as a clock signal Vclk. Here, it fol-lows that the switching means S2,; may also remain floating, with the exception of the specific switching means for the currently measured sensor capacitor Csensor,i, as will be explained in more detail below.
Fig. 3A shows the control bus 6 with individual lines for individual switch signals Vsensorl,2,3... , for the individual sensor capacitors Csensor,l ..., in more detail. These switch signals, as well as the switch signal VB supplied for the switching means.Sl for the reference capacitor Cref via the control line 7, and a further switch signal for the individual bit stages bit5, bit4, bit3, bit2, bitl and bitO in correspondence to the measurement capacitors C5, C91 ... C1, Co, are generated by an initializing and sensor-selecting circuit 15 within the control logic 8. The individual bit stages bitO to bit5 are supplied with the output signal of the differential amplifier 10, i.e. the signal comp_iri, so as to logically combine the same with the pulse sig-nal VA, cf. also Fig. 3B, with the result thereof being used for generating the control signals (on the bus 9, indicated by ar-rows) for the switching means S3,5, S3,4 ... S3,0 assigned to the measurement capacitors C5, CQ... Co. In Fig. 3A, these control signals are shown directly below the individual outputs to the individual switching means S3,5 etc.; here, the horizontal dotted line also indicates that the control signal may also remain high, as a function of the signal output and the output of the differential amplifier, i.e. as a function of the result of the comparison currently being made, as will be explained in further detail below.
Fig. 3B shows in combination the individual pulse signals Vc1k (clocked signal), VB (control signal for the switching means S1 on the control line 7, as well as to the switch So on the con-trol line 12), VA (pulse signal to the bit stages bit5 to bitO) as well as the individual switch signals S3,i, wherein, as regards the initializing phase I preceding the measurement phase M prop-er, it is illustrated that the reference capacitor Cref is applied to the measurement voltage Vdrive according to the signal VB, and also that the feedback branch 11 of the differential amplifier is closed by the switch So, whereas the switching means S3,0 to S3,5 assigned to the measurement capacitors Co to C5 are connected to -earth.
In detail, as regards the pulse signals described, it shall be additionally mentioned that the clock signal V,lk is assumed to trigger all respectiveprocesses with its rising edges. However, the sampling of the output signal comp in of the differential amplifier 10 is effected in case of a rising edge of the inver-ted clock signal, i.e. after the respective bit has had a half period of time for charging. As mentioned above, the outputs of the bit stages bit5 to bitO constitute the control signals for the switching means S3,s, ... S3,0, which are also referred to as SAR switches (SAR - successive approximation register). As men-tioned above, in Fig. 3A, these switch,signals are shown below the bit stages bit5 to bitO, wherein it is also illustrated that they rise successively. The possibility of keeping the signal in the high state, as is illustrated in dotted lines, will result if the respective approximation sum at the input of the compar-ator, i.e. at the input (-) of the differential amplifier 10, is negative.
The signal VA triggers the measurement and approximation phase proper. When the reference capacitor Cref has been charged, with all measurement capacitors C5 to Co having to be connected to earth during said charging time (initializing phase I), this signal VA, when rising, triggers the output of the first array bit, the MSB bit, in the present case the bit no. 5. Here, it should be mentioned that a design with six bits serves as an ex-ample and reflects a good compromise between the number of weighted capacitors Co, C1, ... and the limit of detection. Cer-tainly, also more (or less) measurement capacitors, e.g. 12 measurement capacitors C, may be provided.
During the initializing phase I the signal VB is high prior to the meas.urement proper and urges the switching means S1 for the reference capacitor CLef into its closed state. Likewise, the signal V. causes the switch So to close in the feedback branch 11 of the differential amplifier 10 so as to ensure a uniform feed-back during this phase. The number of clock periods during which the the signal VB is high depends on the decay time necessary after the approximation phase.
The output signal compin of the differential amplifier 10 is of interest when the latter acts as a comparator, i.e. during the measurement or approximation phase proper. The signal comp_in will be positive if the total sum of measurement capa-cities switched on is higher at the respective point of time than the absolute capacity difference.between the sensor capa-citor Csensor,i and. the reference capacitor Cref. However, the signal comp_in will be negative if several measurement capacitors have to be switched on. It is assumed that this output signal compin of the differential amplifier will be rising quickly enough after a respective bit (on the bus 9) has been switched on. This is why attention has to be paid to a quick mode of operation of the differential amplifier 10.
The circuit described allows for sampling rates of up to 15 mega samples per second to be reached at low resolutions, wherein the circuit with the SAR converter 16, which has been realized by the measurement capacitors Co to C5 as well as by a starting capacitor Cmin including the control logic 8 assigned thereto and the differential amplifier or comparator 10, may be implemented with an extraordinary compact circuit with low power input.
During operation, the sensor capacitors Csensor,l == - Csensor,n are multiplexed, wherein, for determining one of these sensor capa-citors, Csensor,ir the measurement capacitors Cn, with n = 0 to 5, are connected one after the other so as to perform SAR al-gorithm. All these capacitors are connected to the node A which is located above the resistance Rcomp at the (-)- input of the differential amplifier 10 and has a high impedance when the feedback branch 11 is open, i.e. when the switch S4 is opened. A
single sensor capacitor is selected from a number of sensor ca-pacitors for each measurement circuit, and those which are cur-rently not being measured remain floating.
If the biosensor arrangement is activated during operation, the whole digital logic plus switch-driving means will be ini-tialized; at this point of time there will be no measurement yet, and there will be no information available at the output.
In this situation the individual switches have the following states:
So open S1 connected to earth S2,a11 floating S3, all connected to earth Then, 'the measurement of the first sensor capacitor, e.g. Csensor,li will be started. As mentioned above, two phases each are provided, i.e. the initializing phase I during which the refer-ence capacitor Crefhaving a fixed capacity value is charged, and the measurement phase M proper during which the respective sensor capacitor Csensor,i is being switched on; thereafter, the measurement capacitors C5 to Co will be connected in the SAR ar-ray. The positions of the switches for the initializing phase during measurement, e.g. of the sensor capacitor Csensor, i ~ are as follows:
So closed S1 connected to Vdrive S2,1 connected to earth Sz,,,Iili floating S3,x connected to earth The addition "x" indicates that in each case all switches are involved, whereas the addition "/{1}" indicates that all switches S2 are floating, except the switch for the first sensor capacitor Csensor, l=
During the measurement phase M proper the states of the in-dividual switches are as follows:
So open S1 connected to earth S2,1 connected to Vdrive S2,%/i11 floating S3,n cn= 5,4, ... 0) these switches S3,n are actuated one after the other and optionally kept connected, depending on the output comp_in of the comparator, as already mentioned above, cf. also the switch or control signals below the control logic 8 in Fig.
3A.
As mentioned above, during the initializing phase I the feedback switch So is closed and the reference capacitor Cref is connected to the measurement voltage Vdrivei while at least the first sensor capacitor Csensor,i to be measured is connected to earth, as are the measurement capacitors C,,. The differential amplifier 10 virtually forces mass potential at its (-)- input.
During this phase the resistance RcomP serves for achieving a phantom-zero frequency compensation in combination with an input capacity. During the measurement phase or approximation phase M
following thereupon the.resistance R,omp will have practically no effect since the current flowing through the same will be negli-gible.
During this approximation or measurement phase M the ampli-fication is high at the operation frequency of the differential amplifier 10 so as to amplify the smallest input voltages during the last step of the approximation to a logic "high" or "low".
The smaller the sensor capacitors to be measured, the higher the amplification of the amplifier or comparator 10 has to be.
During this measurement phase M one sensor capacitor after the other is connected to the measurement voltage Vdrivei wherein the voltage at the node A will then become proportional to the capacity difference. The algorithm of the successive approxima-tion is used for converging to a digital measurement signal rep-resenting said capacity difference. To this end, as mentioned above, the measurement capacitors are connected to the voltage Varra,,, starting with the capacitor C5 of the highest capacity value (32 Cmin), and are kept connected, if occasion arises. This depends on the resulting sign of the voltage in the node A.
In the following, this will be explained in more detail by way of a specific example: it is assumed that all sensor capa-citors have a nominal value (without biological reaction) of about 10pF. Furthermore, it is assumed that the first sensor ca-pacitor Csensor,if influenced by a biological reaction, has de-creased to a value of 8.58pF. This change in the capacity is substantial enough to be representative for the process and may thus be detected.
The reference capacitor Cref is a capacitor on which no reac-tion takes place; its capacity value is lOpF. Correspondingly to a binary order, the measurement capacitors Co to C5 have the ca-pacity values 50fF, 100fF, 200fF, 400fF, 800fF and 1.6pF. During measurement, the difference of the capacity of the sensor capa-citor Csensor,l and the reference capacitor Cref will be stored at the node A. The charge will thus be equivalent to a capacity of 8.58pF - lOpF =-1.42pF. In the following, the measurement capa-citors CS to Co will one after the other be connected to the voltage VarraY, as described above, so as to find out whether.the equivalent charge is positive or negative. In the present ex-ample, this results in the following table of values:
C5,..0 node A equivalent compin. binary output.
remaining capacity 1.6pF -1.42pF +0.18pF negative 0 800fF -1.42pF -0.62pF positive 1 400fF -0.62pF -0.22pF positive 1 200fF -0.22pF -0.02pF positive 1 100fF -0.02pF +0.08pF negative 0 50fF -0.02pF +0.03pF negative 0 The following result will be obtained:
6-bit output: 011100; corresponding measurement result =
8.6pF (in this example, the resolution is 50fF).
The measurement result of 8.6pF results from the difference of the capacity of the reference capacitor (lOpF) minus the ca-pacity measured (1.4pF, corresponding to the 6-bit output 011100). -As mentioned above, the measurement capacitors Co to C5 are preferably also realized on the chip 4 so as to achieve a par-ticularly compact design of the whole biosensor arrangement 1.
In Fig. 2A, another capacitor Cmin is shown which is provided in parallel to the measurement capacitors Co to C5 proper, which is always switched between the node A and earth and which is of no function for the sensor activity and only advantageous for a calibration and testing procedure prior to putting the sensor into operation. The capacitor Cmin may also be omitted.
If, during a measurement and as a function of the sign of the voltage at the node A, i.e. at the (-)- input of the differ-ential amplifier 10, the respective measurement capacitor Cn (wherein n = 0... 5, in the example shown) is either kept con-nected to the supply voltage Varrayr thus contributing to the cur-rent approximation sum, or is connected to earth, thus being inactive for the further conversion, the switching means S3,n will be driven by the control logic 8, as above described, so as to implement the above modified SAR algorithm. During this conver-sion, the voltage V at the node A may generally be as follows:
V_ `Csensor, 1-Crej/ V drive+C arrayVarray Csensor,l+Cref+
C/wrasii+C'array,TOT ' .. -In this relation, Catray,TOT represents the total capacity of the measurement capacitor, whereas Carray indicates the linear com-bination of those weighed measurement capacitors that represent the approximation until the just-described point. After the com-plete measurement circuit, on the bus 9 driving the switching means S3,n those bits are present in series that produce the capa-city difference measured (in the present example: 1.4pF).
The total capacity of the measurement capacitors, Carray,TOTf is determined by the number of bits N (in the present example, N
= 6), as it is desired for the accuracy of measurement and res-olution, as well as by the smallest capacitor which can be real-ized using a certain technology in a manner still providing suf-ficient accuracy. The value of this total capacity is Carray,TOT -21+1'Cmin= If desired, the maximum possible difference, (absolute) value =1 Csensor,i - Cref I, be adapted to the complete region of the SAR converter, i.e. according to v - I C.senanrI i - Crejl v array 2N+1 drire min As a consequence of the mode of two-phase operation, includ-ing the initializing phase I and the measurement phase M proper, the offset sensitivity of the arrangement can be substantially reduced. During the initializing phase I the offset of the dif-ferential amplifier 10 is stored in the node A and, then, it will be eliminated by the difference amplification during the approximation phase (measurement phase M). Assuming that the bi-osensor arrangement 1 is operated at high switching frequencies, the 1/f noise may be considered offset.
When the switch So is being opened in the feedback branch 11, the charge is injected in the node A with high impedance so as to cause a small offset voltage in this node. This offset voltage can only be detected if it is present with a magnitude as is the voltage V at the node A, which occurs when the LSB ca-pacitor (capacitor Co) is being measured during the measurement phase. This may be the case if small biosensors are measured since Carray,TOT will not change and the measurement voltage Vdrive has an upper limit as regards the supply voltage. If necessary, the offsetvoltage may also be compensated for in a digital man-ner by a calibration step.
8.6pF (in this example, the resolution is 50fF).
The measurement result of 8.6pF results from the difference of the capacity of the reference capacitor (lOpF) minus the ca-pacity measured (1.4pF, corresponding to the 6-bit output 011100). -As mentioned above, the measurement capacitors Co to C5 are preferably also realized on the chip 4 so as to achieve a par-ticularly compact design of the whole biosensor arrangement 1.
In Fig. 2A, another capacitor Cmin is shown which is provided in parallel to the measurement capacitors Co to C5 proper, which is always switched between the node A and earth and which is of no function for the sensor activity and only advantageous for a calibration and testing procedure prior to putting the sensor into operation. The capacitor Cmin may also be omitted.
If, during a measurement and as a function of the sign of the voltage at the node A, i.e. at the (-)- input of the differ-ential amplifier 10, the respective measurement capacitor Cn (wherein n = 0... 5, in the example shown) is either kept con-nected to the supply voltage Varrayr thus contributing to the cur-rent approximation sum, or is connected to earth, thus being inactive for the further conversion, the switching means S3,n will be driven by the control logic 8, as above described, so as to implement the above modified SAR algorithm. During this conver-sion, the voltage V at the node A may generally be as follows:
V_ `Csensor, 1-Crej/ V drive+C arrayVarray Csensor,l+Cref+
C/wrasii+C'array,TOT ' .. -In this relation, Catray,TOT represents the total capacity of the measurement capacitor, whereas Carray indicates the linear com-bination of those weighed measurement capacitors that represent the approximation until the just-described point. After the com-plete measurement circuit, on the bus 9 driving the switching means S3,n those bits are present in series that produce the capa-city difference measured (in the present example: 1.4pF).
The total capacity of the measurement capacitors, Carray,TOTf is determined by the number of bits N (in the present example, N
= 6), as it is desired for the accuracy of measurement and res-olution, as well as by the smallest capacitor which can be real-ized using a certain technology in a manner still providing suf-ficient accuracy. The value of this total capacity is Carray,TOT -21+1'Cmin= If desired, the maximum possible difference, (absolute) value =1 Csensor,i - Cref I, be adapted to the complete region of the SAR converter, i.e. according to v - I C.senanrI i - Crejl v array 2N+1 drire min As a consequence of the mode of two-phase operation, includ-ing the initializing phase I and the measurement phase M proper, the offset sensitivity of the arrangement can be substantially reduced. During the initializing phase I the offset of the dif-ferential amplifier 10 is stored in the node A and, then, it will be eliminated by the difference amplification during the approximation phase (measurement phase M). Assuming that the bi-osensor arrangement 1 is operated at high switching frequencies, the 1/f noise may be considered offset.
When the switch So is being opened in the feedback branch 11, the charge is injected in the node A with high impedance so as to cause a small offset voltage in this node. This offset voltage can only be detected if it is present with a magnitude as is the voltage V at the node A, which occurs when the LSB ca-pacitor (capacitor Co) is being measured during the measurement phase. This may be the case if small biosensors are measured since Carray,TOT will not change and the measurement voltage Vdrive has an upper limit as regards the supply voltage. If necessary, the offsetvoltage may also be compensated for in a digital man-ner by a calibration step.
Claims (5)
1. An electronic biosensor arrangement (1), comprising a re-ceiving region (2) for biological material, a sensor-electrode arrangement (3) with sensor electrodes (3a, 3b) which intermesh in a comb-like manner being associated therewith, to which sensor electrodes (3a, 3b) a measuring circuit is connectable for measuring an electrical measurement parameter on the sensor electrodes (3a 3b) and influenced by biological material, char-acterized in that the sensor electrodes (3a, 3b) form a plural-ity of sensor capacitors (C sensor,1 to C sensor,N), with which electronic switching means (S2,1 to S2,N) driven by a control logic (8) are associated for connecting one side of the sensor capa-citors to earth or to a voltage source carrying a measurement voltage (V drive), respectively, comprising a reference capacitor (C ref) with which a switching means (S1) driven by the control lo-gic (8) and arranged for optionally connecting one side to earth or to the measurement voltage (V drive) is likewise associated, wherein the other sides of the capacitors (C ref, C sensor,l - C sensor,N) are combined in a node (A), which is connected to an input (-) of a differential amplifier (10), wherein measurement capacitors (C0 to C5) having binary-weighted capacities (C min to 32C min) are furthermore connected to the node (A) for forming an SAR compar-ison unit, with the measurement capacitors being also select-ively connectable via switching means (S3,0 to S3,5) driven by the control logic (8) to a voltage source (V array) or to earth for the purpose of providing charge-difference for SAR conversion.
2. The biosensor arrangement according to claim 1, character-ized in that the differential amplifier (10) has a feedback branch (11) extending from its output to the one input (-), wherein a switch (S0) driven by the control logic (8) is arranged in said feedback branch (11) for closing and opening the latter, and wherein the differential amplifier (10) acts as a comparator during the SAR approximation phase (M) when the feedback branch (11) is opened, whereas when the feedback branch (11) is closed, the differential amplifier (10), during an initializing phase (I), feedbacks an amplifier-offset voltage (V os) to the node (A) for a subsequent compensation.
3. The biosensor arrangement according to claim 2, character-ized in that after the initializing phase, the switch (S0) provided in the feedback branch (11) will be driven by the con-trol logic (8) either simultaneously with the switching of the switching means (S1) associated with the reference capacitor (C ref) or directly before said switching means (S1) will be switched so as to connect the reference capacitor (C ref) to earth instead of to the measurement voltage (V drive).
4. The biosensor arrangement according to claim 2 or 3, charac-terized in that a resistance (R comp) is connected to the one input (-) of the differential amplifier (10) for frequency damping in connection with an input capacity of the differential amplifier during the initializing phase (I).
5. The biosensor arrangement according to any one of claims 1 to 4, characterized in that the capacity value of the reference capacitor (C ref) is selected so as to equate to the highest-pos-sible capacity value of the individual sensor capacitors (C sensor,i), wherein the capacity values of the sensor capacitors will be reduced by reactions occurring in the biological materi-al.
Applications Claiming Priority (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| AT0083006A AT503742B8 (en) | 2006-05-15 | 2006-05-15 | ELECTRONIC BIOSENSOR ARRANGEMENT |
| ATA830/2006 | 2006-05-15 | ||
| PCT/AT2007/000233 WO2007131255A1 (en) | 2006-05-15 | 2007-05-15 | Electronic biosensor arrangement |
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| Publication Number | Publication Date |
|---|---|
| CA2652407A1 true CA2652407A1 (en) | 2007-11-22 |
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ID=38268834
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| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002652407A Abandoned CA2652407A1 (en) | 2006-05-15 | 2007-05-15 | Electronic biosensor arrangement |
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| Country | Link |
|---|---|
| US (1) | US8043563B2 (en) |
| EP (1) | EP2018438A1 (en) |
| KR (1) | KR20090056932A (en) |
| CN (1) | CN101512015A (en) |
| AT (1) | AT503742B8 (en) |
| CA (1) | CA2652407A1 (en) |
| WO (1) | WO2007131255A1 (en) |
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| CA3074169A1 (en) * | 2017-09-29 | 2019-04-04 | Boehringer Ingelheim Vetmedica Gmbh | Testing and calibration of a circuit arrangement |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4860232A (en) * | 1987-04-22 | 1989-08-22 | Massachusetts Institute Of Technology | Digital technique for precise measurement of variable capacitance |
| US5027814A (en) * | 1989-05-19 | 1991-07-02 | Ventritex, Inc. | Implantable medical device employing an improved waveform digitization network |
| US5235333A (en) * | 1992-03-05 | 1993-08-10 | Burr-Brown Corporation | Hysteresis-insensitive single-comparator successive approximation analog-to-digital converter |
| US5920275A (en) * | 1996-09-09 | 1999-07-06 | Iowa State University Research Foundation, Inc. | Analog-to-digital converter using weighted capacitor array and interpolating comparator |
| US6225678B1 (en) * | 1998-12-23 | 2001-05-01 | Microchip Technology Incorporated | Layout technique for a matching capacitor array using a continuous top electrode |
| US20030226768A1 (en) * | 2000-03-30 | 2003-12-11 | Franz Hoffman | Method for detecting macromolecular biopolymers by means of an electrode arrangement |
| US7413859B2 (en) * | 2001-11-14 | 2008-08-19 | Siemens Aktiengesellschaft | Method and biosensors for detecting macromolecular biopolymers |
| DE10204652B4 (en) * | 2002-02-05 | 2004-07-22 | Infineon Technologies Ag | Circuit arrangement, electrochemical sensor, sensor arrangement and method for processing a current signal provided via a sensor electrode |
| JP2003258639A (en) * | 2002-02-27 | 2003-09-12 | Nec Microsystems Ltd | Analog-to-digital converter |
| US6600437B1 (en) * | 2002-04-01 | 2003-07-29 | Stmicroelectronics S.R.L. | High resolution, high speed, low power switched capacitor digital to analog converter |
| DE10224567B4 (en) * | 2002-06-03 | 2014-10-23 | Boehringer Ingelheim Vetmedica Gmbh | Sensor arrangement and method for operating a sensor arrangement |
| US6714151B2 (en) * | 2002-06-21 | 2004-03-30 | Fujitsu Limited | A/D converter |
| DE10259819B4 (en) | 2002-12-19 | 2006-07-13 | Siemens Ag | Method for PCR amplification and detection of nucleotide sequences |
| DE10328136A1 (en) * | 2003-06-23 | 2005-01-27 | Infineon Technologies Ag | Sensor element, sensor array and method for detecting particles possibly contained in an analyte |
| DE102004045210A1 (en) | 2004-09-17 | 2006-04-06 | Infineon Technologies Ag | Sensor arrangement and method for determining a sensor event |
-
2006
- 2006-05-15 AT AT0083006A patent/AT503742B8/en not_active IP Right Cessation
-
2007
- 2007-05-15 CN CNA2007800235363A patent/CN101512015A/en active Pending
- 2007-05-15 KR KR1020087029901A patent/KR20090056932A/en not_active Application Discontinuation
- 2007-05-15 CA CA002652407A patent/CA2652407A1/en not_active Abandoned
- 2007-05-15 WO PCT/AT2007/000233 patent/WO2007131255A1/en active Application Filing
- 2007-05-15 EP EP07718445A patent/EP2018438A1/en not_active Withdrawn
- 2007-05-15 US US12/300,978 patent/US8043563B2/en not_active Expired - Fee Related
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| Publication number | Publication date |
|---|---|
| KR20090056932A (en) | 2009-06-03 |
| US20090206847A1 (en) | 2009-08-20 |
| AT503742B8 (en) | 2011-08-15 |
| WO2007131255A1 (en) | 2007-11-22 |
| WO2007131255A8 (en) | 2008-02-28 |
| AT503742A1 (en) | 2007-12-15 |
| AT503742B1 (en) | 2011-06-15 |
| US8043563B2 (en) | 2011-10-25 |
| EP2018438A1 (en) | 2009-01-28 |
| CN101512015A (en) | 2009-08-19 |
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