CN104811038A - 具有动态阈值的磁滞降压变换器 - Google Patents

具有动态阈值的磁滞降压变换器 Download PDF

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CN104811038A
CN104811038A CN201510225358.9A CN201510225358A CN104811038A CN 104811038 A CN104811038 A CN 104811038A CN 201510225358 A CN201510225358 A CN 201510225358A CN 104811038 A CN104811038 A CN 104811038A
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voltage adjuster
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CN104811038B (zh
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约翰·L·梅安森
丁磊
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Cirrus Logic Inc
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/1563—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators without using an external clock
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H02M1/0019—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters the disturbance parameters being load current fluctuations
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0032—Control circuits allowing low power mode operation, e.g. in standby mode
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Abstract

磁滞降压变换器可改善调节控制，尤其可改善降压变换器的备用操作。比较电路(Kl,K2)将降压变换器的输出电压(Vout)与由变换器的输出电流指示(+VILOAD)产生的波形(VLOW)作比较，从而使变换器的接通时间随输出电流需量的增加而提前。合成作用预期：输出电压(Vout)会由于电流增大而降低，由此避免使输出电压(Vout)的偏移小于脉动电压最小值。变换器的断开时间是由限制脉动电压最大值的上限阈值(VHIGH)控制的。输出电流指示可以是对输出电流的量度，或是由输入电压(Vin)和输出电压(Vout)的波形计算得到的动态值。

Description

具有动态阈值的磁滞降压变换器
本申请是中国专利申请200980133008.2(母案)的分案申请，母案的申请日是2009年6月19日，发明名称是“具有动态阀值的磁滞降压变换器”。母案的国际申请号是PCT/US2009/048016。
技术领域
本发明通常涉及磁滞降压变换器控制方案，尤其涉及一种使用输出电流指示来动态调节接通定时的降压变换器控制电路。
背景技术
在低输出电流电平，脉宽调制器(pulse width modulator,PWM)控制器以及其他类型的在高负载条件下输送高电流电平的切换功率调节器是低效的。由于脉宽对于低输出电流电平变得非常窄，变换器的功耗受用于操作切换电路和控制/传感电路且通常不随负载电流需量变化的功率支配，由此使变换器在低需量条件下非常低效。在备用低功率模式下经常使用可选的控制器模式操作变换器，由此虽不能获得完全的变换器动态特性，但可保持最小输出电压，以在低负载电流需量时提供所需的电源电压。由于可在负载需量基础上任意降低脉冲频率，脉冲频率调制器(Pulse-frequency modulator,PFM)电路经常用于低功率操作模式。已使用输出电压被保持在两个预设定点之间的磁滞控制电路提供这类低功率操作模式。由于磁滞变换器只在输出电压小于容许极限时起作用，因而它们具有宽的动态范围和可能的低功耗。
此外，在复杂性或PWM操作所需功率不符合要求的低功率应用中，由于控制电路本身可被完全置于备用模式下，且下限电压传感电路是唯一需要操作的电路，因而磁滞控制器有时用于对所有输出电流电平提供电源控制算法。当必须提高输出电压时，下限电压传感电路的输出于是可起动变换器的剩余者。另外，在PWM变换器的瞬态响应不能足够快地对负载瞬态作出响应的任何应用中，磁滞变换器也用于对变化的负载条件作出快速响应。
在典型的磁滞变换器中，当输出电压小于下限阈值时可提供恒定宽度脉冲，由此向输出电容器内注入电荷，从而使输出电压升高预定的量。然而，如果输出电流或输入电压条件发生变化，由于恒定宽度脉冲不能对不同电平的负载电流或输入电压作出响应，这种变换器会产生不符合要求的脉动水平。在其他类型的磁滞变换器中，对输入电压和其他电源条件进行监测，并对输出脉宽进行控制，以将脉动水平控制在比恒定宽度控制器可提供的更高程度。
然而，上述磁滞控制器、负载瞬态或输入压降均可由于变换器需要响应时间而引起下限阈值的负尖峰。在恒定宽度变换器中，可能需要一些脉冲使输出电压恢复，且在宽度受控磁滞变换器中存在初始负尖峰，这些负尖峰由已被触发的脉冲进行校正。
因此，希望提供一种通过控制负尖峰减小磁滞变换器中脉动的电源电路和控制方法。
发明内容
上述控制磁滞变换器中负尖峰的目的可通过一种降压切换电压调节器电路以及一种用于操作该降压切换电压调节器电路的方法实现。
通过预期由于负载电流变化而产生的脉动的大小，降压切换电压调节器电路可改善脉动控制。该电路可以是在切换调节器的备用模式下有源的控制电路，例如在较高电流输出需量条件下使用PWM控制模式、且在较低电流输出需量条件下进入备用模式的PWM调节器。比较电路将变换器的输出电压与由变换器输出电流指示产生的波形作比较，从而使变换器的接通时间随输出电流需量的增加而提前，且调节电压下限，以避免使负尖峰低于指定的脉动电压下限。变换器的断开时间由限制脉动电压最大值的上限阈值控制。输出电流指示可以是对输出电流的量度，或可以是由输入电压和输出电压波形计算得到的值。
从下文尤其是对本发明优选实施方式的说明，并结合附图，显而易见本发明的前述以及其他目的、特征和优点。
附图说明
图1A-1B是本发明实施方式所述功率切换电路的方框图。
图2A是图1A中控制电路10A的简化示意图。
图2B是图1B中控制电路10B的简化示意图。
图3A是显示在不连续传导模式(discontinuous conductionmode,DCM)下在图2A中的阈值产生器和控制电路20A内、以及在图2B中的阈值产生器和控制电路20B内所进行的计算的信号波形图。
图3B是显示在连续传导模式(continuous conduction mode,CCM)下在图2A中的阈值产生器和控制电路20A内、以及在图2B中的阈值产生器和控制电路20B内所进行的计算的信号波形图。
图4A-4B是显示图1A-1B和图2A-2B中所示电路内信号的信号波形图。
具体实施方式
本发明包括用于对降压切换电压调节器进行控制的电路和方法，其中根据流经负载的输出电流的指示通过控制接通阈值而避免脉动负尖峰。可通过直接测量输出电流提供输出电流指示，或如下文所述，可由输出电压波形和输入电压值计算得到输出电流指示。
图1A显示本发明的一实施方式所述的降压切换电压调节器电路。当使用控制信号/sa起动晶体管P1时，一控制电路即控制器10A向切换电路SWA提供栅极驱动信号，且该切换电路SWA将电感器L1串联耦合在输入电压源VIN与输出端子VOUT之间。当使用控制信号sb起动晶体管N1时，切换电路SWA将电感器L1并联耦合在输出端子VOUT与公共返回路径(地面)之间，且该公共返回路径(地面)与输入电压源VIN和输出端子VOUT相关。输出电容器C1过滤降压切换电压调节器电路的输出，以使在输出端子VOUT处产生的电压除脉动电压外大体上保持恒定。在低需量条件下，即在向负载ZL供应低负载电流IL的条件下，控制器10A在不连续传导模式(discontinuous conduction mode,DCM)下工作，即它首先起动晶体管P1通过电感器Ll向输出电容器C1充电，然后停用晶体管P1，并起动晶体管N1将储存在电感器Ll上的能量释放给电容器，最后停用晶体管N1直到输出端子VOUT的电压小于阈值大小。在较高电平的负载电流IL下，控制器10A可在连续传导模式(continuous conduction mode,CCM)下用作磁滞变换器，或可直接转换为另一CCM控制算法，例如脉宽调制(pulse-width modulation,PWM)。即使对于变换器的磁滞操作实施CCM操作，在超出使用磁滞CCM操作时的负载电流IL的范围之后，也可起动PWM或其他操作模式。
在本发明的降压切换电压调节器中，用于确定晶体管P1的接通时间的阈值大小是一种时变波形，该时变波形由从输出端子VOUT向负载提供的输出电流指示、以及由从电压源VIN提供的输入电压大小产生。由此，当输出电流增加或输入电压降低时，晶体管P1的起动时间提前；且当输出电流减小或输入电压升高时，晶体管P1的起动时间推后。从输出电流指示计算得到阈值大小，该输出电流指示可以是对由感应电阻器Rs产生的输出电流的量度；由此提供不同于与负载电流I0成比例的输出端子VOUT电压的电压+VILOAD。(为说明起见，VOUT也被指定为-VILOAD。)控制器10A由电压源VIN提供的输入电压以及输出端子VOUT的电压计算晶体管P1的断开时间，该断开时间也是晶体管N1的接通时间，以使在输出端子VOUT处的脉动电压不超过指定的最大值。最后，晶体管N1的断开时间可由通过电感器L向电容器C1提供的电流IL控制，该电流IL由电阻器RL两端的电压(+/-VIL)测得，且达到零或稍负的值，以确保在电感器L中未储存剩余能量。
图1B显示本发明的另一实施方式所述的降压切换电压调节器。图1B中的降压切换电压调节器与图1A中的降压切换电压调节器类似，因此下文将只描述它们之间的差异。此外，图1B中降压切换电压调节器的各种特征可作为图1A中降压切换电压调节器的特征选项，反之亦然。切换电路SWB使用两个N型晶体管N2和N1，且接收来自控制器10B的栅极控制信号sa和sb。N沟道对也可用于图1A中的降压切换电压调节器，且栅极控制信号/sa的极性发生适当的变化。控制器10B只接收两个控制输入值：输入源VIN的电压、以及输出端子VOUT的电压。控制器10B根据两个控制输入信号值(Vout,Vin)执行所有切换控制，以产生栅极控制信号sa和sb，这两个控制输入信号值(Vout,Vin)由各自的输入源VIN和输出端子VOUT提供。可使用诸如图1A中切换电路SWA的P-N切换级，且使栅极控制信号sa的极性发生适当变化。由于输出电流与输出端子VOUT的电压波形和电压源VIN的电压有关，可由两个输入控制信号值(Vout,Vin)确定晶体管N1的接通时间，如下文详述。
图2A显示图1A中控制器10A的细节。当负载电流I0小于阈值时，或者当经由外部控制信号将PWM控制器24置于备用模式下时，采用由PWM控制器24提供的控制信号模式起动一对三态缓冲器26A和26B。当控制信号模式有效时，PWM 24的栅极控制输出端被置于高阻抗状态下，以使通过在控制器10A内的电路平衡实现的磁滞控制器提供栅极控制输出信号/sa和sb。阈值产生器和控制电路20A向比较器Kl提供用于设定阈值大小(电压信号VLOW)的阈值电压，在小于该阈值大小时输入控制信号Vout将起动开始信号，该开始信号通过起动触发器22A的设置输入而触发栅极控制信号/sa的起始缘。当输入控制信号Vout的大小上升超过另一阈值电压VHIGH时，另一比较器K2起动触发器22A的重设输入、以及触发器22B的设置输入，由此触发栅极控制信号/sa的后缘、以及栅极控制信号sb的前缘。当电感器电流IL小于零或为稍负的值时，阈值产生器和控制电路20A也向触发器22B的重设输入提供控制信号，以触发栅极控制信号sb的后缘。
图2B显示图1B中控制器10B的细节。控制器10B与图2A中的控制器10A类似，因此下文将只描述它们之间的差异。控制器10B被用作最小输入控制器的一个例子，且也可用于说明在磁滞模式下提供所有控制的控制器。然而，应了解可在诸如PWM控制器24的另一控制器类型的备用模式下使用最小输入配置，如图2A所示。如下文详述，阈值产生器和控制电路20B接收控制输入信号(Vout,Vin)，并产生由控制输入信号(Vout,Vin)计算得到的阈值电压VLOW和VHIGH。由于控制器20B不直接测量电感器电流IL，因而提供定时器28，它通常是由时钟信号操作的计数链，以对栅极控制信号sa的持续时间进行计时，用于计算栅极控制信号sb的适当宽度。
图3A显示在不连续传导模式(discontinuous conduction mode,DCM)下在图2A中的阈值产生器和控制电路20A内、以及在图2B中的阈值产生器和控制电路20B内进行的计算。通过使用图2A中控制电路20A的输出电流I0的测量值、或使用由图2B中控制电路20B的控制信号值(Vin,Vout)计算得到的输出电流I0的指示，可确定为确保输出端子VOUT上的脉动不小于指定最小值VMIN而要求的阈值电压大小VLOW的值。假设输出端子VOUT的电压是恒定的，同时栅极控制信号sa是有效的，则电感器电流IL可被近似为：
IL(t)＝(Vin-Vout)(t–t0)/L
其中L是电感器L1的感应系数，因此在时刻t1时，
IL(t1)＝I0＝(Vin-Vout)(t1-t0)/L
所以，
t1–t0＝I0L/(Vin-Vout)
假设电感器电流IL呈线性，则从时刻t0至时刻t1在输出端子VOUT上的实际压降为：
△V＝IL△t/2C
其中C是在输出端子VOUT处的总输出电容。为将在输出端子VOUT处的电压保持为大于最小电压VMIN，应在控制信号VOUT降至以下阈值水平之前起动栅极控制信号sa：
VLOW＝VMIN+ΔV＝VMIN+I0(t1–t0)/2C
因为
t1–t0＝I0L/(Vin-Vout)
在上述对恒定电感器电流IL的近似中，可由下式计算阈值大小VLOW：
VLOW＝I0 2L/2C(Vin-Vout)
所以，一旦已知电感器和电容器的值，由于此时超过阈值大小VLOW，IL(t1)＝I0，因而可由输入电压信号Vin和控制信号Vout以及负载电流IL确定阈值电压VLOW的波形。
为确定用于由VLOW＝I0 2L/2C(Vin-Vout)计算阈值大小VLOW的负载电流，可直接测量或估计负载电流。由于从时刻t0至时刻t2电容器C1两端的电压变化为VHIGH–VLOW，则电容器上电荷的增加可被表示为：
C(VHIGH–VLOW)＝[(Vin-Vout)(t2-t0)2]/2L-I0(t2-t0)
由此，电流I0可被表示为：
I0＝[(Vin-Vout)(t2-t0)]/2L-C(VHIGH–VLOW)/(t2-t0)
通过测量充电开关(例如图1A中的晶体管P1或图1B中的晶体管N2)的导通时间(例如计数时钟周期)，可计算I0并将其用作电感器电流IL的估计值，这是因为除在非常轻的负载状态下外两者差异通常很小。或者，可根据下式计算从时刻t2至时刻t5的时段内的I0：
C(VHIGH–VLOW)＝(Vin-Vout)(t4-t2)2/2L-I0(t5-t2)
由此得到：
I0＝Vout(t4-t2)2/2L(t5-t2)-C(VHIGH–VLOW)/(t5-t2)
进行另一计算，以确定当栅极控制信号sa应无效且栅极控制信号sb应有效时在输出端子VOUT上的电压的大小VHIGH。在VOUT＝VHIGH的时刻t2，输出端子VOUT上的电压为VMAX-Q(t3-t2)/C，其中Q(t3-t2)是在栅极控制信号sa无效且栅极控制信号sb有效之后增加到电容器C1上的总电荷，它等于VMAX-(IMAX–I0)(t3-t2)/2C，其中最大电流IMAX是峰值电感器电流。因此，设定：
VHIGH＝VMAX-(IMAX-I0)(t3-t2)/2C
该式将提供所需的切换时间。也可根据VMIN表达阈值大小VHIGH：
VHIGH＝VMIN+(IMAX-I0)(t2–t1)/2C
可由下两式确定峰值电流IMAX：
IMAX＝I0+(Vin–Vout)(t2–t1)/L
以及
IMAX＝I0+Vout(t3-t2)/L
它们假设输出电压大体上不变化，且电感器电流在充电和放电期间是恒定的。因此，
(Vin-Vout)(t2–t1)/L＝Vout(t3-t2)/L
由此得到：
t3-t2＝(t2–t1)(Vin-Vout)/Vout
可将上述t3-t2的表达式代入到上述VHIGH的表达式中，得到：
VHIGH＝VMAX-[(IMAX-I0)(t2-t1)(Vin-Vout)]/2CVout
可根据最小电压VMIN由阈值大小VHIGH的表达式确定IMAX-I0，得到：
IMAX-I0＝2C(VHIGH-VMIN)/(t2-t1)
最后，结合以上两个等式得到：
VHIGH＝VMAX-(VHIGH-VMIN)(Vin-Vout)/Vout
因此，
VHIGH＝VMIN+(VMAX-VMIN)Vout/Vin
可使用上述表达式在控制信号Vin和Vout的前值基础上得出或计算出作为离散值的阈值大小VHIGH，或可使用上述表达式产生用于控制上限阈值大小的连续波形。
在上述每个计算中，假设电路在DCM下工作，即其工作的结果可使储存在电感器Ll中的所有能量在时刻t0和t5时被释放。然而，在更高负载条件下，本发明的电路可在连续传导模式(continuousconduction mode,CCM)下工作，且为实现最佳操作，可改变下限阈值大小VLOW的计算方式。不过，上限阈值大小VHIGH的计算方式与上述相同。
图3B显示这种操作。由于不存在两切换晶体管(充电和放电)均断电的有效时段，在图3B的信号图中不存在时刻t4。在连续传导模式下，假设输出电压V0恒定：
dIL/dt＝(Vin–Vout)/L
在充电晶体管(例如图1A中的晶体管P1或图1B中的晶体管N2)接通之后，如果给出在接通时间时为起始非零电流值的较低电流IMIN，则：
IL(t)＝(Vin-Vout)(t–t0)/L+IMIN
由此根据在不连续传导模式下的上述定义I0＝IL(t)，得到：
t-to＝(I0 -IMIN)L/(Vin–Vout)
因此，为确保输出端子VOUT的电压不小于最小电压VMIN，应将阈值大小VLOW设定为：
VLOW＝VMIN+L(I0-IMIN)2/2C(Vin–Vout)
为确定在CCM下的负载电流，如上文中就DCM所述，可直接测量或估计负载电流。由于从时刻t0至时刻t2电容器C1两端的电压变化为VHIGH–VLOW，则电容器上增加的电荷可被表示为：
C(VHIGH-VLOW)＝(Vin-Vout)(t2-t0)2/2L-(I0-IMIN)(t2-t0)且I0-IMIN可被表示为：
I0-IMIN＝(Vin-Vout)(t2-t0)/2L-C(VHIGH-VLOW)/(t2-t0)
它与在DCM下I0的表达式相同。由于用于确定上述VLOW所需的量是I0-IMIN，可使用相同的计算方式来估计在CCM下I0-IMIN的值，该值用于估计在DCM下的I0。或者也可使用如下基于从时刻t2至时刻t4的时段的可选表达式：
I0-IMIN＝Vout(t4-t2)2/2L(t3-T2)-C(VHIGH-VLOW)/(t5-t2)
在CCM下，有可能进一步优化对阈值大小VLOW的控制。因为在从时刻t5至时刻t6的时段，从电容器C1损失的总电荷如下：
(I0-IMIN)(t6–t5)/2
所以，下限阈值大小VLOW为：
VLOW＝VMIN+(I0-IMIN)(t6–t5)/2
下限阈值大小VLOW也可被表示为：
VLOW＝VMAX-(I0-IMIN)(t5–t3)/2
在时刻t5时电感器电流IL＝IMIN，此时V0＝VLOW，且可被表示为：
IMIN＝I0-(Vin–Vout)(t6-t5)/L＝I0 _Vout(t5-t3)/L
所以，
(Vin-Vout)(t6-t5)＝Vout(t5-t3)
可结合上述关系得到：
VLOW＝VMIN+Vout(I0-IMIN)(t5-t3)/2(Vin–Vout)
＝VMAX-(I0-IMIN)(t5-t3)/2
则：
VLOW＝VMIN+(VMAX-VMIN)Vout/Vin
它与在CCM和DCM下VHIGH的表达式相同。因此，对于CCM，VLOW＝VHIGH。
图4A和图4B显示图1A和图1B中的降压切换电压调节器电路、图2A中的控制器10A、以及图1B中的控制器10B的操作，且下文将更详细地说明在控制器10A和10B中用于确定上述切换时间的计算方式。虽然只显示DCM，但该图解也适用于CCM操作。图4A显示对负载电流I0的变化作出响应的本发明磁滞变换器电路的操作。根据上述公式，当负载电流I0增加时，阈值电压VLOW增加，由此使充电晶体管渐进提前接通。图4B类似地显示对在输入源VIN处的电压下降作出响应的磁滞变换器的操作，例如使用正在放电的电池工作。放电速率被扩大，以说明输入电压的下降对阈值电压VLOW的影响；可增大阈值电压VLOW，以使充电晶体管提前接通，从而补偿输入源VIN电压的下降。
已结合优选实施方式对本发明作了详细显示和说明，但本领域技术人员应了解无需脱离本发明的精神和保护范围，即可获得前述和其他在形式方面的变化以及细节。

Claims (22)

1.一种降压切换电压调节器电路，包括：
比较电路，其用于当所述降压切换电压调节器电路的输出端的电压不超过第一阈值大小时提供第一指示，以及当所述降压切换调节器电路的所述输出端的电压超过第二阈值大小时提供第二指示，其中所述第二阈值大小大于所述第一阈值大小；
电感式存储元件；
切换电路，其用于为对所述第一指示作出响应而将所述电感式存储元件耦合在所述降压切换电压调节器电路的输入端与所述降压切换电压调节器电路的所述输出端之间，以及为对所述第二指示作出响应而将所述电感式存储元件耦合在所述降压切换电压调节器电路的所述输出端与所述降压切换电压调节器电路的所述输出端的返回路径之间；以及
控制电路，其用于产生作为时间变化电压信号的所述第一阈值大小和所述第二阈值大小，以使所述第一阈值大小随由所述降压切换电压调节器电路的所述输出端向负载提供的输出电流的增加而增加，且随所述输出电流的下降而下降，其中所述控制电路计算所述输出电流的指示，并据此改变所述第一阈值大小。
2.根据权利要求1所述的降压切换电压调节器电路，其中所述控制电路还根据所述输出电流的值改变所述第二阈值大小。
3.根据权利要求1所述的降压切换电压调节器电路，其中所述控制电路由所述降压切换电压调节器电路的所述输出端的电压、以及从所述第二指示在所述切换电路的给定周期内的起始处以及从所述第一指示在所述切换电路的下一周期内的起始处延伸的时段，计算所述输出电流的所述指示。
4.根据权利要求3所述的降压切换电压调节器电路，其中所述控制电路根据下列公式计算所述输出电流的指示：
(Vin–Vout)(t2–t0)/2L-C(VHIGH-VLOW)/(t2-t0)
其中Vin是所述降压切换调节器电路的所述输入端的电压，Vout是所述降压切换调节器电路的所述输出端的电压，C是所述降压切换调节器电路的所述输出端的电容，L是所述电感式存储元件的感应系数，t2是所述第二指示在所述给定周期内的起始处的时刻，t0是所述第一指示在所述给定周期内的起始处的时刻，VHIGH是所述第二阈值，且VLOW是所述第一阈值。
5.根据权利要求3所述的降压切换电压调节器电路，其中所述控制电路根据下列公式计算所述输出电流的指示：
Vout(t4-t2)2/2L(t5-t2)-C(VHIGH-VLOW)/(t5-t2)
其中Vout是所述降压切换调节器电路的所述输出端的电压，C是所述降压切换调节器电路的所述输出端的电容，L是所述电感式存储元件的感应系数，t2是所述第二指示在给定周期内的起始处的时刻，t5是所述第一指示在所述下一周期内的起始处的时刻，VHIGH是所述第二阈值，VLOW是所述第一阈值，且t4是所述第二指示在所述给定周期内结束的时刻。
6.根据权利要求1所述的降压切换电压调节器电路，其中所述控制电路测量所述输出电流，以提供所述输出电流的所述指示。
7.根据权利要求1所述的降压切换电压调节器电路，其中所述控制电路还根据在所述降压切换电压调节器电路的所述输入端的电压与所述降压切换电压调节器电路的所述输出端的所述电压之间的差值来根据所述第一阈值大小产生时间变化信号。
8.根据权利要求7所述的降压切换电压调节器电路，其中所述控制电路根据下列公式产生对应于所述第一阈值大小的所述时间变化信号：
I0 2L/2C(Vin–Vout)
其中Vin是所述降压切换调节器电路的所述输入端的电压，Vout是所述降压切换调节器电路的所述输出端的所述电压，C是所述降压切换调节器电路的所述输出端的电容，L是所述电感式存储元件的感应系数，且I0是所述输出电流的指示。
9.根据权利要求1所述的降压切换电压调节器电路，其中所述控制电路在连续传导模式下工作，且输出电流的所述指示是在输送给所述降压切换电压调节器电路的所述输出端的电流与在所述电感式存储元件中的最小电流之间的差值。
10.根据权利要求1所述的降压切换电压调节器电路，其中所述控制电路由所述降压切换电压调节器电路的所述输出端的所述电压、以及所述降压切换电压调节器电路的所述输入端的电压产生对应于所述第二阈值大小的所述时间变化。
11.根据权利要求10所述的降压切换电压调节器电路，其中所述控制电路根据以下公式产生对应于所述第二阈值大小的所述时间变化：
VHIGH＝VMIN+(VMAX-VMIN)Vout/Vin
其中Vin是所述降压切换调节器电路的所述输入端的电压，Vout是所述降压切换调节器电路的所述输出端的电压，VMAX是电压Vout的最大脉动电压值，VMIN是电压Vout的最小脉动电压值，VHIGH是所述第二阈值。
12.一种用于调节降压切换电压调节器电路的输出电压的方法，包括：
首先将所述降压切换电压调节器电路的一输出端的电压与第一阈值大小作比较；
其次将所述降压切换电压调节器电路的所述输出端的所述电压与第二阈值大小作比较，其中所述第二阈值大小大于所述第一阈值大小；
当所述首先比较者显示所述降压切换电压调节器电路的所述输出端的所述电压不超过所述第一阈值大小时，将电感式存储元件耦合在所述降压切换电压调节器电路的一输入端与所述降压切换电压调节器电路的所述输出端之间；
当所述其次比较者显示所述降压切换电压调节器电路的所述输出端的所述电压超过所述第二阈值大小时，将所述电感式存储元件耦合在所述降压切换电压调节器电路的所述输出端与所述切换电压调节器电路的所述输出端的返回路径之间；以及
通过计算由所述降压切换电压调节器电路的所述输出端向负载提供的输出电流的指示、以及据此改变所述第一阈值大小，产生作为时间变化电压信号的所述第一阈值大小和所述第二阈值大小的值，以使所述第一阈值大小随由所述降压切换电压调节器电路的输出端向负载提供的输出电流的增加而增加，且随所述输出电流的下降而下降。
13.根据权利要求12所述的方法，其中所述产生还包括根据所述输出电流的值改变所述第二阈值大小。
14.根据权利要求12所述的方法，其中所述产生由所述降压切换电压调节器电路的所述输出端的电压、以及从第二指示在所述降压切换电压调节器电路的给定周期内的起始处以及从第一指示在所述降压切换电压调节器电路的下一周期内的起始处延伸的时段，计算所述输出电流的所述指示。
15.根据权利要求14所述的方法，其中所述产生根据下列公式计算所述输出电流的指示：
(Vin–Vout)(t2–t0)/2L-C(VHIGH-VLOW)/(t2-t0)
其中Vin是所述降压切换调节器电路的所述输入端的电压，Vout是所述降压切换调节器电路的所述输出端的所述电压，C是所述降压切换调节器电路的所述输出端的电容，L是所述电感式存储元件的感应系数，t2是所述第二指示在给定周期内的起始处的时刻，t0是所述第一指示在所述给定周期内的起始处的时刻，VHIGH是所述第二阈值，且VLOW是所述第一阈值。
16.根据权利要求14所述的方法，其中所述产生根据下列公式计算所述输出电流的指示：
Vout(t4-t2)2/2L(t5-t2)-C(VHIGH-VLOW)/(t5-t2)
其中Vout是所述降压切换调节器电路的所述输出端的所述电压，C是所述降压切换调节器电路的所述输出端的电容，L是所述电感式存储元件的感应系数，t2是所述第二指示在给定周期内的起始处的时刻，t5是所述第一指示在所述下一周期内的起始处的时刻，VHIGH是所述第二阈值，VLOW是所述第一阈值，且t4是所述第二指示在所述给定周期内结束的时刻。
17.根据权利要求12所述的方法，还包括测量所述输出电流，以提供所述输出电流的指示。
18.根据权利要求12所述的方法，其中所述产生还根据在所述降压切换电压调节器电路的所述输入端的电压与所述降压切换电压调节器电路的所述输出端的所述电压之间的差值产生所述第一阈值大小。
19.根据权利要求18所述的方法，其中所述产生根据下列公式产生所述第一阈值大小：
I0 2L/2C(Vin–Vout)
其中Vin是所述降压切换调节器电路的所述输入端的电压，Vout是所述降压切换调节器电路的所述输出端的所述电压，C是所述降压切换调节器电路的所述输出端的电容，L是所述电感式存储元件的感应系数，且I0是所述输出电流的指示。
20.根据权利要求12所述的方法，其中所述降压切换电压调节器电路在连续传导模式下工作，且输出电流的指示是在输送给所述降压切换电压调节器电路的所述输出端的电流与在所述电感式存储元件中的最小电流之间的差值。
21.根据权利要求12所述的方法，其中所述产生由所述降压切换电压调节器电路的所述输出端的所述电压、以及所述降压切换电压调节器电路的所述输入端的电压产生所述第二阈值大小。
22.根据权利要求21所述的方法，其中所述产生根据下列公式产生所述第二阈值大小：
VHIGH＝VMIN+(VMAX-VMIN)Vout/Vin
其中Vin是所述降压切换调节器电路的所述输入端的电压，Vout是所述降压切换调节器电路的所述输出端的电压，VMAX是电压Vout的最大脉动电压值，VMIN是电压Vout的最小脉动电压值，VHIGH是所述第二阈值。
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CN102132478A (zh) 2011-07-20
TW201010255A (en) 2010-03-01
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