US20060232326A1

US20060232326A1 - Reference circuit that provides a temperature dependent voltage

Info

Publication number: US20060232326A1
Application number: US11/108,186
Authority: US
Inventors: Helmut Seitz; Russell Houghton; Ernst Stahl
Original assignee: Infineon Technologies AG
Current assignee: Infineon Technologies AG
Priority date: 2005-04-18
Filing date: 2005-04-18
Publication date: 2006-10-19
Also published as: KR20060109843A; US7196947B2; JP2006309756A; US20060233029A1; DE102006017794A1; KR100798436B1

Abstract

A reference circuit that includes a first circuit configured to provide a temperature dependent current, a second circuit configured to provide a first current, and a third circuit. The third circuit is configured to provide a temperature dependent voltage based on the first current and the temperature dependent current. The temperature dependent voltage has a voltage versus temperature slope established by the third circuit and a voltage level established by the first current.

Description

BACKGROUND

Memory speed and memory capacity continue to increase to meet the demands of system applications. Some of these system applications include mobile electronic systems that have limited space and limited power resources. In mobile applications, such as cellular telephones and personal digital assistants (PDAs), memory cell density and power consumption are issues for future generations.
To address these issues, the industry is developing random access memories (RAMs) for mobile applications. One type of RAM, referred to as CellularRAM, is a high performance and low power memory designed to meet the growing memory density and bandwidth demands of future designs. CellularRAM is a pseudo static RAM (PSRAM) that offers a lower cost per bit ratio than current solutions. Also, CellularRAM offers static RAM (SRAM) pin and function compatibility, external refresh-free operation, and a low power design. CellularRAM devices are drop-in replacements for most asynchronous low power SRAMs currently used in mobile applications, such as cellular telephones.
Typically, a PSRAM is based on a dynamic RAM (DRAM) that provides significant advantages in density and speed over traditional SRAM. The DRAM can include one transistor and one capacitor memory cells that are arranged in one or more arrays of memory cells, which are arranged in memory banks. To read and write memory cells, each DRAM includes one or more row decoders, one or more column decoders, and sense amplifiers. The sense amplifiers can be differential sense amplifiers, wherein each sense amplifier receives one bit line at each of two differential inputs.
To read or write memory cells, the DRAM receives a row address, a column address, and control signals, such as row address select (RAS) and column address select (CAS) signals. A row decoder receives the row address to select a row of memory cells and the row address is latched into the row decoder via the RAS signal. A column decoder receives the column address to select one or more columns of memory cells and the column address is latched into the column decoder via the CAS signal. Each memory cell at the intersection of a selected row and a selected column provides a data bit to a sense amplifier.
At each sense amplifier that receives data, one of the bit lines receives the data bit from a selected memory cell and the other bit line is used as a reference. To read the data bit, the sense amplifier amplifies the difference between the data bit value and the reference value and provides a sensed output value to an output driver. To write a data bit into a selected memory cell, input drivers overdrive the sense amplifier. One input driver overdrives a data bit value onto the bit line that is connected to the selected memory cell and another input driver overdrives the inverse of the data bit value onto the reference bit line.
Typically, a PSRAM receives SRAM control signals that do not include DRAM control signals, such as RAS and CAS signals. As a result, the PSRAM internally provides the DRAM control signals to read data from and write data into the DRAM memory. The DRAM control signals include timing constraints that must be met to operate the DRAM. These timing constraints include timing parameters such as the minimum amount of time that the RAS signal is active (tRAS), the minimum amount of time that the RAS signal is inactive, referred to as RAS precharge time (tRP), the minimum amount of time for a RAS cycle (tRC), and the minimum amount of time to recover from a write operation, referred to as write recovery time (tWR).
The PSRAM includes delay chains to provide the DRAM control signals and meet the timing constraints of the DRAM control signals. Typically, the speeds of the delay chains are temperature dependent, such that the timing of the DRAM control signals will vary with temperature. As the timing of the DRAM control signals change with temperature, the PSRAM may become unreliable or stop working altogether.
For these and other reasons there is a need for the present invention.

SUMMARY

One aspect of the present invention provides a reference circuit that includes a first circuit configured to provide a temperature dependent current, a second circuit configured to provide a first current, and a third circuit. The third circuit is configured to provide a temperature dependent voltage based on the first current and the temperature dependent current. The temperature dependent voltage has a voltage versus temperature slope established by the third circuit and a voltage level established by the first current.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating one embodiment of an electronic system according to the present invention.
FIG. 2 is a block diagram illustrating one embodiment of an electronic system according to the present invention including a controller and a RAM.
FIG. 3 is a diagram illustrating one embodiment of a memory cell in an array of memory cells.
FIG. 4 is a diagram illustrating one embodiment of a temperature dependent reference circuit.
FIG. 5 is a diagram illustrating one embodiment of a regulator circuit.
FIG. 6 is a diagram illustrating one embodiment of a temperature dependent reference circuit including a band gap reference circuit and a current mirror circuit.

DETAILED DESCRIPTION

In the following Detailed Description, reference is made to the accompanying drawings, which form a part hereof, and in which is shown by way of illustration specific embodiments in which the invention may be practiced. In this regard, directional terminology, such as “top,” “bottom,” “front,” “back,” “leading,” “trailing,” etc., is used with reference to the orientation of the Figure(s) being described. Because components of embodiments of the present invention can be positioned in a number of different orientations, the directional terminology is used for purposes of illustration and is in no way limiting. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope of the present invention. The following detailed description, therefore, is not to be taken in a limiting sense, and the scope of the present invention is defined by the appended claims.
FIG. 1 is a block diagram illustrating one embodiment of an electronic system 20 according to the present invention. The electronic system 20 includes a temperature dependent reference circuit 22. Electronic system 20 can be any suitable electronic system, such as an integrated circuit chip, multiple integrated circuit chips, or a computer system including a controller and memory.
Temperature dependent reference circuit 22 provides a temperature dependent reference voltage VREF at 24. Temperature dependent reference circuit 22 includes circuitry that provides a temperature dependent current, a constant current, and the temperature dependent reference voltage VREF at 24 based on the constant current and the temperature dependent current. The temperature dependent reference voltage VREF at 24 has a voltage versus temperature slope that is established via one part of the temperature dependent reference circuit 22 and a voltage level that is established via another part of the temperature dependent reference circuit 22. The voltage versus temperature slope and the voltage level of the temperature dependent reference voltage VREF at 24 are individually established parameters of the temperature dependent reference voltage VREF at 24. In one embodiment, as the temperature increases, the temperature dependent reference voltage VREF at 24 increases and as the temperature decreases, the temperature dependent reference voltage VREF at 24 decreases.
Electronic system 20 includes circuitry that receives the temperature dependent reference voltage VREF at 24 and uses the temperature dependent reference voltage VREF at 24 for any suitable purpose in electronic system 20. In one embodiment, electronic system 20 includes a delay chain, wherein the speed of the delay chain varies with temperature due to device characteristics. As the temperature increases, the speed of the delay chain decreases and as the temperature decreases, the speed of the delay chain increases. The speed of the delay chain is also dependent on the delay chain supply voltage. As the supply voltage increases, the speed of the delay chain increases and as the supply voltage decreases, the speed of the delay chain decreases. To keep the speed of the delay chain constant, electronic circuit 20 provides a temperature dependent output voltage as a supply voltage to the delay chain. The temperature dependent output voltage is based on the temperature dependent reference voltage VREF at 24. As the temperature increases, the temperature dependent reference voltage VREF at 24 and the temperature dependent output voltage increase to increase the speed of the delay chain at higher temperatures. As the temperature decreases, the temperature dependent reference voltage VREF at 24 and the temperature dependent output voltage decrease to decrease the speed of the delay chain at lower temperatures. The speed of the delay chain can be made constant over temperature, i.e., independent of temperature, using the temperature dependent reference voltage VREF at 24 to provide a temperature dependent output voltage as the delay chain supply voltage.
FIG. 2 is a block diagram illustrating one embodiment of an electronic system 40 according to the present invention. Electronic system 40 includes a controller 42 and a RAM 44. Controller 42 is electrically coupled to RAM 44 via memory communications path 46 and data communications path 48. Controller 42 provides row and column addresses and control signals to RAM 44 via memory communications path 46. Controller 42 provides data to RAM 44 and receives data from RAM 44 via data communications path 48. In one embodiment, RAM 44 is a PSRAM. In other embodiments, RAM 44 can be any suitable RAM.
RAM 44 includes an array of memory cells 50, a row address latch and decoder 52, a column address latch and decoder 54, a sense amplifier circuit 56, a RAM I/O circuit 58, an address register 60, a control circuit 62, and a temperature dependent voltage source 64. Conductive word lines 66, referred to as row select lines, extend in the x-direction across the array of memory cells 50. Conductive bit lines 68, referred to as bit lines, extend in the y-direction across the array of memory cells 50. A memory cell 70 is located at each cross point of a word line 66 and a bit line 68.
Each word line 66 is electrically coupled to row address latch and decoder 52 and each bit line 68 is electrically coupled to one of the sense amplifiers in sense amplifier circuit 56. The sense amplifier circuit 56 is electrically coupled to column address latch and decoder 54 via conductive column select lines 72. Also, sense amplifier circuit 56 is electrically coupled to row address latch and decoder 52 via communications path 74 and to RAM I/O circuit 58 via I/O communications path 76. Data is transferred between RAM I/O circuit 58 and controller 42 via data communications path 48.
Controller 42 is electrically coupled to address register 60 and control circuit 62 via memory communications path 46. Address register 60 is electrically coupled to row address latch and decoder 52 and column address latch and decoder 54 via row and column address lines 78. Control circuit 62 is electrically coupled to row address latch and decoder 52 and column address latch and decoder 54 via control communications path 80. Also, control circuit 62 is electrically coupled to temperature dependent voltage source 64 via voltage source path 82.
Temperature dependent voltage source 64 includes a temperature dependent reference circuit 84 and a regulator circuit 86. Temperature dependent reference circuit 84 is electrically coupled to regulator circuit 86 via reference voltage path 88. Control circuit 62 includes a delay block 90 that is electrically coupled to regulator circuit 86 via voltage source path 82.
Address register 60 receives row and column addresses from controller 42 via memory communications path 46. Address register 60 supplies a row address to row address latch and decoder 52 via row and column address lines 78, and control circuit 62 supplies a RAS signal to row address latch and decoder 52 via control communications path 80 to latch the supplied row address into row address latch and decoder 52. Address register 60 supplies a column address to column address latch and decoder 54 via row and column address lines 78, and control circuit 62 supplies a CAS signal to column address latch and decoder 54 via control communications path 80 to latch the supplied column address into column address latch and decoder 54.
Controller 42 and I/O circuit 58 communicate data between controller 42 and RAM 44 via data communications path 48. I/O circuit 58 includes a suitable number of transmitter and receiver pairs and controller 42 includes a suitable number of transmitter and receiver pairs. Each transmitter and receiver pair in I/O circuit 58 corresponds to a transmitter and receiver pair in controller 42. Data communications path 48 includes one or more signal lines and each transmitter and receiver pair in I/O circuit 58 is electrically coupled to the corresponding transmitter and receiver pair in controller 42 via one of the signal lines in data communications path 48.
Sense amplifier circuit 56 includes sense amplifiers, equalization and precharge circuits, and switches. The sense amplifiers are differential input sense amplifiers and each sense amplifier receives one bit line 68 at each of the two differential inputs. One of the bit lines 68 receives a data bit from a selected memory cell 70 and the other bit line 68 is used as a reference. The equalization and precharge circuits equalize the voltage on bit lines 68 connected to the same sense amplifier prior to a read or write operation. To read a data bit, a sense amplifier amplifies the difference between the data bit value and the reference value and provides a sensed output value to I/O circuit 58 via I/O communications path 76. One of the transmitter and receiver pairs in I/O circuit 58 receives the sensed output value and provides the sensed output value to the corresponding transmitter and receiver pair in controller 42 via data communications path 48. To write a data bit, one of the transmitter and receiver pairs in controller 42 provides a data bit to the corresponding transmitter and receiver pair in I/O circuit 58 in RAM 44 via data communications path 48. I/O circuit 58 provides the data bit to a sense amplifier in sense amplifier circuit 56 via I/O communications path 76. I/O circuit 58 overdrives the sense amplifier to overdrive the data bit value onto the bit line 68 that is connected to one of the memory cells 70 and to overdrive the inverse of the data bit value onto the reference bit line 68. The sense amplifier writes the received data bit value into the selected memory cell 70.
Row address latch and decoder 52 receives row addresses and RAS signals and latches the row addresses into row address latch and decoder 52. Row address latch and decoder 52 decodes each of the row addresses to select a row of memory cells 70. In addition, row address latch and decoder 52 provides sense amplifier activation signals and equalization and precharge signals to sense amplifier circuit 56 via communications path 74.
Column address latch and decoder 54 activates column select lines 72 to connect sense amplifiers in sense amplifier circuit 56 to transmitter and receiver pairs in I/O circuit 58. Column address latch and decoder 54 receives a column address and latches the column address into column address latch and decoder 54. Column address latch and decoder 54 decodes the column address to select addressed column select lines 72. In addition, column address latch and decoder 54 receives column select line activation signals from control circuit 62 via control communications path 80. The column select line activation signals indicate which of the addressed column select lines 72 are to be activated by column address latch and decoder 54. Column address latch and decoder 54 activates column select lines 72 that are addressed by the column address and selected for activation by the column select line activation signals. Activated column select lines 72 are provided to sense amplifier circuit 56 to connect sense amplifiers in sense amplifier circuit 56 to transmitter and receiver pairs in I/O circuit 58.
Controller 42 provides addresses and control signals to control circuit 62 via memory communications path 46. Control circuit 62 receives the addresses and control signals from controller 42 and provides internal control signals to read data from or write data into the array of memory cells 50. Control circuit 62 provides RAS signals to row address latch and decoder 52 and CAS signals to column address latch and decoder 54. Also, control circuit 62 provides internal control signals to column address latch and decoder 52 to selectively activate column select lines 72.
In one embodiment, RAM 44 is a PSRAM and controller 42 provides SRAM control signals to control circuit 62. The SRAM control signals do not include DRAM control signals, such as RAS and CAS signals, and control circuit 62 provides the DRAM control signals in response to the SRAM control signals. The DRAM control signals include timing constraints to read data from or write data into the array of memory cells 50. These timing constraints include timing parameters such as the minimum amount of time that the RAS signal is active (tRAS), the minimum amount of time that the RAS signal is inactive, referred to as RAS precharge time (tRP), the minimum amount of time for a RAS cycle (tRC), and the minimum amount of time to recover from a write operation, referred to as write recovery time (tWR).
Delay block 90 includes delay chains that are used to provide internal control signals and to meet the timing constraints of the internal control signals. The speed of each of the delay chains in delay block 90 varies with temperature due to device characteristics. As the temperature increases, the speed of each of the delay chains decreases and as the temperature decreases, the speed of each of the delay chains increases. Also, the speed of each of the delay chains is dependent on the supply voltage provided to delay block 90. As the supply voltage increases, the speed of each of the delay chains increases and as the supply voltage decreases, the speed of each of the delay chains decreases. To provide constant delay times over temperature, the supply voltage provided to delay block 90 increases at higher temperatures and decreases at lower temperatures.
Temperature dependent voltage source 64 provides a temperature dependent output voltage VOUT to delay block 90 via voltage source path 82. The temperature dependent output voltage VOUT is used as the supply voltage for the delay chains in delay block 90. As the temperature increases, the temperature dependent output voltage VOUT increases and as the temperature decreases, the temperature dependent output voltage VOUT decreases.
Temperature dependent reference circuit 84 is similar to temperature dependent reference circuit 22 (shown in FIG. 1). Temperature dependent reference circuit 84 provides a temperature dependent reference voltage VREF to regulator circuit 86 via reference voltage path 88. Temperature dependent reference circuit 84 includes circuitry that provides a temperature dependent current, a constant current, and the temperature dependent reference voltage VREF based on the constant current and the temperature dependent current. The temperature dependent reference voltage VREF has a voltage versus temperature slope that is established via one part of the temperature dependent reference circuit 84 and a voltage level that is established via another part of the temperature dependent reference circuit 84. The voltage versus temperature slope and the voltage level of the temperature dependent reference voltage VREF are individually established parameters of the temperature dependent reference voltage VREF. As the temperature increases, the temperature dependent reference voltage VREF increases and as the temperature decreases, the temperature dependent reference voltage VREF decreases.
Regulator circuit 86 receives the temperature dependent reference voltage VREF via reference voltage path 88 and provides the temperature dependent output voltage VOUT to delay block 90 via voltage source path 82. The temperature dependent output voltage VOUT is based on the temperature dependent reference voltage VREF. As the temperature increases, the temperature dependent reference voltage VREF and the temperature dependent output voltage VOUT increase. As the temperature decreases, the temperature dependent reference voltage VREF and the temperature dependent output voltage VOUT decrease. The speed of the delay chains in delay block 90 can be made constant over temperature, i.e., independent of temperature, using the temperature dependent reference voltage VREF to provide a temperature dependent output voltage VOUT as the supply voltage for delay block 90.
During a read operation, control circuit 62 receives read control signals and address register 60 receives the row address of a selected memory cell or cells 70. The row address is supplied from address register 60 to row address latch and decoder 52 and latched into row address latch and decoder 52 by control circuit 62 that provides a RAS signal. Row address latch and decoder 52 decodes the row address and activates the selected word line 66. As the selected word line 66 is activated, the value stored in each memory cell 70 coupled to the selected word line 66 is passed to the respective bit line 68. The bit value stored at a memory cell 70 is detected by a sense amplifier that is electrically coupled to the respective bit line 68.
Next, control circuit 62 and address register 60 receive the column address of the selected memory cell or cells 70. The column address is supplied from address register 60 to column address latch and decoder 54 and latched into column address latch and decoder 54 by control circuit 62 that provides a CAS signal. The column address latch and decoder 54 decodes the column address to select column select lines 72. Control circuit 62 provides internal control signals to column address latch and decoder 54 to selectively activate column select lines 72 and connect selected sense amplifiers to transmitter and receiver pairs in I/O circuit 58. Sensed output values are provided to transmitter and receiver pairs in I/O circuit 58 and to the corresponding transmitter and receiver pairs in controller 42 via data communications path 48.
During a write operation, data to be stored in the array of memory cells 50 is supplied from transmitter and receiver pairs in controller 42 to transmitter and receiver pairs in I/O circuit 58 via data communications path 48. Control circuit 62 receives write control signals and address register 60 receives the row address of a selected memory cell or cells 70. The row address is supplied from address register 60 to row address latch and decoder 52 and latched into row address latch and decoder 52 by control circuit 62 that provides a RAS signal. The row address latch and decoder 52 decodes the row address and activates the selected word line 66. As the selected word line 66 is activated, the value stored in each memory cell 70 coupled to the selected word line 66 is passed to the respective bit line 68 and the sense amplifier that is electrically coupled to the respective bit line 68.
Next, control circuit 62 and address register 60 receive the column address of the selected memory cell or cells 70. Address register 60 supplies the column address to column address latch and decoder 54 and the column address is latched into column address latch and decoder 54 by control circuit 62 via a CAS signal. Column address latch and decoder 54 receives column select line activation signals from control circuit 62 and activates selected column select lines 72 to connect sense amplifiers in sense amplifier circuit 56 to transmitter and receiver pairs in I/O circuit 58. I/O circuit 58 passes data from controller 42 to the sense amplifiers and overdrives the sense amplifiers to write data to the selected memory cell or cells 70 via bit lines 68.
FIG. 3 is a diagram illustrating one embodiment of a memory cell 70 in the array of memory cells 50. Memory cell 70 includes a transistor 92 and a capacitor 94. The gate of transistor 92 is electrically coupled to word line 66. One side of the drain-source path of transistor 92 is electrically coupled to bit line 68 and the other side of the drain-source path is electrically coupled to one side of capacitor 94. The other side of capacitor 94 is electrically coupled to a reference 96, such as one-half the supply voltage. Capacitor 94 is charged and discharged to represent a logic 0 or a logic 1.
During a read operation, word line 66 is activated to turn on transistor 92 and the value stored on capacitor 94 is read by a sense amplifier via bit line 68. During a write operation, word line 66 is activated to turn on transistor 92 to access capacitor 94. The sense amplifier connected to bit line 68 is overdriven to write a data value on capacitor 94 via bit line 68 and transistor 92.
A read operation on memory cell 70 is a destructive read operation. After each read operation, capacitor 94 is recharged or discharged to the data value that was just read. In addition, even without read operations, the charge on capacitor 94 discharges over time. To retain a stored value, memory cell 70 is refreshed periodically by reading and/or writing memory cell 70. All memory cells 70 in the array of memory cells 50 are periodically refreshed to maintain their values.
FIG. 4 is a diagram illustrating one embodiment of a temperature dependent reference circuit 100. Temperature dependent reference circuit 100 is similar to temperature dependent reference circuit 22 (shown in FIG. 1) and temperature dependent reference circuit 84 (shown in FIG. 2). Temperature dependent reference circuit 100 provides a temperature dependent reference voltage VREF at 102. As the temperature increases, the temperature dependent reference voltage VREF at 102 increases, and as the temperature decreases, the temperature dependent reference voltage VREF at 102 decreases.
Temperature dependent reference circuit 100 includes a differential amplifier 104, a first p-channel metal oxide semiconductor (PMOS) transistor 106, a capacitor 108, a first resistor 110, a diode 112, a second resistor 114, a second PMOS transistor 116, and a third resistor 118. The negative input of differential amplifier 104 receives input voltage VIN at 120 and the output of differential amplifier 104 is electrically coupled at 122 to one side of capacitor 108, the gate of first PMOS transistor 106, and the gate of second PMOS transistor 116. The other side of capacitor 108 is electrically coupled to a reference, such as ground, at 124.
One side of the drain-source path of first PMOS transistor 106 is electrically coupled at 126 to first resistor 110, second resistor 114, and the positive input of differential amplifier 104. The other side of second resistor 114 is electrically coupled at 128 to a reference, such as ground, and the other side of first resistor 110 is electrically coupled at 130 to the anode of diode 112. The cathode of diode 112 is electrically coupled to a reference, such as ground, at 132.
The other side of the drain-source path of first PMOS transistor 106 is electrically coupled to a power supply voltage VDD at 134. One side of second PMOS transistor 116 is electrically coupled to the power supply voltage VDD at 134. The other side of the drain-source path of second PMOS transistor 116 is electrically coupled at 102 to third resistor 118 and the other side of third resistor 118 is electrically coupled to a reference, such as ground, at 136.
Differential amplifier 104 receives input voltage VIN at 120 and provides an output voltage at 122 that turns on first PMOS transistor 106 and second PMOS transistor 116. The input voltage VIN is a substantially constant voltage. Current I1 passes through first PMOS transistor 106 and is divided into a substantially constant current ICC that passes through second resistor 114 and a temperature dependent current ITD that passes through the series combination of first resistor 110 and diode 112. The voltage at the drain of first PMOS transistor 106 and at the positive input of differential amplifier 104 is regulated to substantially the value of the constant input voltage VIN by differential amplifier 104. Capacitor 108 stabilizes the feedback loop from the output of differential amplifier 104 to the gate of first PMOS transistor 106 and back to the positive input of differential amplifier 104.
The current I1 is equal to the sum of the constant current ICC and the temperature dependent current ITD. Second resistor 114 is part of a constant-current source that provides constant current ICC. The constant current ICC is equal to the value of the input voltage VIN divided by the resistance value R2 of second resistor 114. First resistor 110 and diode 112 are part of a temperature dependent current source that provides temperature dependent current ITD. The temperature dependent current ITD is equal to the value of the input voltage VIN minus the temperature dependent voltage VBE across diode 112, divided by the resistance value R1 of first resistor 110. As the temperature increases, the voltage VBE across diode 112 decreases and the temperature dependent current ITD increases. As the temperature decreases, the voltage VBE across diode 112 increases and the temperature dependent current ITD decreases.
The current I1 through first PMOS transistor 106 is mirrored as current I2 through second PMOS transistor 116. Current I2 is substantially equal to current I1 and current I2 passes through second PMOS transistor 116 and third resistor 118 to provide the temperature dependent output voltage VREF at 102. With the reference at 136 at ground, the current I2 times the resistance value R3 of third resistor 118 equals the temperature dependent output voltage VREF at 102.
In operation, as the temperature increases, the voltage VBE across diode 112 decreases and the temperature dependent current ITD increases. This increases current I1 and current 12, which increases the temperature dependent output voltage VREF at 102. As the temperature decreases, the voltage VBE across diode 112 increases and the temperature dependent current ITD decreases. This decreases current I1 and current I2, which decreases temperature dependent output voltage VREF at 102.
The temperature dependent output voltage VREF at 102 is determined as follows in Equation I: $\begin{matrix} VREF = (\frac{VIN}{R 2} + \frac{VIN - VBE}{R 1}) * R 3 & Equation I \end{matrix}$
Equation I can be rearranged into Equation II: $\begin{matrix} VREF = ((\frac{R 1 + R 2}{R 1 * R 2}) * R 3 * VIN) - (\frac{R 3}{R 1} * VBE) & Equation II \end{matrix}$
In Equation II, temperature dependent reference voltage VREF includes a temperature independent part and a temperature dependent part. The temperature independent part comes before the minus sign and is based on the resistance values R1, R2, and R3 and the input voltage VIN. The temperature dependent part follows the minus sign and is based on the resistance values R3 and R1 and the temperature dependent voltage VBE across diode 112. The temperature dependent voltage VBE across diode 112 is determined roughly as follows in Equation III:
VBE=0.662−0.0025(T−25) Equation III
where T is the temperature in degrees Celsius (C.).
From Equation III, the temperature dependent voltage VBE across diode 112 changes at a rate of 2.5 millivolts (mV) per degrees C. In Equation II, the temperature dependent voltage VBE is multiplied by resistance value R3 of third resistor 118 and divided by the resistance value R1 of first resistor 110. The voltage versus temperature slope of the temperature dependent reference voltage VREF is established by setting the resistance values R1 and R3.
For example, to change the temperature dependent reference voltage VREF 150 mV over a 140 degree C. temperature range of −30 degrees C. to 110 degrees C., the temperature dependent part of temperature dependent reference voltage VREF is as follows in Equation IV: $\begin{matrix} 2.5 \frac{mV}{°C} * \frac{R 3}{R 1} = \frac{150 mV}{140 °C} = 1.071 \frac{mV}{°C} & Equation IV \end{matrix}$
Setting the resistance value R1 of first resistor 110 equal to 1 mega Ohm (MOhm) and solving for the resistance value R3 of third resistor 118, results in a resistance value R3 of 0.429 MOhms. If the resistance value R1 of first resistor 110 is constant, the voltage versus temperature slope of the temperature dependent reference voltage VREF is established by setting the resistance value R3 of third resistor 118.
To further the example, input voltage VIN is set to 1.2 volts and Equation III is combined with Equation II. Substituting in a resistance value of 1 MOhm for resistance value R1 and a resistance value of 0.429 MOhms for resistance value R3, results in Equation V. $\begin{matrix} VREF = 0.514 * (\frac{1 + R 2}{R 2}) - 0.311 + (.001071 * T) & Equation V \end{matrix}$
Setting the temperature dependent reference voltage VREF to 0.9 volts and the temperature to 30 degrees C. and solving for the resistance value R2 of second resistor 114, results in a resistance value R2 of 0.773 MOhms. At a given temperature, the voltage level of the temperature dependent reference voltage VREF is established by setting the resistance value R2 of second resistor 114. Substituting the resistance value R2 of 0.773 MOhms back into Equation V results in Equation VI.
VREF=0.868+(0.001071*T) Equation VI
The temperature dependent reference voltage VREF in Equation VI is equal to 0.9 volts at 30 degrees C. and has a voltage versus temperature slope of 150 mV over 140 degrees C., between −30 degrees C. and 110 degrees C.
Second resistor 114 and third resistor 118 are variable resistors configured to establish the voltage level and the voltage versus temperature slope of the temperature dependent reference voltage VREF. With the resistance value R1 constant, the resistance value R3 of third resistor 118 establishes the slope of the temperature dependent reference voltage VREF, which also changes the voltage level of the temperature dependent reference voltage VREF. The resistance value R2 of second resistor 114 is then configured to establish the voltage level of the temperature dependent reference voltage VREF. The voltage versus temperature slope and the voltage level of the temperature dependent reference voltage VREF are established independently via third resistor 118 and second resistor 114.
In one embodiment, second resistor 114 includes multiple resistors connected in series and each of the multiple resistors can be selected or shorted to provide resistance value R2. In one embodiment, third resistor 118 includes multiple resistors connected in series and each of the multiple resistors can be selected or shorted to provide resistance value R3. In one embodiment, second resistor 114 is a digitally variable resistor. In one embodiment, third resistor 118 is a digitally variable resistor. In other embodiments, first resistor 110 can be a variable resistor.
FIG. 5 is a diagram illustrating one embodiment of a regulator circuit 200. Regulator circuit 200 is similar to regulator circuit 86 (shown in FIG. 2). Regulator circuit 200 receives the temperature dependent reference voltage VREF at 202 and provides the temperature dependent output voltage VOUT at 204. The temperature dependent output voltage VOUT at 204 is based on the temperature dependent reference voltage VREF at 202. As the temperature increases, the temperature dependent reference voltage VREF at 202 increases and the temperature dependent output voltage VOUT at 204 increases. As the temperature decreases, the temperature dependent reference voltage VREF at 202 decreases and the temperature dependent output voltage VOUT at 204 decreases.
Regulator circuit 200 includes an input circuit 206 and an output circuit 208. Input circuit 206 is a differential amplifier including a first n-channel metal oxide semiconductor (NMOS) transistor 210, a second NMOS transistor 212, a third NMOS transistor 214, a first PMOS transistor 216, and a second PMOS transistor 218. The first NMOS transistor 210 and the second NMOS transistor 212 are input transistors and the third NMOS transistor 214 provides a current source for the differential amplifier. The first PMOS transistor 216 and the second PMOS transistor 218 are arranged as a current mirror.
The gate of the first NMOS transistor 210 receives the temperature dependent reference voltage VREF at 202 and one side of the drain-source path of first NMOS transistor 210 is electrically coupled at 220 to one side of the drain-source path of second NMOS transistor 212 and to one side of the drain-source path of third NMOS transistor 214. The other side of the drain-source path of third NMOS transistor 214 is electrically coupled to a reference, such as ground, at 222. The gate of third NMOS transistor 214 receives a current source voltage VC1 at 224 that turns on third NMOS transistor 214 to provide a current source for the differential amplifier.
The other side of the drain-source path of first NMOS transistor 210 is electrically coupled at 226 to one side of the drain-source path of first PMOS transistor 216 and to the gates of first PMOS transistor 216 and second PMOS transistor 218. The other side of the drain-source path of first PMOS transistor 216 is electrically coupled to a power supply voltage VPP at 228. One side of the drain-source path of second PMOS transistor 218 is electrically coupled to a power supply voltage VPP at 228. The other side of the drain-source path of second PMOS transistor 218 is electrically coupled at 230 to the other side of the drain-source path of second NMOS transistor 212 and to output circuit 208. The gate of the second NMOS transistor 212 is electrically coupled at 232 to output circuit 208.
Output circuit 208 includes a third PMOS transistor 234, a fourth PMOS transistor 236, a fourth NMOS transistor 238, a fifth NMOS transistor 240, a first resistor 242, and a second resistor 244. The gate of the third PMOS transistor 234 is electrically coupled at 230 to the drain-source path of second PMOS transistor 218 and the drain-source path of second NMOS transistor 212. One side of the drain-source path of the third PMOS transistor 234 is electrically coupled at 246 to one side of the drain source path of fourth NMOS transistor 238 and the gate of fourth PMOS transistor 236. The other side of the drain-source path of fourth NMOS transistor 238 is electrically coupled to a reference, such as ground, at 248. The gate of fourth NMOS transistor 238 receives a current source voltage VC2 at 250 that turns on fourth NMOS transistor 238 to provide a current source. The other side of the drain-source path of third PMOS transistor 234 is electrically coupled to the power supply voltage VPP at 228. One side of the drain source path of fourth PMOS transistor 236 is electrically coupled to the power supply voltage VPP at 228.
The other side of the drain-source path of the fourth PMOS transistor 236 is electrically coupled at 204 to one side of first resistor 242. The other side of first resistor 242 is electrically coupled at 232 to the gate of second NMOS transistor 212 and one side of second resistor 244. The other side of second resistor 244 is electrically coupled at 252 to one side of the drain-source path of fifth NMOS transistor 240. The other side of the drain-source path of fifth NMOS transistor 240 is electrically coupled to a reference, such as ground, at 254. The gate of fifth NMOS transistor 240 receives a current source voltage VC3 at 256 that turns on fifth NMOS transistor 240 to provide a current source. In one embodiment, current source voltages VC1, VC2, and VC3 are provided by the same voltage supply and are the same voltage value.
First NMOS transistor 210 receives the temperature dependent reference voltage VREF at 202 and is biased to pass a first current from first PMOS transistor 216 to third NMOS transistor 214 and the reference at 222. The current mirror's second PMOS transistor 218 is biased to pass a second current that mirrors the first current. The output circuit 208 responds to the voltage at 230 to provide the temperature dependent output voltage VOUT and stabilize the voltage at the gate of second NMOS transistor 212 to substantially the voltage value of the temperature dependent reference voltage VREF.
If second NMOS transistor 212 is biased to pass less current than first NMOS transistor 210, the voltage at 230 increases and third PMOS transistor 234 is biased to pass less current. Fourth NMOS transistor 238 pulls the voltage at 246 lower, which biases fourth PMOS transistor 236 to pass more current. Current that passes through fourth PMOS transistor 236 flows through first and second resistors 242 and 244 and fifth NMOS transistor 240 to the reference at 254. A larger current through fourth PMOS transistor 236 raises the temperature dependent output voltage VOUT and the gate voltage of second NMOS transistor 212, between the first and second resistors 242 and 244. This biases second NMOS transistor 212 to pass more current and lower the voltage at 230. The lower voltage at 230 biases third PMOS transistor 234 to pass more current, which raises the voltage at 246 and biases fourth PMOS transistor 236 to pass less current. The regulator circuit 200 stabilizes with the gate voltage of second NMOS transistor 212 substantially equal to the temperature dependent reference voltage VREF at 202.
If second NMOS transistor 212 is biased to pass more current than first NMOS transistor 210, the voltage at 230 decreases and third PMOS transistor 234 is biased to pass more current. The voltage at 246 increases and fourth PMOS transistor 236 is biased to pass less current. The smaller current through fourth PMOS transistor 236 lowers the temperature dependent output voltage VOUT and the gate voltage of second NMOS transistor 212. Second NMOS transistor 212 is biased to pass less current, which raises the voltage at 230. The higher voltage at 230 biases third PMOS transistor 234 to pass less current, which lowers the voltage at 246 and fourth PMOS transistor 236 is biased to pass more current. The regulator circuit 200 stabilizes with the gate voltage of second NMOS transistor 212 substantially equal to the temperature dependent reference voltage VREF at 202.
In one embodiment, the resistance value of first resistor 242 is equal to the resistance value of second resistor 244 and the temperature dependent output voltage VOUT is substantially twice the temperature dependent reference voltage VREF. In one embodiment, temperature dependent reference circuit 100 of FIG. 4 receives a power supply voltage VDD of 1.8 volts that is boosted to provide a power supply voltage VPP of 2.9 volts to regulator circuit 200. In one embodiment, temperature dependent reference circuit 100 provides a temperature dependent reference voltage VREF that ranges from 0.85-1.00 volt and regulator circuit 200 provides a temperature dependent output voltage VOUT that ranges from 1.7-2.0 volts.
FIG. 6 is a diagram illustrating one embodiment of a temperature dependent reference circuit 300. Temperature dependent reference circuit 300 is similar to temperature dependent reference circuit 22 (shown in FIG. 1) and temperature dependent reference circuit 84 (shown in FIG. 2). Temperature dependent reference circuit 300 provides a temperature dependent reference voltage VREF at 302. As the temperature increases, the temperature dependent reference voltage VREF at 302 increases, and as the temperature decreases, the temperature dependent reference voltage VREF at 302 decreases.
Temperature dependent reference circuit 300 includes a bandgap reference circuit 304, a current mirror circuit 306, a differential amplifier 308, a first PMOS transistor 310, a capacitor 312, a first resistor 314, a second PMOS transistor 316, and a second resistor 318. Bandgap reference circuit 304 provides a constant, temperature independent input voltage VIN at 320 to the negative input of differential amplifier 308. The output of differential amplifier 308 is electrically coupled at 322 to one side of capacitor 312, the gate of first PMOS transistor 310, and the gate of second PMOS transistor 316. The other side of capacitor 312 is electrically coupled to a reference, such as ground, at 324.
One side of the drain-source path of first PMOS transistor 310 is electrically coupled at 326 to current mirror circuit 306, first resistor 314, and the positive input of differential amplifier 308. The other side of first resistor 314 is electrically coupled at 328 to a reference, such as ground. The other side of the drain-source path of first PMOS transistor 310 is electrically coupled to a power supply voltage VDD at 330. One side of second PMOS transistor 316 is electrically coupled to the power supply voltage VDD at 330. The other side of the drain-source path of second PMOS transistor 316 is electrically coupled at 302 to second resistor 318 and the other side of second resistor 318 is electrically coupled to a reference, such as ground, at 332.
Bandgap reference circuit 304 includes a first bandgap circuit PMOS transistor 334, a second bandgap circuit PMOS transistor 336, a first bandgap circuit NMOS transistor 338, a second bandgap circuit NMOS transistor 340, a first diode 342, a second diode 344, and a first bandgap circuit resistor 346. Bandgap reference circuit 304 also includes a third bandgap circuit PMOS transistor 348, a third diode 350, and a second bandgap circuit resistor 352.
First bandgap circuit PMOS transistor 334 and second bandgap circuit PMOS transistor 336 are arranged as a current mirror. The gate of first bandgap circuit PMOS transistor 334 is electrically coupled at 354 to the gate and one side of the drain-source path of second bandgap circuit PMOS transistor 336 and to one side of the drain-source path of second bandgap circuit NMOS transistor 340. The other side of the drain-source path of second bandgap circuit PMOS transistor 336 is electrically coupled to power supply voltage VDD at 356. One side of the drain-source path of first bandgap circuit PMOS transistor 334 is electrically coupled to the power supply voltage VDD at 356. The other side of the drain-source path of first bandgap circuit PMOS transistor 334 is electrically coupled at 358 to the gate and one side of the drain-source path of first bandgap circuit NMOS transistor 338 and to the gate of second bandgap circuit NMOS transistor 340.
The other side of the drain-source path of first bandgap circuit NMOS transistor 338 is electrically coupled at 360 to the anode of first diode 342. The other side of the drain-source path of second bandgap circuit NMOS transistor 340 is electrically coupled at 362 to one side of first bandgap circuit resistor 346. The other side of first bandgap circuit resistor 346 is electrically coupled at 364 to the anode of second diode 344. The cathode of first diode 342 is electrically coupled at 366 to the cathode of second diode 344 and to a reference such as ground.
Third bandgap circuit PMOS transistor 348 is arranged in the current mirror including first bandgap circuit PMOS transistor 334 and second bandgap circuit PMOS transistor 336. The gate of third bandgap circuit PMOS transistor 348 is electrically coupled at 354 to the gates of first bandgap circuit PMOS transistor 334 and second bandgap circuit PMOS transistor 336. Also, one side of the drain-source path of third bandgap circuit PMOS transistor 348 is electrically coupled to the power supply voltage VDD at 356. The other side of the drain-source path of third bandgap circuit PMOS transistor 348 is electrically coupled at 320 to one side of second bandgap circuit resistor 352 and the negative input of differential amplifier 308. The other side of second bandgap circuit resistor 352 is electrically coupled at 368 to the anode of third diode 350 and the cathode of third diode 350 is electrically coupled to a reference such as ground, at 370.
The current mirror circuit 306 includes a mirror circuit PMOS transistor 372, a first mirror circuit NMOS transistor 374, and a second mirror circuit NMOS transistor 376. Mirror circuit PMOS transistor 372 is arranged in the current mirror including first bandgap circuit PMOS transistor 334 and second bandgap circuit PMOS transistor 336. The gate of mirror circuit PMOS transistor 372 is electrically coupled at 354 to the gates of first bandgap circuit PMOS transistor 334 and second bandgap circuit PMOS transistor 336. Also, one side of the drain-source path of mirror circuit PMOS transistor 372 is electrically coupled to the power supply voltage VDD at 356.
The other side of the drain-source path of mirror circuit PMOS transistor 372 is electrically coupled at 378 to the gate and one side of the drain-source path of first mirror circuit NMOS transistor 374 and to the gate of second mirror circuit NMOS transistor 376. The other side of the drain-source path of first mirror circuit NMOS transistor 374 is electrically coupled to a reference, such as ground, at 380. One side of the drain-source path of second mirror circuit NMOS transistor 376 is electrically coupled to a reference, such as ground, at 382 and the other side of the drain-source path of second mirror circuit NMOS transistor 376 is electrically coupled at 326 to first PMOS transistor 310, first resistor 314, and the positive input of differential amplifier 308.
In bandgap reference circuit 304, second bandgap circuit PMOS transistor 336 is biased to pass a bandgap reference current IBGR from the power supply voltage VDD to second bandgap circuit NMOS transistor 340. The bandgap reference current IBGR flows through second bandgap circuit NMOS transistor 340 and first bandgap circuit resistor 346 and second diode 344 to the reference at 366. First bandgap circuit PMOS transistor 334 is biased to pass a mirror current that is substantially equal in value to the bandgap reference current IBGR. The mirror current flows from the power supply voltage VDD and through first bandgap circuit PMOS transistor 334 and first bandgap circuit NMOS transistor 338 and first diode 342 to the reference at 366.
The bandgap reference current IBGR is a temperature dependent current that increases as the temperature increases and decreases as the temperature decreases. The bandgap reference current IBGR changes as a function of V=kT/q, where k is Boltzmann's constant, T is temperature, and q is the charge of an electron. As the temperature increases, the voltage across second diode 344 decreases, which increases the bandgap reference current IBGR. As the temperature decreases, the voltage across second diode 344 increases, which decreases the bandgap reference current IBGR.
Third bandgap circuit PMOS transistor 348 is biased to pass a mirror current that is substantially equal in value to the bandgap reference current IBGR. The mirror current flows from the power supply voltage VDD and through third bandgap circuit PMOS transistor 348 and second bandgap circuit resistor 352 and third diode 350. The first diode 342, second diode 344, and third diode 350 are thermally matched to have substantially the same voltage versus temperature slopes and the input voltage VIN at 320 is substantially constant over temperature. As the temperature increases, a decrease in the voltage across third diode 350 is offset by the increase in the bandgap reference current IBGR and increase in the voltage across second bandgap circuit resistor 352. As the temperature decreases, an increase in the voltage across third diode 350 is offset by a decrease in the bandgap reference current IBGR and decrease in the voltage across second bandgap circuit resistor 352. Bandgap reference circuit 304 provides a constant, temperature independent input voltage VIN at 320 to the negative input of differential amplifier 308.
Current mirror circuit 306 provides a temperature dependent current ITD via second mirror circuit NMOS transistor 376. Mirror circuit PMOS transistor 372 is biased to pass a mirror current that is substantially equal in value to the bandgap reference current IBGR. The mirror current flows from the power supply voltage VDD and through mirror circuit PMOS transistor 372 and first mirror circuit NMOS transistor 374 to the reference at 380. The current through first mirror circuit NMOS transistor 374 is mirrored through second mirror circuit NMOS transistor 376. The temperature dependent current ITD is directly related to the bandgap reference current IBGR and increases as the temperature increases and decreases as the temperature decreases.
Differential amplifier 308 receives input voltage VIN at 320 and provides an output voltage at 322 that turns on first PMOS transistor 310 and second PMOS transistor 316. Current I1 passes through first PMOS transistor 310 and is divided into a substantially constant current ICC that passes through first resistor 314 and temperature dependent current ITD. The voltage at the drain of first PMOS transistor 310 and at the positive input of differential amplifier 308 is regulated to substantially the value of the constant input voltage VIN by differential amplifier 308. Capacitor 312 stabilizes the feedback loop from the output of differential amplifier 308 to the gate of first PMOS transistor 310 and back to the positive input of differential amplifier 308.
The current I1 is equal to the sum of the constant current ICC and the temperature dependent current ITD. First resistor 314 is part of a constant current source that provides constant current ICC. The constant current ICC is equal to the value of the input voltage VIN divided by the resistance value of first resistor 314. Bandgap reference circuit 304 and current mirror circuit 306 are part of a temperature dependent current source that provides temperature dependent current ITD. As the temperature increases, the temperature dependent current ITD increases and, as the temperature decreases, the temperature dependent current ITD decreases.
The current I1 through first PMOS transistor 310 is mirrored as current I2 through second PMOS transistor 316. Current I2 is substantially equal to current I1 and current I2 passes through second PMOS transistor 316 and second resistor 318 to provide the temperature dependent output voltage VREF at 302. With the reference at 332 at ground, the current I2 times the resistance value of second resistor 318 equals the temperature dependent output voltage VREF at 302.
In operation, as the temperature increases, the temperature dependent current ITD increases. This increases the currents I1 and I2 and the temperature dependent output voltage VREF at 302. As the temperature decreases, the temperature dependent current ITD decreases. This decreases the currents I1 and I2 and the temperature dependent output voltage VREF at 302. The voltage versus temperature slope of the temperature dependent reference voltage VREF is established by setting the resistance value of second resistor 318. The voltage level of the temperature dependent reference voltage VREF is established by setting the resistance value of first resistor 314.
First resistor 314 and second resistor 318 are variable resistors configured to establish the voltage level and the voltage versus temperature slope of the temperature dependent reference voltage VREF. The resistance value of second resistor 318 establishes the slope of the temperature dependent reference voltage VREF, which also changes the voltage level of the temperature dependent reference voltage VREF. The resistance value of first resistor 314 is then configured to establish the voltage level of the temperature dependent reference voltage VREF. The voltage versus temperature slope and the voltage level of the temperature dependent reference voltage VREF are established independently via second resistor 318 and first resistor 314.
In one embodiment, first resistor 314 includes multiple resistors connected in series and each of the multiple resistors can be selected or shorted to provide a resistance value. In one embodiment, second resistor 318 includes multiple resistors connected in series and each of the multiple resistors can be selected or shorted to provide a resistance value. In one embodiment, first resistor 314 is a digitally variable resistor. In one embodiment, second resistor 318 is a digitally variable resistor.
The slope and voltage level of the temperature dependent reference voltage VREF can be established to accommodate circuit needs. In one embodiment, the slope and voltage level of the temperature dependent reference voltage VREF is established to provide a regulated, temperature dependent output voltage VOUT that is used as a supply voltage for delay chains in a RAM. The speed of the delay chains can be made constant over temperature, i.e., temperature independent, via the temperature dependent output voltage VOUT to help ensure that the RAM functions reliably over temperature. The voltage level of the temperature dependent reference voltage VREF can be established independently to conserve power, while the slope of the temperature dependent reference voltage VREF can be established to provide temperature independent delay chains.
Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternate and/or equivalent implementations may be substituted for the specific embodiments shown and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the specific embodiments discussed herein. Therefore, it is intended that this invention be limited only by the claims and the equivalents thereof.

Claims

1. A reference circuit comprising:

a first circuit configured to provide a temperature dependent current;

a second circuit configured to provide a first current; and

a third circuit configured to provide a temperature dependent voltage based on the first current and the temperature dependent current, wherein the temperature dependent voltage has a voltage versus temperature slope established by the third circuit and a voltage level established by the first current.

2. The reference circuit of claim 1, wherein the second circuit provides a second current that comprises the first current and the temperature dependent current and the third circuit is configured to provide a third current that mirrors the second current.

3. The reference circuit of claim 1, wherein the third circuit includes a resistive element configured to receive a current that mirrors the first current and the temperature dependent current and to establish the voltage versus temperature slope of the temperature dependent voltage.

4. The reference circuit of claim 1, wherein the first circuit comprises:

a first resistive element; and

a diode coupled to the first resistive element, wherein the diode has a diode voltage that is dependent on temperature and a constant voltage is applied across the first resistive element and the diode to provide the temperature dependent current.

5. The reference circuit of claim 4, wherein the second circuit includes a second resistive element coupled across the first resistive element and the diode and configured to provide the first current.

6. The reference circuit of claim 1, wherein the first circuit comprises:

a bandgap reference circuit; and

a current mirror coupled to the bandgap reference to provide the temperature dependent current.

7. An electronic system comprising:

a random access memory that includes a reference circuit configured to provide a temperature dependent reference voltage, wherein the reference circuit comprises:

a temperature dependent current source configured to provide a temperature dependent current;

a constant current source configured to provide a constant current; and

a circuit configured to mirror the constant current and the temperature dependent current and provide the temperature dependent reference voltage based on the mirrored constant current and temperature dependent current, wherein the circuit is configured to provide a voltage level of the temperature dependent reference voltage based on the constant current and to establish a voltage versus temperature slope of the temperature dependent reference voltage.

8. The electronic system of claim 7, wherein the random access memory includes a regulator configured to receive the temperature dependent reference voltage and provide a temperature dependent output voltage based on the temperature dependent reference voltage.

9. The electronic system of claim 8, wherein the random access memory includes a temperature dependent delay chain configured to receive the temperature dependent output voltage and provide a substantially constant delay versus temperature.

10. The electronic system of claim 7, wherein the circuit comprises:

a differential amplifier configured to receive a first constant voltage; and

a first transistor configured to be controlled by the differential amplifier to provide a second constant voltage and to provide a first current comprising the temperature dependent current and the constant current.

11. The electronic system of claim 10, wherein the circuit comprises:

a second transistor configured to be controlled by the differential amplifier and to provide a second current that mirrors the first current; and

a resistive element configured to receive the second current to provide the temperature dependent reference voltage and the voltage versus temperature slope of the temperature dependent reference voltage.

12. A reference circuit comprising:

a constant current source configured to provide a constant current;

a differential amplifier configured to receive a first constant voltage and provide an output voltage;

a first transistor configured to receive the output voltage to provide a second constant voltage and to provide a first current comprising the temperature dependent current and the constant current;

a second transistor configured to receive the output voltage to provide a second current that mirrors the first current; and

a first resistive element configured to receive the second current to provide a temperature dependent voltage and a voltage versus temperature slope of the temperature dependent voltage, wherein a voltage level of the temperature dependent voltage is based on the constant current.

13. The reference circuit of claim 12, wherein the constant current source comprises a resistive element configured to receive the second constant voltage to provide the constant current.

14. The reference circuit of claim 12, wherein the temperature dependent current source comprises:

a resistive element; and

a diode coupled to the resistive element.

15. The reference circuit of claim 12, wherein the temperature dependent current source comprises:

a bandgap reference circuit; and

a current mirror coupled to the bandgap reference.

16. A random access memory comprising:

means for providing a temperature dependent current;

means for providing a current to establish a voltage level of a temperature dependent voltage;

means for mirroring the current and the temperature dependent current to provide the temperature dependent voltage; and

means for establishing a voltage versus temperature slope of the temperature dependent voltage.

17. The random access memory of claim 16, wherein the means for providing a temperature dependent current comprises:

means for applying a constant voltage across a resistor and a diode.

18. The random access memory of claim 16, wherein the means for providing a temperature dependent current comprises:

means for mirroring a bandgap reference temperature dependent current.

19. The random access memory of claim 16, comprising:

means for regulating a temperature dependent output voltage based on the temperature dependent voltage.

20. A method for providing a temperature dependent voltage comprising:

providing a temperature dependent current;

providing a current;

mirroring the current and the temperature dependent current to provide the temperature dependent voltage;

establishing a voltage versus temperature slope of the temperature dependent voltage; and

establishing a voltage level of the temperature dependent voltage.

21. The method of claim 20, wherein providing a temperature dependent current comprises:

applying a constant voltage across a resistor and a diode.

22. The method of claim 20, wherein providing a temperature dependent current comprises:

mirroring a bandgap reference temperature dependent current.

23. The method of claim 20, wherein providing a current comprises:

applying a constant voltage across a resistor.

24. The method of claim 20, comprising:

regulating a temperature dependent output voltage based on the temperature dependent voltage.

25. A method for providing a temperature dependent output voltage in a random access memory, comprising:

receiving a temperature dependent current;

receiving a constant current;

combining the temperature dependent current and the constant current;

mirroring the combined current;

receiving the mirrored current at a resistor to provide a voltage level of a temperature dependent reference voltage based on the constant current and to establish a voltage versus temperature slope of the temperature dependent reference voltage.

26. The method of claim 25, comprising:

receiving the temperature dependent reference voltage; and

regulating a temperature dependent output voltage based on the temperature dependent reference voltage.

27. The method of claim 25, comprising:

receiving a first constant voltage at a differential amplifier; and

controlling a first transistor via the differential amplifier to provide a second constant voltage and the combined current.

28. The method of claim 27, wherein mirroring the combined current comprises controlling a second transistor via the differential amplifier to provide a second current that mirrors the combined current.

29. A method for providing a temperature dependent voltage comprising:

receiving a temperature dependent current;

receiving a constant current;

receiving a first constant voltage at a differential amplifier to provide an output voltage;

receiving the output voltage at a first transistor to provide a second constant voltage and a first current comprising the temperature dependent current and the constant current;

receiving the output voltage at a second transistor to provide a second current that mirrors the first current; and

receiving the second current at a first resistive element to provide a temperature dependent voltage and a voltage versus temperature slope of the temperature dependent voltage and to establish a voltage level of the temperature dependent voltage based on the constant current.

30. The method of claim 29, comprising:

receiving the second constant voltage at a second resistive element to provide the constant current.

31. The method of claim 29, comprising:

receiving the second constant voltage across a third resistive element and a diode coupled to the third resistive element to provide the temperature dependent current.

32. The method of claim 29, comprising:

providing the temperature dependent current via a current mirror coupled to a bandgap reference.