US20100173598A1 - Method and system for filter calibration using fractional-n frequency synthesized signals - Google Patents
Method and system for filter calibration using fractional-n frequency synthesized signals Download PDFInfo
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- US20100173598A1 US20100173598A1 US12/724,166 US72416610A US2010173598A1 US 20100173598 A1 US20100173598 A1 US 20100173598A1 US 72416610 A US72416610 A US 72416610A US 2010173598 A1 US2010173598 A1 US 2010173598A1
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/06—Receivers
- H04B1/16—Circuits
- H04B1/30—Circuits for homodyne or synchrodyne receivers
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
- H04B1/40—Circuits
- H04B1/403—Circuits using the same oscillator for generating both the transmitter frequency and the receiver local oscillator frequency
- H04B1/405—Circuits using the same oscillator for generating both the transmitter frequency and the receiver local oscillator frequency with multiple discrete channels
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04H—BROADCAST COMMUNICATION
- H04H60/00—Arrangements for broadcast applications with a direct linking to broadcast information or broadcast space-time; Broadcast-related systems
- H04H60/76—Arrangements characterised by transmission systems other than for broadcast, e.g. the Internet
- H04H60/81—Arrangements characterised by transmission systems other than for broadcast, e.g. the Internet characterised by the transmission system itself
- H04H60/90—Wireless transmission systems
- H04H60/91—Mobile communication networks
Abstract
Description
- This application makes reference to, claims priority to, and claims the benefit of:
- U.S. Provisional Application Ser. No. 60/778,232, filed on Mar. 2, 2006.
- This application also makes reference to:
- U.S. application Ser. No. ______ (Attorney Docket 17540US02) filed on even date herewith;
U.S. application Ser. No. 11/385,390 filed on Mar. 21, 2006; and
U.S. application Ser. No. 11/385,389 filed on Mar. 21, 2006. - Each of the above stated applications is hereby incorporated herein by reference in its entirety.
- Certain embodiments of the invention relate to on-chip RF tuners. More specifically, certain embodiments of the invention relate to a method and system for filter calibration using fractional-N frequency synthesized signals.
- Broadcasting and telecommunications have historically occupied separate fields. In the past, broadcasting was largely an “over-the-air” medium while wired media carried telecommunications. That distinction may no longer apply as both broadcasting and telecommunications may be delivered over either wired or wireless media. Present development may adapt broadcasting to mobility services. One limitation has been that broadcasting may often require high bit rate data transmission at rates higher than could be supported by existing mobile communications networks. However, with emerging developments in wireless communications technology, even this obstacle may be overcome.
- Terrestrial television and radio broadcast networks have made use of high power transmitters covering broad service areas, which enable one-way distribution of content to user equipment such as televisions and radios. By contrast, wireless telecommunications networks have made use of low power transmitters, which have covered relatively small areas known as “cells”. Unlike broadcast networks, wireless networks may be adapted to provide two-way interactive services between users of user equipment such as telephones and computer equipment.
- The introduction of cellular communications systems in the late 1970's and early 1980's represented a significant advance in mobile communications. The networks of this period may be commonly known as first generation, or “1G” systems. These systems were based upon analog, circuit-switching technology, the most prominent of these systems may have been the advanced mobile phone system (AMPS). Second generation, or “2G” systems, ushered improvements in performance over 1G systems and introduced digital technology to mobile communications. Exemplary 2G systems include the global system for mobile communications (GSM), digital AMPS (D-AMPS), and code division multiple access (CDMA). Many of these systems have been designed according to the paradigm of the traditional telephony architecture, often focused on circuit-switched services, voice traffic, and supported data transfer rates up to 14.4 kbits/s. Higher data rates were achieved through the deployment of “2.5G” networks, many of which were adapted to existing 2G network infrastructures. The 2.5G networks began the introduction of packet-switching technology in wireless networks. However, it is the evolution of third generation, or “3G” technology that may introduce fully packet-switched networks, which support high-speed data communications.
- Standards for digital television terrestrial broadcasting (DTTB) have evolved around the world with different systems being adopted in different regions. The three leading DTTB systems are, the advanced standards technical committee (ATSC) system, the digital video broadcast terrestrial (DVB-T) system, and the integrated service digital broadcasting terrestrial (ISDB-T) system. The ATSC system has largely been adopted in North America, South America, Taiwan, and South Korea. This system adapts trellis coding and 8-level vestigial sideband (8-VSB) modulation. The DVB-T system has largely been adopted in Europe, the Middle East, Australia, as well as parts of Africa and parts of Asia. The DVB-T system adapts coded orthogonal frequency division multiplexing (COFDM). The OFDM spread spectrum technique may be utilized to distribute information over many carriers that are spaced apart at specified frequencies. The OFDM technique may also be referred to as multi-carrier or discrete multi-tone modulation. This technique may result in spectral efficiency and lower multi-path distortion, for example. The ISDB-T system has been adopted in Japan and adapts bandwidth segmented transmission orthogonal frequency division multiplexing (BST-OFDM). The various DTTB systems may differ in important aspects; some systems employ a 6 MHz channel separation, while others may employ 7 MHz or 8 MHz channel separations.
- While 3G systems are evolving to provide integrated voice, multimedia, and data services to mobile user equipment, there may be compelling reasons for adapting DTTB systems for this purpose. One of the more notable reasons may be the high data rates that may be supported in DTTB systems. For example, DVB-T may support data rates of 15 Mbits/s in an 8 MHz channel in a wide area single frequency network (SFN). There are also significant challenges in deploying broadcast services to mobile user equipment. Because of form factor constraints, many handheld portable devices, for example, may require that PCB area be minimized and that services consume minimum power to extend battery life to a level that may be acceptable to users. Another consideration is the Doppler Effect in moving user equipment, which may cause inter-symbol interference in received signals. Among the three major DTTB systems, ISDB-T was originally designed to support broadcast services to mobile user equipment. While DVB-T may not have been originally designed to support mobility broadcast services, a number of adaptations have been made to provide support for mobile broadcast capability. The adaptation of DVB-T to mobile broadcasting is commonly known as DVB handheld (DVB-H). The broadcasting frequencies for Europe are in UHF (bands IVN) and in the US, the 1670-1675 MHz band that has been allocated for DVB-H operation. Additional spectrum is expected to be allocated in the L-band world-wide.
- To meet requirements for mobile broadcasting the DVB-H specification supports time slicing to reduce power consumption at the user equipment, addition of a 4K mode to enable network operators to make tradeoffs between the advantages of the 2K mode and those of the 8K mode, and an additional level of forward error correction on multi-protocol encapsulated data—forward error correction (MPE-FEC) to make DVB-H transmissions more robust to the challenges presented by mobile reception of signals and to potential limitations in antenna designs for handheld user equipment. DVB-H may also use the DVB-T modulation schemes, like QPSK and 16-quadrature amplitude modulation (16-QAM).
- While several adaptations have been made to provide support for mobile broadcast capabilities in DVB-T, concerns regarding device size, cost, and/or power requirements still remain significant constraints for the implementation of handheld portable devices enabled for digital video broadcasting operations. For example, typical DVB-T tuners or receivers in mobile terminals may employ super-heterodyne architectures with one or two intermediate frequency (IF) stages and direct sampling of the passband signal for digital quadrature down-conversion. Moreover, external tracking and SAW filters may generally be utilized for channel selection and image rejection. Such approaches may result in increased power consumption and high external component count, which may limit their application in handheld portable devices. As a result, the success of mobile broadcast capability of DVB-T may depend in part on the ability to develop TV tuners that have smaller form factor, are produced at lower cost, and consume less power during operation. Furthermore, process and temperature variations within conventional tuners or receivers in mobile terminals result in deviation in the characteristics of many sub-circuits of the transceiver. A very important case is the deviation of the frequency response of analog filters used within the tuners or receivers. Such deviation of the frequency response results in deterioration of channel selection capabilities of the tuners or receivers.
- As mobile terminals support a wider range of content from voice to data to video, they may be required to receive a correspondingly wider range of frequencies. Consequently, filtering circuitry may be required to filter signals for correspondingly wider ranges of frequencies.
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FIG. 1 is diagram for a conventional filter calibration scheme utilizing a matched oscillator. This is an indirect filter calibration techniques, meaning that filter bandwidth is calibrated through the calibration of a circuit other than the filter itself (the oscillator). Referring toFIG. 1 , there is shown afilter 202, an oscillator 204, acrystal oscillator 206, afrequency divider block 208, an exclusive-or (XOR) block 210, and acontrol block 212. - The oscillator 204 may comprise suitable logic, circuitry, and/or code that may enable generation of a clock signal. The oscillator 204 may comprise resistive (R) and capacitive (C) components. The R and C components may be variable or fixed. When the frequency associated with the clock signal is based on the values for the R and C components, the oscillator 204 may comprise an RC oscillator circuit.
- The oscillator 204 may comprise active components, for example operational amplifier (op-amp) and C components. The op-amp component may comprise one or more electrical devices characterized by one or more transconductance (Gm) values. The C component may comprise one or more electrical devices characterized by one or more fixed or variable capacitive values. When the frequency associated with the clock signal is based on the values for the op-amp and C components, the oscillator 204 may comprise a GmC oscillator circuit.
- The
frequency divider block 208 may comprise suitable logic, circuitry, and/or code that may enable generation of an output signal based on an input signal, wherein the input signal is characterized by a frequency that is a multiple of the corresponding frequency of the output signal. The value of each corresponding frequency may be determined by thefrequency divider block 208. - The
XOR block 210 may comprise suitable logic, circuitry, and/or code that may enable generation of an output signal in which the value of the output signal is based on a comparison of respective values associated with two input signals. TheXOR block 210 may output a LOW value when the respective values of the two input signals are approximately equal. TheXOR block 210 may output a HIGH value when the respective values of the two input signals are not approximately equal. - The
control block 212 may comprise suitable logic, circuitry, and/or code that may enable generating a control signal, fControl, based on an input signal. The control signal may comprise an analog signal, such as a value for a voltage or a current for example, based on the input signal. The control signal may comprise a digital representation comprising one or more bits for example, based on the input signal. Thecontrol block 212 may receive an input signal from an external circuit. Thecontrol block 212 may generate the control signal based on the input signal. The control signal may be communicated to control at least a portion of the circuitry from which the input signal was received. - In operation, the
crystal oscillator 206 may enable generation of a crystal (xtal) timing signal. The crystal timing signal may be characterized by a crystal frequency, fXtal. Thefrequency divider 208 may receive the crystal timing signal as an input signal. Thefrequency divider 208 may utilize a frequency division factor, fp, to generate a reference timing signal characterized by a reference frequency, fRef, and a reference phase φRef. The value of the reference frequency may be about equal to the ratio of the value of the reference frequency and the value of the frequency division factor, fRef/fD. - The oscillator 204 may enable generation of an oscillator timing signal characterized by an oscillator frequency, fOsc, and an oscillator phase φOsc. For an oscillator 204 comprising an RC oscillator, the oscillator frequency may be referred to as an RC oscillator frequency, fOsc(RC). The corresponding oscillator phase may be referred to as an RC oscillator phase, φOsc(RC). The value of the RC oscillator frequency and/or phase may be based on values for the R and C components. The values for the R and/or C components may be determined based on the control signal fControl.
- The
XOR block 210 may concurrently compare a value for the reference timing signal and a corresponding value for the oscillator timing signal at various time instants. Based on the comparison, the XOR block 210 may generate a difference signal. The difference signal may be nonzero when there are differences between the frequencies fRef and fOsc(RC), at a given time instant. The difference signal may be nonzero when there are differences between the phases φRef and φOsc(RC), at a given time instant. - The
control block 212 may receive the difference signal and generate the control signal, fControl, based on the value of the difference signal. Thecontrol block 212 may communicate the control signal, comprising feedback information, to the oscillator 204. The feedback information may cause the oscillator 204 to adjust the R and/or C values. As a result of the adjustment, the corresponding frequency and/or phase values, fOsc(RC)/φOsc(RC), may be adjusted. - For an oscillator 204 comprising an GmC oscillator, the oscillator frequency may be referred to as an GmC oscillator frequency, fOsc(GmC). The corresponding oscillator phase may be referred to as an GmC oscillator phase, φOsc(GmC). The value of the GmC oscillator frequency and/or phase may be based on values for the op-amp and C components. The values for the op-amp and/or C components may be determined based on the control signal fControl.
- The difference signal generated by the XOR block 210 may be nonzero when there are differences between the frequencies fRef and fOsc(GmC), at a given time instant. The difference signal may be nonzero when there are differences between the phases φRef and φOsc(GmC), at a given time instant.
- The
control block 212 may receive the difference signal and generate the control signal, fControl, based on the value of the difference signal. Thecontrol block 212 may communicate the control signal, comprising feedback information, to the oscillator 204. The feedback information may cause the oscillator 204 to adjust the Gm and/or C values. As a result of the adjustment, the corresponding frequency and/or phase values, fOsc(GmC)/φOsc(GmC), may be adjusted. - The oscillator 204 may utilize shared or common components with the
filter 202. For example, for an oscillator 204 that comprises R and C components, thefilter 202 may comprise equivalent R and C components. When the value for the f−3dB filter cut-off frequency is based on the values of the R and C components, thefilter 202 may comprise an RC filter circuit. For an oscillator 204 that comprises op-amp components and C components, thefilter 202 may comprise equivalent op-amp and C components. When the value for the f−3dB filter cut-off frequency is based on the values of the op-amp and C components, thefilter 202 may comprise a GmC filter circuit. - For a
filter 202 comprising an RC filter circuit, the control signal, fControl, generated by thecontrol block 212 may cause thefilter 202 to adjust the R and/or C values for the equivalent R and/or C components. As a result of the adjustment, the corresponding value for the f−3dB filter cut-off frequency may be adjusted. For afilter 202 comprising a GmC filter circuit, the control signal, fControl, generated by thecontrol block 212 may cause thefilter 202 to adjust the Gm and/or C values for the equivalent op-amp and/or C components. As a result of the adjustment, the corresponding value for the f−3dB filter cut-off frequency may be adjusted. - The oscillator 204 may be utilized to calibrate the
filter 202 since the control signal, fControl, is generated based on the oscillator frequency fOsc, and/or oscillator phase φOsc. The control signal may cause thefilter 202 to compute a value for the f−3dB filter cut-off frequency. A disadvantage in this method is that the accuracy of the calibration may be limited based on the extent to which the values for the R and C components in the oscillator 204 are equal to corresponding values for the equivalent R and C components in thefilter 202, for a given value of the control signal fControl. The accuracy of the calibration may also be limited based on the range of values for frequency, fRef, and/or phase, φRef, which may be generated by thefrequency divider block 208. - Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings.
- A system and/or method is provided for filter calibration using fractional-N frequency synthesized signals, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims.
- These and other advantages, aspects and novel features of the present invention, as well as details of an illustrated embodiment thereof, will be more fully understood from the following description and drawings.
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FIG. 1 is diagram for a conventional filter calibration scheme utilizing a matched oscillator. -
FIG. 2A is a block diagram illustrating an exemplary mobile terminal, in accordance with an embodiment of the invention. -
FIG. 2B is a block diagram illustrating exemplary communication between a multi-band RF receiver and a digital baseband processor in a mobile terminal, in accordance with an embodiment of the invention. -
FIG. 2C is a block diagram illustrating an exemplary single-chip multi-band RF receiver with an integrated LNA in each front-end, in accordance with an embodiment of the invention. -
FIG. 2D is a diagram for an exemplary direct filter calibration scheme utilizing filtering, which may be utilized in connection with an embodiment of the invention. -
FIG. 3A is a block diagram of an exemplary analog baseband processing system supporting auto-calibration, in accordance with an embodiment of the invention. -
FIG. 3B is a schematic diagram of an exemplary opamp-RC baseband filter that may be used in accordance with an embodiment of the invention. -
FIG. 3C is a block diagram of an exemplary baseband processing system using opamp-RC filters and an auto-calibration loop, in accordance with an embodiment of the invention. -
FIG. 4 is a flow diagram illustrating exemplary steps in a filter calibration scheme using fractional-N frequency synthesized signals, in accordance with an embodiment of the invention. - Certain aspects of the invention provide a method and system for filter calibration using fractional-N frequency synthesized signals. Aspects of the method may comprise generating a LO signal by a PLL circuit within a chip. A reference signal may be generated based on the generated LO signal and a synthesizer control signal. The method may further comprise calibrating a frequency response for a filter circuit integrated within the chip by adjusting parameters associated with the filter circuit based on the generated reference signal. Aspects of the system may include a single-chip multi-band RF receiver that enables generation of a LO signal by a PLL circuit within the single-chip. A reference signal may be generated based on the generated LO signal and a synthesizer control signal. The single-chip multi-band RF receiver may enable calibration of a frequency response for a filter circuit integrated within the chip. The frequency response may be calibrated by adjusting the filter based on the generated reference signal.
- Various embodiments of the invention may comprise a scheme for calibrating a filter in a communication receiver without requiring additional circuitry. In such embodiments, a Σ−Δ fractional-N synthesizer may be utilized for synthesizing RF signals. The Σ−Δ fractional-N synthesizer may enable the generation of a wide range of reference signal frequencies. In addition, the Σ−Δ fractional-N synthesizer may enable more accurate generation of a specified filter cutoff frequency f−3dB than may be the case with many conventional filter calibration schemes. Various embodiments of the invention may utilize a digital frequency synthesizer to enable even greater accuracy in the generation of a specific frequency f−3dB.
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FIG. 2A is a block diagram illustrating an exemplary mobile terminal, in accordance with an embodiment of the invention. Referring toFIG. 2A , there is shown amobile terminal 120 that may comprise anRF receiver 123 a, anRF transmitter 123 b, a digital baseband processor 129, a processor 125, and a memory 127. A receiveantenna 121 a may be communicatively coupled to theRF receiver 123 a. A transmitantenna 121 b may be communicatively coupled to theRF transmitter 123 b. Themobile terminal 120 may be operated in a system, such as the cellular network and/or digital video broadcast network described inFIG. 2A , for example. - The
RF receiver 123 a may comprise suitable logic, circuitry, and/or code that may enable processing of received RF signals. TheRF receiver 123 a may enable receiving RF signals in a plurality of frequency bands. For example, theRF receiver 123 a may enable receiving DVB-H transmission signals via the VHF band, from about 174 MHz to about 240 MHz, the UHF band, from about 470 MHz to about 890 MHz, the 1670-1675 MHz band, and/or the L-band, from about 1400 MHz to about 1700 MHz, for example. Moreover, theRF receiver 123 a may enable receiving signals in cellular frequency bands, for example. Each frequency band supported by theRF receiver 123 a may have a corresponding front-end circuit for handling low noise amplification and down conversion operations, for example. In this regard, theRF receiver 123 a may be referred to as a multi-band receiver when it supports more than one frequency band. In another embodiment of the invention, themobile terminal 120 may comprise more than oneRF receiver 123 a, wherein each of theRF receivers 123 a may be a single-band or a multi-band receiver. - The
RF receiver 123 a may quadrature down convert the received RF signal to a baseband frequency signal that comprises an in-phase (I) component and a quadrature (Q) component. TheRF receiver 123 a may perform direct down conversion of the received RF signal to a baseband frequency signal, for example. In some instances, theRF receiver 123 a may enable analog-to-digital conversion of the baseband signal components before transferring the components to the digital baseband processor 129. In other instances, theRF receiver 123 a may transfer the baseband signal components in analog form. - The digital baseband processor 129 may comprise suitable logic, circuitry, and/or code that may enable processing and/or handling of baseband frequency signals. In this regard, the digital baseband processor 129 may process or handle signals received from the
RF receiver 123 a and/or signals to be transferred to theRF transmitter 123 b, when theRF transmitter 123 b is present, for transmission to the network. The digital baseband processor 129 may also provide control and/or feedback information to theRF receiver 123 a and to theRF transmitter 123 b based on information from the processed signals. The digital baseband processor 129 may communicate information and/or data from the processed signals to the processor 125 and/or to the memory 127. Moreover, the digital baseband processor 129 may receive information from the processor 125 and/or to the memory 127, which may be processed and transferred to theRF transmitter 123 b for transmission to the network. - The
RF transmitter 123 b may comprise suitable logic, circuitry, and/or code that may enable processing of RF signals for transmission. TheRF transmitter 123 b may enable transmission of RF signals in a plurality of frequency bands. Moreover, theRF transmitter 123 b may enable transmitting signals in cellular frequency bands, for example. Each frequency band supported by theRF transmitter 123 b may have a corresponding front-end circuit for handling amplification and up conversion operations, for example. In this regard, theRF transmitter 123 b may be referred to as a multi-band transmitter when it supports more than one frequency band. In another embodiment of the invention, themobile terminal 120 may comprise more than oneRF transmitter 123 b, wherein each of theRF transmitters 123 b may be a single-band or a multi-band transmitter. - The
RF transmitter 123 b may quadrature up convert the baseband frequency signal comprising I/Q components to an RF signal. TheRF transmitter 123 b may perform direct up conversion of the baseband frequency signal to a baseband frequency signal, for example. In some instances, theRF transmitter 123 b may enable digital-to-analog conversion of the baseband signal components received from the digital baseband processor 129 before up conversion. In other instances, theRF transmitter 123 b may receive baseband signal components in analog form. - The processor 125 may comprise suitable logic, circuitry, and/or code that may enable control and/or data processing operations for the
mobile terminal 120. The processor 125 may be utilized to control at least a portion of theRF receiver 123 a, theRF transmitter 123 b, the digital baseband processor 129, and/or the memory 127. In this regard, the processor 125 may generate at least one signal for controlling operations within themobile terminal 120. The processor 125 may also enable executing of applications that may be utilized by themobile terminal 120. For example, the processor 125 may execute applications that may enable displaying and/or interacting with content received via DVB-H transmission signals in themobile terminal 120. - The memory 127 may comprise suitable logic, circuitry, and/or code that may enable storage of data and/or other information utilized by the
mobile terminal 120. For example, the memory 127 may be utilized for storing processed data generated by the digital baseband processor 129 and/or the processor 125. The memory 127 may also be utilized to store information, such as configuration information, that may be utilized to control the operation of at least one block in themobile terminal 120. For example, the memory 127 may comprise information necessary to configure theRF receiver 123 a to enable receiving DVB-H transmission in the appropriate frequency band. -
FIG. 2B is a block diagram illustrating exemplary communication between a multi-band RF receiver and a digital baseband processor in a mobile terminal, in accordance with an embodiment of the invention. Referring toFIG. 2B , there is shown amulti-band RF receiver 130, an analog-to-digital converter (ADC) 134, and a digital baseband processor 132. Themulti-band RF receiver 130 may comprise a UHF front-end 131 a, an L-band front-end 131 b, a VHF front-end 131 c, abaseband block 133 a, a received signal strength indicator (RSSI) block 133 b, and asynthesizer 133 c. Themulti-band RF receiver 130, the analog-to-digital converter (ADC) 134, and/or the digital baseband processor 132 may be part of a mobile terminal, such as themobile terminal 120 inFIG. 2A , for example. - The
multi-band RF receiver 130 may comprise suitable logic, circuitry, and/or code that may enable handling of VHF, UHF and L-band signals. Themulti-band RF receiver 130 may be enabled via an enable signal, such as thesignal RxEN 139 a, for example. In this regard, enabling themulti-band RF receiver 130 via thesignal RxEN 139 a by a 1:10 ON/OFF ratio may allow time slicing in DVB-H while reducing power consumption. At least a portion of the circuitry within themulti-band RF receiver 130 may be controlled via thecontrol interface 139 b. Thecontrol interface 139 b may receive information from, for example, a processor, such as the processor 125 inFIG. 2A , or from the digital baseband processor 132. Thecontrol interface 139 b may comprise more than one bit. For example, when implemented as a 2-bit interface, thecontrol interface 139 a may be an inter-integrated circuit (12C) interface. - The VHF front-
end 131 c may comprise suitable logic, circuitry, and/or code that may enable low noise amplification and direct down conversion of VHF signals. In this regard, the VHF front-end 131 c may utilize an integrated low noise amplifier (LNA) and mixers, such as passive mixers, for example. The VHF front-end 131 c may communicate the resulting baseband frequency signals to the baseband block 133 a for further processing. - The UHF front-
end 131 a may comprise suitable logic, circuitry, and/or code that may enable low noise amplification and direct down conversion of UHF signals. In this regard, the UHF front-end 131 a may utilize an integrated low noise amplifier (LNA) and mixers, such as passive mixers, for example. The UHF front-end 131 a may communicate the resulting baseband frequency signals to the baseband block 133 a for further processing. - The L-band front-
end 131 b may comprise suitable logic, circuitry, and/or code that may enable low noise amplification and direct down conversion of L-band signals. In this regard, the L-band front-end 131 b may utilize an integrated LNA and mixers, such as passive mixers, for example. The L-band front-end 131 b may communicate the resulting baseband frequency signals to the baseband block 133 a for further processing. Themulti-band RF receiver 130 may enable one of the VHF front-end 131 c, the UHF front-end 131 a and the L-band front-end 131 b based on current communication conditions. - The
synthesizer 133 c may comprise suitable logic, circuitry, and/or code that may enable generating the appropriate local oscillator (LO) signal for performing direct down conversion in either the VHF front-end 131 c, the UHF front-end 131 a or the L-band front-end 131 b. Since thesynthesizer 133 c may enable fractional division of a source frequency when generating the LO signal, a large range of crystal oscillators may be utilized as a frequency source for thesynthesizer 133 c. This approach may enable the use of an existing crystal oscillator in a mobile terminal PCB, thus reducing the number of external components necessary to support the operations of themulti-band RF receiver 130, for example. Thesynthesizer 133 c may generate a common LO signal for the VHF front-end 131 c, the UHF front-end 131 a and for the L-band front-end 131 b. In this regard, the VHF front-end 131 c, the UHF front-end 131 a and the L-band front-end 131 b may enable dividing the LO signal in order to generate the appropriate signal to perform down conversion from the VHF band, from the UHF band and from the L-band respectively. In some instances, thesynthesizer 133 c may have at least one integrated voltage controlled oscillator (VCO) for generating the LO signal. In other instances, the VCO may be implemented outside thesynthesizer 133 c. - The baseband block 133 a may comprise suitable logic, circuitry, and/or code that may enable processing of I/Q components generated from the direct down conversion operations in the VHF front-
end 131 c, the UHF front-end 131 a and the L-band front-end 131 b. The baseband block 133 a may enable amplification and/or filtering of the I/Q components in analog form. The baseband block 133 a may communicate the processed I component, that is, signal 135 a, and the processed Q component, that is, signal 135 c, to the ADC 134 for digital conversion. - The
RSSI block 133 b may comprise suitable logic, circuitry, and/or code that may enable measuring the strength, that is, the RSSI value, of a received RF signal, whether VHF, UHF or L-band signal. The RSSI measurement may be performed, for example, after the received RF signal is amplified in either the VHF front-end 131 c, the UHF front-end 131 a or the L-band front-end 131 b. TheRSSI block 133 b may communicate the analog RSSI measurement that is, signal 135 e, to the ADC 134 for digital conversion. - The ADC 134 may comprise suitable logic, circuitry, and/or code that may enable digital conversion of
signals signals multi-band RF receiver 130 or into the digital baseband processor 132. - The digital baseband processor 132 may comprise suitable logic, circuitry, and/or code that may enable processing and/or handling of baseband frequency signals. In this regard, the digital baseband processor 132 may be the same or substantially similar to the digital baseband processor 129 described in
FIG. 2A . The digital baseband processor 132 may enable generating at least one signal, such as thesignals AGC_BB 137 a andAGC_RF 137 b, for adjusting the operations of themulti-band RF receiver 130. For example, thesignal AGC_BB 137 a may be utilized to adjust the gain provided by the baseband block 133 a on the baseband frequency signals generated from either the VHF front-end 131 c, the UHF front-end 131 a or the L-band front-end 131 b. In another example, thesignal AGC_RF 137 b may be utilized to adjust the gain provided by an integrated LNA in either the VHF front-end 131 c, the UHF front-end 131 a or the L-band front-end 131 b. In another example, the digital baseband processor 132 may generate at least one control signal or control information communicated to themulti-band RF receiver 130 via thecontrol interface 139 b for adjusting operations within themulti-band RF receiver 130. -
FIG. 2C is a block diagram illustrating an exemplary single-chip multi-band RF receiver with an integrated LNA in each front-end, in accordance with an embodiment of the invention. Referring toFIG. 2C , there is shown a single-chipmulti-band RF receiver 140 a that may comprise a VHF front-end 148 c, a UHF front-end 148 a, an L-band front-end 148 b, abaseband block 164, adigital frequency synthesizer 188, a logarithmic amplifier (logarithmic amplifier) 172, a EA fractional-N synthesizer 174, aVCO block 176, adigital interface 160, anADC 162, anoscillator 180, and abuffer 182. - The single-chip
multi-band RF receiver 140 a may be fabricated using any of a plurality of semiconductor manufacturing processes, for example, complimentary metal-oxide-semiconductor (CMOS) processes, bipolar CMOS (BiCMOS), or Silicon Germanium (SiGe). The single-chipmulti-band RF receiver 140 a may be implemented using differential structures to minimize noise effects and/or substrate coupling, for example. The single-chipmulti-band RF receiver 140 a may utilize low drop out (LDO) voltage regulators to regulate and clean up on-chip voltage supplies. In this regard, the LDO voltage regulators may be utilized to transform external voltage sources to the appropriate on-chip voltages. - When the single-chip
multi-band RF receiver 140 a is implemented utilizing a CMOS process, some design considerations may include achieving low noise figure (NF) values, wide-band operation, high signal-to-noise ration (SNR), performing DC offset removal, achieving high input second-order and third-order intercept points (IIP2 and IIP3), and/or reducing I/Q mismatch, for example. - The single-chip
multi-band RF receiver 140 a may receive UHF signals via afirst antenna 142 a, aUHF filter 144 a, and afirst balum 146 a. TheUHF filter 144 a enables band pass filtering, wherein the band pass may be about 470 to about 702 MHz for cellular signals, for example, or about 470 to about 862 MHz, for other types of received signals, for example. The balum 146 a enables balancing the filtered signals before being communicated to the UHF front-end 148 a. - The single-chip
multi-band RF receiver 140 a may receive L-band signals via asecond antenna 142 b, an L-band filter 144 b, and asecond balum 146 b. The L-band filter 144 b enables band pass filtering, wherein the band pass may be about 1670 to about 1675 MHz for signals in US systems, for example, or about 1450 to about 1490 MHz, for signals in European systems, for example. Thebalum 146 b enables balancing the filtered signals before being communicated to the L-band front-end 148 a. In some instances,antennas multi-band RF receiver 140 a that may support receiving radio signals operating in the UHF IV/V and/or L-band, for example. - The single-chip
multi-band RF receiver 140 a may receive VHF signals via athird antenna 142 c, aVHF filter 144 c, and athird balum 146 c. TheVHF filter 144 c enables band pass filtering, wherein the band pass may be about 174 to about 240 MHz, for example. Thebalum 146 c enables balancing the filtered signals before being communicated to the VHF front-end 148 c. - The UHF front-
end 148 a may comprise a variable low noise amplifier (LNA) 150 a, amixer 152 a, amixer 154 a, and aLO signal divider 156. Thevariable LNA 150 a may comprise suitable logic and/or circuitry that may enable amplification of the UHF signals received. Matching between the output of the balum 146 a and the input of thevariable LNA 150 a may be achieved by utilizing off-chip series inductors, for example. Thevariable LNA 150 a may implement continuous gain control by current steering that may be controlled by a replica scheme within thevariable LNA 150 a. The gain of thevariable LNA 150 a may be adjusted via thesignal AGC_RF 137 b, for example. - The
mixers divider block 156. Themixers LO signal divider 156 may comprise suitable logic, circuitry, and/or code that may enable dividing of the LO signal 186 by a factor of 2 (:/2) or a factor of 3 (:/3) and at the same time providequadrature outputs 186I and 186Q, wherein 186I and 186Q have 90 degrees separation between them. The factor of 3 division may be used when the received UHF signal band is about 470 to about 600 MHz, for example. The factor of 2 division may be used when the received UHF signal band is about 600 to about 900 MHz, for example. The I/Q components generated by themixers baseband block 164. - The L-band front-
end 148 b may comprise avariable LNA 150 b, amixer 152 a, amixer 154 a, and aLO signal generator 158. Thevariable LNA 150 a may comprise suitable logic and/or circuitry that may enable amplification of the L-band signals received. Matching between the output of thebalum 146 b and the input of thevariable LNA 150 b may be achieved by utilizing off-chip series inductors, for example. Thevariable LNA 150 b may implement continuous gain control by current steering that may be controlled by a replica scheme within thevariable LNA 150 b. The gain of thevariable LNA 150 b may be adjusted via thesignal AGC_RF 137 b, for example. - The
mixers LO generator block 158. Themixers LO signal generator 158 may comprise suitable logic, circuitry, and/or code that may enable generation of quadrature LO signals 158I and 158Q, that is, signals with 90 degree phase split between them, from theLO signal 186. The I/O components generated by themixers baseband block 164. - The VHF front-end 148 c may comprise a variable low noise amplifier (LNA) 150 c, a
mixer 152 c, a mixer 154 c, and aLO signal divider 157. Thevariable LNA 150 c may comprise suitable logic and/or circuitry that may enable amplification of the VHF signals received. Matching between the output of thebalum 146 c and the input of thevariable LNA 150 c may be achieved by utilizing off-chip series inductors, for example. Thevariable LNA 150 c may implement continuous gain control by current steering that may be controlled by a replica scheme within thevariable LNA 150 c. The gain of thevariable LNA 150 c may be adjusted via thesignal AGC_RF 137 b, for example. - The
mixers 152 c and 154 c may comprise suitable logic and/or circuitry that may enable generating in-phase (I) and quadrature (Q) components of the baseband frequency signal based on direct down conversion of the amplified received VHF signal with the quadrature signals 187I and 187Q generated by thedivider block 157. Themixers 152 c and 154 c may be passive mixers in order to achieve high linearity and/or low flicker noise, for example. TheLO signal divider 157 may comprise suitable logic, circuitry, and/or code that may enable dividing of the LO signal 187 by a factor of 6 (:/6) or a factor of 8 (:/8) and at the same time provide quadrature outputs 187I and 187Q, wherein 187I and 187Q have 90 degrees separation between them. The I/Q components generated by themixers 152 c and 154 c may be communicated to thebaseband block 164. - The
digital frequency synthesizer 188 may comprise suitable logic, circuitry, and/or code that may enable generation of a reference signal based on a clock timing signal, and on a control input signal. In various embodiments of the invention, thedigital frequency synthesizer 188 may implement a look up table (LUT) function wherein a given clock timing signal and control input signal combination may correspond to a frequency, phase, and/or magnitude for a generated reference signal. Data utilized for the LUT function may be stored and/or retrieved from the memory 127 (FIG. 2A ), for example. In other embodiments of the invention, thedigital frequency synthesizer 188 may comprise an over-sampling digital to analog conversion (DAC) function in which thedigital frequency synthesizer 188 performs digital sampling of the clock timing signal. A rate of digital sampling may be determined based on the control input signal. - The logarithmic amplifier 172 may comprise suitable logic, circuitry, and/or code that may enable generation of a wideband, received signal strength indicator (RSSI) signal, such as the
signal 135 e, based on the output of thevariable LNA 150 a. The RSSI signal indicates the total amount of signal power that is present at the output of the LNA, for example. The RSSI signal may be utilized by, for example, the digital baseband processor 132 inFIG. 2C , to adjust the gain of thevariable LNA 150 a in the presence of RF interference to achieve NF and/or linearity performance that meets blocking and/or intermodulation specifications, for example. In this regard, interference may refer to blocker signals, for example. Blocker signals may be unwanted signals in frequency channels outside the wanted or desired channel that may disturb the reception of the wanted signals. This effect may be a result of blockers generating large signals within the receiver path. These large signals may introduce harmonics, intermodulation products, and/or unwanted mixing products that crosstalk with the wanted signals. In another embodiment of the invention, the logarithmic amplifier 172 may enable generating a wideband, RSSI signal, such as thesignal 135 e, based on the output of thevariable LNA 150 b. In this instance, the RSSI signal may be utilized by to adjust the gain of thevariable LNA 150 b. - The
baseband block 164 may comprise an in-phase component processing path and a quadrature component processing path. The in-phase processing path may comprise at least one programmable gain amplifier (PGA) 166 a, abaseband filter 168 a, and at least onePGA 170 a. The quadrature component processing path may comprise at least onePGA 166 b, abaseband filter 168 b, and at least onePGA 170 b. ThePGAs PGAs PGAs multi-band RF receiver 140 a, thePGAs PGAs ADC 162 may be utilized to provide digital control of thePGAs - The baseband filters 168 a and 168 b may comprise suitable logic, circuitry, and/or code that may enable channel selection, for example. Channel selection may be performed by filters, such as an Nth order lowpass Chebyschev filter implemented by opamp-RC active integrators in a leapfrog configuration, for example. For the correct tuning of the characteristics of the filters, an on-chip auto-calibration loop may be activated upon power-up. The auto-calibration loop may set up the cut-off frequency, f−3dB to the correct vale required to meet the requirements of the communications standard for which the receiver is designed. For example, in DVB-T/DVB-H, the value f−3dB of the filter cut-off frequency may be set to a value from 2 to 5 MHz thus supporting the different channel bandwidths of 5-8 MHz specified by DVB-T/DVB-H standards. During auto-calibration, a tone at the appropriate f−3dB may be generated by the
digital frequency synthesizer 188 and may be applied at the input of the baseband filters 168 a and 168 b for comparison with the filter output of a root-mean-squared (RMS) detector. A digitally controlled loop may be utilized to adjust the baseband filter bandwidth until the output of the baseband filter and the RMS detector are the same. - The Σ−Δ fractional-N synthesizer 174 may comprise suitable logic, circuitry, and/or code that may enable LO generation that may be independent of the reference crystal frequency, such as the
crystal 178, for example. In this regard, the synthesizer 174 may generate a signal, such as thesignal 190, for example, to control the operation of theVCO block 176 and therefore the generation of theLO signal 186. Since the synthesizer 174 may enable fractional synthesis, the single-chip multiband RF receiver 140 a may utilize the same crystal utilized by other operations in the mobile terminal while maintaining fine tuning capability. The synthesizer 174 may receive a reference frequency signal from thecrystal 178 via anoscillator 180, for example. The output of theoscillator 180 may also be buffered by thebuffer 182 to generate aclock signal 184, for example. - The
VCO block 176 may comprise suitable logic, circuitry, and/or code that may enable generating the LO signal 186 utilized by the VHF front-end 148 c, the UHF front-end 148 a and the L-band front-end 148 b for direct down conversion of the received RF signals. TheVCO block 176 may comprise at least one VCO, wherein each VCO may have cross-coupled NMOS and PMOS devices and metal-oxide-semiconductor (MOS) varactors in an accumulation mode for tuning. In this regard, a switched varactor bank may be utilized for providing coarse tuning. TheVCO block 176 may provide a range of about 1.2 to about 1.8 GHz when implemented utilizing two VCOs, for example. When more than one VCO is utilized in implementing theVCO block 176, selecting the proper VCO for generating theLO signal 186 may be based on the type of RF signal being received by the single-chip multiband RF receiver 140 a. - The
digital interface 160 may comprise suitable logic, circuitry, and/or code that may enable controlling circuitry within the single-chip multiband RF receiver 140 a. Thedigital interface 160 may comprise a plurality of registers for storing control and/or operational information for use by the single-chipmulti-band RF receiver 140 a. Thedigital interface 160 may enable receiving thesignal RxEN 139 a that may be utilized to perform 1:10 ON/OFF ratio time slicing in DVB-H while reducing power consumption. Moreover, thedigital interface 160 may enable receiving thecontrol interface 139 b from, for example, a processor, such as the processor 125 inFIG. 2A , or from the digital baseband processor 132 inFIG. 2C . Thecontrol interface 139 b may comprise more than one bit. Thecontrol interface 139 b may be utilized to control the synthesis operations of the synthesizer 174 and/or the filtering operations of the baseband filters 168 a and 168 b. Thecontrol interface 139 b may also be utilized to adjust the bias of circuits within the single-chipmulti-band RF receiver 140 a, such as those of thevariable LNAs PGAs -
FIG. 2D is a diagram for an exemplary direct filter calibration scheme utilizing filtering, which may be utilized in connection with an embodiment of the invention. Referring toFIG. 2D , there is shown acrystal oscillator 206, afrequency divider block 208, a filtering block 222, a filter block 224, and acontrol block 226. Thecrystal oscillator block 206 may substantially comprise the functions of thecrystal 178, and the oscillator 180 (FIG. 2C ). Thefrequency divider block 208 may be substantially as described inFIG. 1 . Thecontrol block 226 may be substantially as described for thecontrol block 212 inFIG. 1 . The filter block 224 may substantially comprise the functions of thebaseband filter 168 a. When the value for the f−3dB filter cut-off frequency is based on the values of R and C components, the filter 224 may comprise an opamp-RC filter circuit. For the opamp-RC filter circuit, the f−3dB filter cut-off frequency may be referred to as a f−3dB (RC) filter cut-off frequency. When the value for the f−3dB filter cut-off frequency is based on the values of transconductors and C components, the filter 224 may comprise a GmC filter circuit. For the RC filter circuit, the f−3dB filter cut-off frequency may be referred to asa f —3 dB (GmC) filter cut-off frequency. - The filtering block 222 may comprise suitable logic, circuitry, and/or code that may enable generation of an output signal by removing harmonic frequency components from a received input signal. The function of the filtering block 222 may be practiced by implementations comprising a low pass filter, a high pass filter, and/or a band pass filter.
- In operation, the
crystal oscillator 206 may enable generation of a crystal (xtal) timing signal. The crystal timing signal may be characterized by a crystal frequency, fXtal. Thefrequency divider 208 may receive the crystal timing signal as an input signal. Thefrequency divider 208 may utilize a frequency division factor, fD, to generate a reference timing signal characterized by a reference frequency, fRef. The value of the reference frequency may be approximately equal to the ratio of the value of the reference frequency and the value of the frequency division factor, fRef/fD. - The filtering block 222 may enable generation of a filtered reference timing signal based on the reference timing signal. The filtered reference timing signal may also be characterized by the reference frequency. The filtering block 222 may generate the filtered reference timing signal may attenuating frequency components in the reference timing signal, which are characterized by harmonic frequencies of the reference frequency.
- For a filter block 224 comprising an opamp-RC filter circuit, the control signal, fControl, generated by the
control block 226 may cause the filter block 224 to adjust the R and/or C values for the R and/or C components. As a result of the adjustment, the corresponding value for the f−3dB (RC) filter cut-off frequency may be adjusted. The filter block 224 may generate a filter calibration signal by filtering the filtered reference timing signal. The filter block 224 may perform the filtering function by attenuating certain frequency components in the filtered reference timing signal based on the f−3dB (RC) filter cut-off frequency. The filtering function performed by the filter block 224 may modify an amplitude parameter and/or phase parameter, which characterizes at least a portion of the frequency components contained within the filtered reference timing signal. - For a filter block 224 comprising an GmC filter circuit, the control signal, fControl, generated by the
control block 226 may cause the filter block 224 to adjust the Gm and/or C values for the transconductor and/or C components. As a result of the adjustment, the corresponding value for the f−3dB (GmC) filter cut-off frequency may be adjusted. The filter block 224 may generate a filter calibration signal by filtering the filtered reference timing signal. The filter block 224 may perform the filtering function by attenuating frequency components in the filtered reference timing signal based on the f−3dB (GmC) filter cut-off frequency. The filtering function performed by the filter block 224 may modify an amplitude parameter and/or phase parameter, which characterizes at least a portion of the frequency components contained within the filtered reference timing signal. - The
control block 226 may compare values of the amplitude parameters and/or phase parameters, which characterize the corresponding frequency components in the filter calibration signal received from the filter block 224. The values of the amplitude parameters and/or phase parameters may be compared with expected values for the amplitude and/or phase parameters for the corresponding frequency components. The control block may generate a control signal, fControl, based on the comparison. - The component value mismatch problem may be avoided in comparison to the system illustrated in
FIG. 1 since the control signal, fControl, is generated based on the filter calibration signal generated by the filter block 224. This calibration method is direct as opposed to the indirect method inFIG. 1 . As with the system illustrated inFIG. 1 , a disadvantage in this method is that that the accuracy of the calibration may also be limited based on the range of values for frequency, fRef, and/or phase, φRef, which may be generated by thefrequency divider block 208. In addition, embodiments of the system illustrated inFIG. 2D may require additional circuitry to implement the filtering block 222. -
FIG. 3A is a block diagram of an exemplary analog baseband processing system supporting auto-calibration, in accordance with an embodiment of the invention. Referring toFIG. 3A , the baseband processing block 300 a may comprise a plurality of programmable gain amplifiers (PGAs) 304 a, 308 a, 310 a, and 314 a, andbaseband filters - For example, the baseband processing block 300 a may comprise an in-phase (I) component processing
path comprising PGAs baseband filter 306 a. The in-phase component processing path of the baseband processing block 300 a may process an input in-phase (I) signal 316 a to generate an output in-phase signal 318 a. The input in-phase signal 316 a may comprise a down converted component of a baseband frequency signal generated by an RF front end, for example. The baseband processing block 300 a may also comprise a quadrature component (Q) processingpath comprising PGAs baseband filter 312 a. The quadrature component processing path of the baseband processing block 300 a may process an input quadrature (Q) signal 320 a to generate an output quadrature signal 322 a. Theinput quadrature signal 320 a may comprise a down converted component of a baseband frequency signal generated by an RF front end, for example. - The
PGAs PGAs PGAs PGAs -
FIG. 3B is a schematic diagram of an exemplary opamp-RC baseband filter that may be used in accordance with an embodiment of the invention. Referring toFIG. 3B , thebaseband filter 300 b may comprise a sixth order Chebyschev filter, for example. TheChebyschev filter 300 b may comprise a plurality of operational amplifiers (opamps) a1, . . . , a6, a plurality of variable capacitors c1, . . . , c6, and a plurality of resistors r1, . . . , r14. In one embodiment of the invention, the opamp-RC integrators a1-c1-r1 and a6-c6-r8 may be arranged in a leapfrog formation. Each of the capacitors c1, . . . , c6 may be implemented as a binary weighted array of capacitors that may be controlled by 6 bits, for example. - In operation, the cut-off frequency f−3dB of the
Chebyschev filter 300 b may be changed during channel selection. For example, the cut-off frequency f−3dB of theChebyschev filter 300 b may be set to a value from 2 MHz, for example, thereby supporting channel bandwidth of about 5 MHz to about 8 MHz, which is specified by the DVB-T standard. Even though thebaseband filter 300 b comprises a sixth order Chebyschev filter, the present invention may not be so limited and an Nth order low-pass filter (LPF) may be utilized instead. - Even though the
baseband filter 300 b is described as a Chebyschev filter, the present invention may not be so limited. Other types of filters may also be utilized, such as Butterworth, Elliptic etc., for example. Furthermore, even though operational amplifier RC integrators are utilized within thefilter 300 b, the present invention may not be so limited and other integrator implementations may also be utilized, such as a Gm-C integrator. Additionally, topologies other than the leapfrog formation may be utilized, such as cascaded biquads. -
FIG. 3C is a block diagram of an exemplary baseband processing system using opamp-RC filters and an auto-calibration loop, in accordance with an embodiment of the invention. Referring toFIG. 3C , thebaseband processing block 300 c may comprise a plurality of programmable gain amplifiers (PGAs) 302 c, 304 c, 318 c, and 320 c, andbaseband filters switches digital frequency synthesizer 188, a fractional-N synthesizer 174, acrystal 178, anoscillator 180, anamplifier 324 c, root-means-square (rms) blocks 326 c and 328 c, acomparator 330 c, and controllogic block 334 c. Thedigital frequency synthesizer 188, Σ−Δ fractional-N synthesizer 174,crystal 178, andoscillator 180 may be substantially as described inFIG. 2C . - Even though rms blocks are used within the
baseband processing block 300 c, the present invention may not be so limited and peak detectors may be used instead of the rms blocks. - The functionality of the
PGAs PGAs FIG. 3A , respectively. Similarly, the functionality of the baseband filters 310 c and 312 c may be the same as the functionality of the baseband filters 306 a and 312 a inFIG. 3A , respectively. For example, the baseband filters 310 c and 312 c may each comprise a sixth order Chebyschev filters, such as theChebyschev filter 311 c or theChebyschev filter 300 b inFIG. 3B . - During an exemplary auto-calibration of the quadrature-phase signal path, the
switch 308 c may communicatively couple the output from thedigital frequency synthesizer 188 to the input for thebaseband filter 312 c. Theswitch 316 c may communicatively couple the output from thebaseband filter 312 c to the input for the rms block 326 c. Thedigital frequency synthesizer 188 may generate a reference frequency signal f−3 dB. The reference frequency signal may then be applied at the input of thebaseband filter 312 c and theamplifier 324 c. Theamplifier 324 c may attenuate the reference frequency signal by 3 dB, for example. The attenuated frequency signal may then be communicated to the rms block 328 c. After thebaseband filter 312 c filters the reference frequency signal communicated from thedigital frequency synthesizer 188, the filtered reference frequency signal may be communicated to the rms block 326 c. A corresponding exemplary auto-calibration may be performed for the in-phase signal path - Even though the
digital frequency synthesizer 188 generates a reference frequency signal f−3dB that corresponds to the frequency where the filter attenuates by 3 dB the present invention may not be so limited. In this regard,digital frequency synthesizer 188 may generate a reference frequency signal f−x dB that corresponds to the frequency of a main signal attenuated from the filter by x dB in accordance with the expected frequency response of the baseband filters 310 c and 312 c. In such instances, theamplifier 324 c may correspondingly attenuate the signal generated by thedigital frequency synthesizer 188. - While the
digital frequency synthesizer 188 may receive an input signal from the Σ−Δ fractional-N synthesizer 174 in the exemplary system ofFIG. 3C , various embodiments of the invention may not be so limited. In various embodiments of the invention, thedigital frequency synthesizer 188 may receive an LO signal from the phase locked loop (PLL) present in a communication receiver. In one aspect of the invention, a scheme for calibrating a filter in a communication receiver without requiring additional circuitry may be provided. In various embodiments of the invention, the Σ−Δ fractional-N synthesizer 174 may be utilized to enable the generation of a wide range of reference signal frequencies. In addition, the Σ−Δ fractional-N synthesizer 174 may enable more accurate generation of a specified frequency f−3dB than may be the case with many conventional filter calibration schemes. In another embodiment of the invention, adigital frequency synthesizer 188 may provide even greater accuracy in the generation of a specific frequency f−3dB. - The rms blocks 326 c and 328 c may perform an averaging function, for example, on the filtered reference frequency signal and the attenuated reference frequency signal, respectively. The averaged filtered reference frequency signal and the attenuated reference frequency signal may be compared by the
comparator 330 c. A comparator output signal may be communicated from thecomparator 330 c to thecontrol logic block 334 c. Thecontrol logic block 334 c may comprise suitable circuitry, logic, and/or code and may enable generation of a filter control signal 336 c and/orsynthesizer control signal 338 c. Thecontrol logic block 334 c may use aclock signal 332 c during control signal generation. In one exemplary embodiment of the invention, if a sixth order Chebyschev filter is used within thebaseband processing block 300 c, the filter control signal 336 c may comprise a 6-bit signal. In this regard, six bits may be used to program or adjust the capacitance of each variable capacitor c1, . . . , c6 in thefilter 311 c. - The filter control signal 336 c may be communicated to each of the baseband filters 310 c, 312 c. The baseband filters 310 c and 312 c may adjust capacitance of the variable capacitors within the filters and, thereby, change the cut-off frequency that determines filter bandwidth. The cut-off frequency and filter bandwidth of the
filters filter 312 c equals 3 dB, for example. Thesynthesizer control signal 338 c may be communicated to the digital frequency synthesizer from thecontrol logic block 334 c. Thesynthesizer control signal 338 c may be utilized by thedigital frequency synthesizer 188 to generate a reference frequency signal based on a received LO signal. - Even though an auto-calibration loop is described with respect to the quadrature signal path of the
baseband processing block 300 c, the same auto-calibration loop circuitry, such as thedigital frequency synthesizer 188,amplifier 324 c, rms blocks 326 c and 328 c,comparator 330 c and controllogic block 334 c, may be used with regard to the in-phase signal path of thebaseband processing block 300 c. - In one embodiment of the invention, for DVB-T applications, for example, an on-chip auto-calibration loop may be activated within the
baseband processing block 300 c upon power-up. The auto-calibration loop may adjust the cut-off frequency f−3dB of the filter response ofbaseband filters baseband processing block 300 c may support a plurality of channel bandwidths of 4-10 MHz, such as bandwidths specified by the DVB-T standard. A similar principle may apply to a plurality of channel bandwidths required by a specific communication standard. The auto-calibration loop may be utilized to enable adjustment of the filter cut-off frequency to a selected value as required, given that proper component value ranges have been provided in the design of the system. For example, the filter cut-off frequency may be adjusted to 210 kHz as required by the ISDB-T standard. -
FIG. 4 is a flow diagram illustrating exemplary steps in a filter calibration scheme using fractional-N frequency synthesized signals, in accordance with an embodiment of the invention. Referring toFIG. 4 , in step 402 a reference frequency at which to characterize abaseband filter 312 c may be determined. The reference frequency may correspond to an expected L−3dB cutoff frequency for thebaseband filter 312 c. Instep 404, a local oscillator (LO) signal andsynthesizer control signal 338 c may be generated. In various embodiments of the invention, the LO signal may be generated by a Σ−Δ fractional-N synthesizer 174, and thesynthesizer control signal 338 c may be generated by thecontrol logic block 334 c. - In
step 406, a reference signal may be generated. The reference signal may be characterized by the reference frequency, L−3dB. In various embodiments of the invention, the reference signal may be generated by thedigital frequency synthesizer 188. Thedigital frequency synthesizer 188 may generate the reference signal based on the LO signal and thesynthesizer control signal 338 c. - In
step 408, a filtered version of the reference frequency signal and an attenuated version of the reference frequency signal may be generated. In various embodiments of the invention, the filtered version of the reference frequency signal may be generated by applying the filtering characteristics of thebaseband filter 312 c to the reference frequency signal, while the attenuated version of the reference frequency signal may be generated based on the reference frequency signal by theamplifier 324 c. - In
step 410, the filtered version of the reference frequency signal and the attenuated version of the reference frequency signal may be compared by thecomparator 330 c. The comparison may be based on an amplitude and/or phase of the filtered version of the reference frequency signal versus the corresponding amplitude and/or phase of the attenuated version of the reference frequency signal. Prior to comparing, the respective signals may be averaged. - In step 412, a filter control signal 336 c may be generated and/or modified based on the comparison from
step 410. In various embodiments of the invention, the filter control signal may be generated and/or modified by thecontrol logic block 334 c. - In
step 414, a resistance, transconductance, and/or capacitance value may be modified for abaseband filter 312 c based on the filter control signal 336 c generated and/or modified in step 412. In various embodiments of the invention that comprise an RC filter, a resistance and/or capacitance value may be modified based on the filter control signal 336 c. In various embodiments of the invention that comprise a GmC filter, a transconductance and/or capacitance value may be modified based on the filter control signal 336 c. - Aspects of a system for filter calibration using fractional-N synthesized signals may include a single-chip
multi-band RF receiver 140 a that enables generation of a LO signal by a PLL circuit within the single-chip, and enables calibration of a frequency response for a filter circuit integrated within the chip. The frequency response may include a filter cut-off frequency and/or a filter phase shift. The filter cut-off frequency may represent a frequency beyond which the filter circuit may attenuate signal amplitudes by at least 3 dB, for example. The filter phase shift may represent a phase shift induced in signals that are processed by the filter circuit. A reference signal may be generated based on the generated LO signal and a synthesizer control signal. The frequency response may be calibrated by adjusting the filter circuit based on the generated reference signal. For an RC filter, exemplary parameters may comprise resistance values and/or capacitance values. For a GmC filter, exemplary parameters may comprise transconductance values and/or capacitance values. In various embodiments of the invention, the function of the PLL may be implemented by a Σ−Δ fractional-N synthesizer 174. The function of the filter circuit may be implemented by abaseband filter 312 c. - The single-chip
multi-band RF receiver 140 a may enable generation of a reference signal based on the generated LO signal and on a synthesizer control signal. The reference signal may be generated by thedigital frequency synthesizer 188. The synthesizer control signal may be generated by acontrol logic block 334 c. The single-chipmulti-band RF receiver 140 a may also enable generation of an attenuated reference signal by attenuating the reference signal. The reference signal may be attenuated by theamplifier 324 c. The single-chipmulti-band RF receiver 140 a may enable generation of a filtered reference signal based on filtering of the reference signal by the filter circuit. The single-chipmulti-band RF receiver 140 a may enable comparison of the attenuated reference signal and the filtered reference signal. The comparison may be performed by thecomparator 330 c. - The single-chip
multi-band RF receiver 140 a may also enable computation of averages for the attenuated reference signal and for the filtered reference signal, prior to performing the comparison. Each average may be computed by anrms block 326 c. The single-chipmulti-band RF receiver 140 a may enable generation of one or more filter control signals based on the comparison. Thecontrol logic block 334 c may enable generation of the filter control signals 336 c. - The single-chip
multi-band RF receiver 140 a may enable adjustment of a resistance value, and/or a capacitance value, for the filter circuit integrated within the single chip based on the generated one or more control signals. The single-chipmulti-band RF receiver 140 a may enable modification of a cut-off frequency for the filter circuit integrated within the chip based on the adjusting of the resistance value, and/or the capacitance value. - The single-chip
multi-band RF receiver 140 a may enable adjustment of a transconductance value, and/or a capacitance value, for the filter circuit integrated within the chip based on the generated one or more control signals. The single-chipmulti-band RF receiver 140 a may enable modification of a cut-off frequency for the filter circuit integrated within the chip based on the adjusting of the transconductance value, and/or the capacitance value. - The filter circuit integrated within the single-chip
multi-band RF receiver 140 a may comprise a low-pass filter. An exemplary low pass filter may comprise a Chebychev filter. - Accordingly, aspects of the invention may be realized in hardware, software, firmware and/or a combination thereof. The invention may be realized in a centralized fashion in at least one computer system or in a distributed fashion where different elements are spread across several interconnected computer systems. Any kind of computer system or other apparatus adapted for carrying out the methods described herein is suited. A typical combination of hardware, software and firmware may be a general-purpose computer system with a computer program that, when being loaded and executed, controls the computer system such that it carries out the methods described herein.
- One embodiment of the present invention may be implemented as a board level product, as a single chip, application specific integrated circuit (ASIC), or with varying levels integrated on a single chip with other portions of the system as separate components. The degree of integration of the system will primarily be determined by speed and cost considerations. Because of the sophisticated nature of modern processors, it is possible to utilize a commercially available processor, which may be implemented external to an ASIC implementation of the present system. Alternatively, if the processor is available as an ASIC core or logic block, then the commercially available processor may be implemented as part of an ASIC device with various functions implemented as firmware.
- While the invention has been described with reference to certain embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted without departing from the scope of the present invention. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the present invention without departing from its scope. Therefore, it is intended that the present invention not be limited to the particular embodiments disclosed, but that the present invention will include all embodiments falling within the scope of the appended claims.
Claims (25)
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US12/724,166 US20100173598A1 (en) | 2006-03-02 | 2010-03-15 | Method and system for filter calibration using fractional-n frequency synthesized signals |
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US77823206P | 2006-03-02 | 2006-03-02 | |
US11/431,960 US7680227B2 (en) | 2006-03-02 | 2006-05-11 | Method and system for filter calibration using fractional-N frequency synthesized signals |
US12/724,166 US20100173598A1 (en) | 2006-03-02 | 2010-03-15 | Method and system for filter calibration using fractional-n frequency synthesized signals |
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US11/431,960 Continuation US7680227B2 (en) | 2006-03-02 | 2006-05-11 | Method and system for filter calibration using fractional-N frequency synthesized signals |
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US12/724,166 Abandoned US20100173598A1 (en) | 2006-03-02 | 2010-03-15 | Method and system for filter calibration using fractional-n frequency synthesized signals |
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Also Published As
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US7680227B2 (en) | 2010-03-16 |
US20070207760A1 (en) | 2007-09-06 |
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