US20110212692A1 - Cascaded Filter Based Noise and Interference Canceller - Google Patents
Cascaded Filter Based Noise and Interference Canceller Download PDFInfo
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- US20110212692A1 US20110212692A1 US13/014,657 US201113014657A US2011212692A1 US 20110212692 A1 US20110212692 A1 US 20110212692A1 US 201113014657 A US201113014657 A US 201113014657A US 2011212692 A1 US2011212692 A1 US 2011212692A1
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B15/00—Suppression or limitation of noise or interference
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
- H04B1/40—Circuits
- H04B1/50—Circuits using different frequencies for the two directions of communication
- H04B1/52—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa
- H04B1/525—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa with means for reducing leakage of transmitter signal into the receiver
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
- H04B1/40—Circuits
- H04B1/44—Transmit/receive switching
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
- H04B1/40—Circuits
- H04B1/50—Circuits using different frequencies for the two directions of communication
- H04B1/52—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa
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Abstract
Signals propagating from an aggressor communication channel can cause detrimental interference in a victim communication channel. A high input power cascaded filter canceller (“HIPCF”) can obtain a sample of the signal that imposes interference and process the sampled signal to generate an interference compensation signal that, when applied to the victim communication channel, cancels or suppresses the detrimental interference. The HIPCF canceller can include two or more cascaded filters, such as band-pass filters, that block or reduce amplitude of a signal outside the communication frequency band of a victim receiver. The HIPCF canceller also includes an I/Q modulator that receives the filtered signal and generates the interference compensation signal by adjusting or updating one or more aspects of the filtered signal, such as a gain, phase, or delay, based upon feedback from the victim receiver or a power detector and algorithms executed by the controller.
Description
- This patent application claims to the benefit of U.S. Provisional Patent Application No. 61/308,697, entitled “High Power Cascaded Filter Based Noise Canceller” and filed Feb. 26, 2010. This application also claims to the benefit of U.S. Provisional Patent Application No. 61/375,491, entitled “Methods and Systems for Noise and Interference Cancellation” and filed Aug. 20, 2010. This application is related to U.S. patent application Ser. No. ______, [Attorney Docket No. 07982.105115], entitled “Methods and Systems for Noise and Interference Cancellation,” filed on the same date as this application. The entire contents of each of the foregoing priority and related applications are hereby fully incorporated herein by reference.
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FIG. 1 is a functional block diagram of a communication system, in accordance with certain exemplary embodiments. -
FIG. 2 is a block schematic diagram of a high input power cascaded filter (HIPCF) canceller, in accordance with certain exemplary embodiments. -
FIG. 3 is a block schematic diagram of certain components of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIG. 4 depicts a spectral diagram of signals received at a victim receiver antenna, in accordance with certain exemplary embodiments. -
FIG. 5 depicts a spectral diagram of signals received at the input of a victim receiver after cancellation of in-band unwanted spectral components by an HIPCF canceller, in accordance with certain exemplary embodiments. -
FIG. 6 is a block schematic diagram of a Q-enhanced band-pass filter (Q-Enhanced-BPF), in accordance with certain exemplary embodiments -
FIG. 7 is a block schematic diagram illustrating additional components of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIG. 8 depicts a functional block diagram of a communication system, in accordance with certain exemplary embodiments. -
FIG. 9 depicts a lookup table, in accordance with certain exemplary embodiments. -
FIG. 10 is a flow chart depicting a method for calibrating certain components of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIG. 11 is a flow chart depicting a method for configuring the filters of the HIPCF canceller ofFIG. 2 for a desired center frequency, in accordance with certain exemplary embodiments. -
FIG. 12 is a flow chart depicting a method for calibrating an input band-pass filter (Input-BPF) of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIG. 13 is a flow chart depicting a method for calibrating a low noise amplifier band-pass filter (LNA-BPF) of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIGS. 14A and 14B , collectivelyFIG. 14 , depict a flow chart of a method for calibrating a Q-Enhanced-BPF of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIG. 15 is a flow chart depicting a method for calibrating the Input-BPF of the HIPCF canceller ofFIG. 2 , in accordance with certain exemplary embodiments. -
FIG. 16 is a flow chart depicting a method for determining switch settings for a given frequency, in accordance with certain exemplary embodiments. -
FIG. 17 depicts implementation layers of noise and/or interference cancellation algorithms, in accordance with certain exemplary embodiments. -
FIG. 18 is a diagram depicting receiver sensitivity plotted versus coupled power amplifier phase noise, in accordance with certain exemplary embodiments. -
FIG. 19 is a diagram depicting an output signal to noise ratio (SNR) of a mobile TV tuner plotted versus a received mobile TV tuner signal strength, in accordance with certain exemplary embodiments. -
FIG. 20 is a flow chart depicting a fast binary algorithm for canceling noise or interference, in accordance with certain exemplary embodiments. -
FIG. 21 depicts a graph of in-phase (I) and quadrature (Q) values adjusted using binary algorithms, in accordance with certain exemplary embodiments. -
FIG. 22 is a flow chart depicting a minstep algorithm for canceling noise and/or interference, in accordance with certain exemplary embodiments. -
FIG. 23 depicts an I-Q plane with pseudorandom feedback values, in accordance with certain exemplary embodiments. -
FIG. 24 is a graph depicting a receive quality indicator plotted versus I or Q values resulting from an implementation of a dual slope algorithm (DSA), in accordance with certain exemplary embodiments. -
FIG. 25 is a flow chart depicting a DSA for canceling noise and/or interference, in accordance with certain exemplary embodiments. -
FIG. 26 is a graph depicting a receive quality indicator plotted versus I or Q values resulting from an implementation of the dual slope algorithm ofFIG. 24 , in accordance with certain exemplary embodiments. -
FIG. 27 is a flow chart depicting a track and search algorithm (TSA) for canceling noise and/or interference, in accordance with certain exemplary embodiments. -
FIG. 28 is a graph depicting cancellation points along an I-Q plane evaluated in an implementation of the TSA ofFIG. 27 , in accordance with certain exemplary embodiments. -
FIG. 29 is a flow chart depicting a method for finding a preferred noise cancellation point for two noise cancellers disposed in a communication system, in accordance with certain exemplary embodiments. -
FIG. 30 is a flow chart depicting a method for finding a preferred noise cancellation point for two noise cancellers disposed in a communication system, in accordance with certain exemplary embodiments. -
FIG. 31 is a flow chart depicting a method for finding a preferred noise cancellation point for two noise cancellers disposed in a communication system, in accordance with certain exemplary embodiments. - Many aspects of the invention can be better understood with reference to the above drawings. The drawings illustrate only exemplary embodiments of the invention and are therefore not to be considered limiting of its scope, as the invention may admit to other equally effective embodiments. The elements and features shown in the drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of exemplary embodiments of the present invention. Additionally, certain dimensions may be exaggerated to help visually convey such principles. In the drawings, reference numerals designate like or corresponding, but not necessarily identical, elements.
- The present invention is directed to systems and methods for compensating for signal interference occurring between two or more communication channels or between two or more communication elements in a communication system. Compensating for interference can improve signal quality, enhance communication bandwidth or information carrying capability, or improve receiver sensitivity. A communication channel may comprise a transmission line, a printed circuit board (PCB) trace, a flex circuit trace, an electrical conductor, a waveguide, a bus, a communication antenna, a medium that provides a signal path, or an active or passive circuit or circuit element such as a filter, oscillator, diode, VCO, PLL, amplifier, digital or mixed signal integrated circuit. Thus, a channel can comprise a global system for mobile communications (GSM) device, a processor, a detector, a source, a diode, an inductor, an integrated circuit, a connector, a circuit trace, or a digital signal processing (DSP) chip, to name only a few possibilities.
- Exemplary embodiments described herein can include a high input power cascaded filter (HIPCF) noise and interference canceling device. Exemplary HIPCF cancellers described herein can support selectively canceling, correcting, addressing, or compensating for interference, electromagnetic interference (EMI), noise (e.g., phase noise, intermodulation products, and other interfering noise), spurs, or other unwanted spectral components associated with one or more communication paths of a communication system, such as a high speed digital communication system in a portable electronic device. For the purpose of this specification, the term “high power” generally refers to signals having a power ratio up to approximately +33 dBm (decibels relative to one milliwatt) or more. For example, exemplary HIPCF cancellers described herein can be coupled to the output of cellular telephone power amplifiers having output power of this magnitude.
- The HIPCF cancellers can obtain a sample of a communication signal that imposes interference from a communication path of an aggressor transmitting device and process the sampled signal to produce an interference compensation signal. The HIPCF can deliver the interference compensation signal into or onto a communication path of a victim receiver that is a recipient of the interference, to cancel, mitigate, suppress, or otherwise compensate for the received interference.
- Turning now to the drawings, in which like numerals indicate like or corresponding (but not necessarily identical) elements throughout the figures, exemplary embodiments of the invention are described in detail.
FIG. 1 is a functional block diagram of acommunication system 100, in accordance with certain exemplary embodiments. Referring toFIG. 1 , thecommunication system 100 includes atransmitter 105 that transmits electromagnetic signals via a transmittingantenna 115. Atransmit path 107, including one or more electrical conductors, couples thetransmitter 105 to the transmittingantenna 115. In certain exemplary embodiments, thetransmitter 105 conveys data to a remote device using one or more communications standards or methods, such as the Global System for Mobile Communications (GSM), Code Division Multiple Access (CDMA), Long Term Evolution (LTE), Wideband Code Division Multiple Access (W-CDMA), Digital Cellular System (DCS), Personal Communication Service (PCS), and Wireless Local Area Network (WLAN). One of ordinary skill in the art having the benefit of the present disclosure would appreciate that thecommunication system 100 described herein is not limited to the aforementioned communication standards and methods, but instead can be used with many other types of signal transmitting technologies. - Disposed along the
transmit path 107 between thetransmitter 105 and the transmittingantenna 115 is apower amplifier 110. Thepower amplifier 110 adjusts the power level of the transmitter's output signals prior to the signals being propagated by theantenna 115. When apower amplifier 110 adjusts the power level of a signal, unwanted spectral components can be introduced onto the signal. For example, thetransmitter 105 can transmit signals having a certain carrier frequency or a certain fundamental tone. Thepower amplifier 110 can introduce intermodulation products having a different frequency than that carrier frequency or fundamental tone. Other components associated with thetransmitter 105 also can cause noise or other unwanted spectral components to be introduced onto the signal. For example, thetransmitter 105 may include a local oscillator and/or one or more up-conversion mixer(s) that can cause unwanted spectral components, including out-of-band noise (outside the frequency band of the transmitted signal) sometimes referred to as out-of-band blockers, to be introduced onto the signal. - The
communication system 100 also includes areceiver 135 that receives signals via a receivingantenna 120 and a receivepath 133 that electrically couples the receivingantenna 120 to thereceiver 135. In certain exemplary embodiments, thereceiver 135 receives signals within the same or a different frequency band than that of thetransmitter 105. For example, a mobile electronic device, such as a mobile telephone, personal digital assistant (PDA) or mobile computer, may include atransmitter 105 that communicates via one of the communication protocols discussed above and areceiver 135 that communicates in a different frequency band, such as a mobile TV tuner, a Bluetooth receiver, a Worldwide Interoperability for Microwave Access (WiMAX) receiver, or a Global Positioning System (GPS) receiver. In the illustrated embodiment, the receivepath 133 includes an optional receive (RX)filter 140. The optional receivefilter 140 can include a band-pass filter or other filter arrangement that allows communication signals received by theantenna 120 within the frequency band of thereceiver 135 to pass to thereceiver 135, while blocking signals outside the frequency band of thereceiver 135. - The frequency band of the
receiver 135 may be near the frequency band of thetransmitter 105 such that phase noise or other unwanted spectral components produced by thepower amplifier 110 or another component disposed along the transmitpath 107 degrades the sensitivity of thereceiver 135. For example, thecommunication system 100 may be embodied in a mobile device having a CDMA, GSM orLTE transmitter 105 and a mobile TV tuner asreceiver 135. Certain types of CDMA andGSM transmitters 105 transmit signals within a frequency band of approximately 800 MHz to 900 MHz and certain types ofLTE transmitters 105 operate within a frequency band of 698 MHz to 798 MHz. These transmitted signals often include phase noise having a frequency between 450 MHz and 776 MHz, which falls within the receive band of some mobile TV tuners and many other communication devices. If this in-band phase noise is imposed onto the signal path of the receiver 135 (e.g., air coupled from the transmittingantenna 115 to the receiving antenna 120), the phase noise can degrade the sensitivity of thereceiver 135. Generally, receive filters, such as receivefilter 140, do not filter out in-band noise as the noise is within the frequency band of thereceiver 135 and thus, the pass-band of receivefilter 140. Therefore, the phase noise may pass through the receivefilter 140 and degrade the sensitivity of thereceiver 135. - To prevent a degradation of the sensitivity of the
receiver 135 caused by in-band noise (noise having a frequency within the frequency band of the receiver 135) or nearby out-of-band noise caused by the transmissions from the transmittingantenna 115, thecommunication system 100 includes aHIPCF canceller 130. An input of theHIPCF canceller 130 is coupled to the transmitpath 107 at the output of thepower amplifier 110 by way of asampling device 125. Thesampling device 125 can include a capacitor, (e.g., a sampling or tapping capacitor), a resistor, a coupler, a coil, a transformer, a signal trace, or an antenna/detector. Sampling devices having two ports or two terminals, such as a resistor, capacitor, coil, transformer, or signal trace, can have a first port electrically coupled to the transmitpath 107 and a second port electrically coupled to the input of theHIPCF canceller 130. In an antenna/detector, having mostly one terminal, the second terminal is formed by the electromagnetic field protruding from the device, allowing for locating the device close to the transmitpath 107. - In the illustrated embodiment, the
sampling device 125 is connected to the transmitpath 107 at the output of thepower amplifier 110. Thesampling device 125 obtains samples of the signal (“sampled transmit signals”) at the output of thepower amplifier 110 and provides the sampled transmit signals to theHIPCF canceller 130. In certain exemplary embodiments, thesampling device 125 may produce attenuation on the sampled transmit signals. For example, the amplitude of the sampled transmit signal may be 20 dBc (decibels relative to carrier) lower than the signal at the output of thepower amplifier 110. - In certain exemplary embodiments, the
sampling device 125 includes a voltage-controlled capacitor (varactor) for trimming frequency dependent attenuation to a desired value and hence, compensate for gain ripple. In one example, thesampling device 125 includes a voltage controlled varactor. The capacitance of the varactor can be adjusted via a control voltage. This control voltage can be generated by a controller 235 (FIG. 2 ) of theHIPCF Canceller 130 and transmitted to thesampling device 125 via one or moreelectrical conductors 127. - The
HIPCF canceller 130 selectively suppresses or cancels interfering signals (e.g., phase noise, intermodulation products, unwanted spectral components, etc.) produced by the power amplifier 110 (or another component along the transmit path 107) having a frequency within or near the receive frequency band of thereceiver 135 that would otherwise interfere with the sensitivity of thereceiver 135. TheHIPCF canceller 130 obtains samples of the signals output by thepower amplifier 110 and processes the sampled transmit signals to produce an interference compensation signal that, when applied to an input of thereceiver 135, suppresses or cancels the interfering signals. In certain exemplary embodiments, theHIPCF canceller 130 tunes the interference compensation signal using feedback, such as a receive signal quality indicator, obtained from thereceiver 135 via afeedback path 180 including one or more electrical conductors. Theexemplary HIPCF canceller 130 is described in further detail below in connection withFIGS. 2-31 . - The interference compensation signal is applied to the receive
path 133 of thereceiver 135 at acancellation point 134. In certain exemplary embodiments, thecancellation point 134 is implemented by converging an electrical conductor of the receivepath 133 with an electrical conductor along the output path of theHIPCF canceller 130 such that the electrical conductors make electrical contact. For example, a flex circuit trace of the receivepath 134 may be connected to a flex circuit trace of the HIPCF output. In certain exemplary embodiments, a component, such as a coupler, a summation node, an adder, or another suitable technology may be used to apply the interference compensation signal to thereceiver path 133 of thereceiver 135. - The
communication system 100 illustrated inFIG. 1 can transmit electromagnetic signals having a frequency within a first frequency range and receive electromagnetic signals having a frequency within a second frequency range. The first frequency range may be close to the second frequency range or even include frequencies that overlap or are included in the second frequency range. In operation, thetransmitter 105 transmits signals along the transmitpath 107 to thepower amplifier 110. Thepower amplifier 110 adjusts the intensity of the signals received from thetransmitter 105 and outputs the intensity adjusted signal to the transmittingantenna 115. The transmittingantenna 115 transmits the signals received from thepower amplifier 110. A portion of the signals transmitted by the transmittingantenna 115 is coupled to the receivingantenna 120 via air. If received by thereceiver 135, signals coupled onto the receivingantenna 120 originating from the transmittingantenna 105 may interfere with or degrade the sensitivity of thereceiver 135. For example, signals transmitted by the transmittingantenna 115 having a frequency within the frequency band or close to the frequency band of the receiver 135 (e.g., intermodulation spectra appearing like phase noise tails generated by the power amplifier 110) can degrade the sensitivity of thereceiver 135. To compensate for this interference or sensitivity degradation, theHIPCF canceller 130 obtains samples of the signals output by the power amplifier 110 (via the sampling device 125) and processes the sampled transmit signals to produce an interference compensation signal that, when applied to an input of thereceiver 135, compensates for the interference imposed on thereceiver 135 by the signals transmitted by transmittingantenna 115. -
FIG. 2 is a block schematic diagram of theHIPCF canceller 130 ofFIG. 1 , in accordance with certain exemplary embodiments. Theexemplary HIPCF canceller 130 includes a band-pass filter (Input-BPF) 205 that receives signal samples from thesampling device 125. In this exemplary embodiment, the Input-BPF 205 includes an inductor L1 and two switchable capacitors C1 and C2. The resonant frequency of the Input-BPF 205 is tunable by adjusting capacitance of one or both of the switchable capacitors C1 and C2. The switchable capacitors C1 and C2 are described in further detail below in connection withFIG. 3 . - In certain exemplary embodiments, the inductor L1 is a high-Q inductor. The use of a high-Q inductor can provide performance advantages, such as providing additional attenuation to signals outside of the pass band of the Input-
BPF 205 and hence to protect subsequent components in theHIPCF canceller 130, allowing to trade linearity for a lower noise floor. In certain exemplary embodiments, the inductor L1 is a low-Q inductor. In certain exemplary embodiments, the Input-BPF 205 includes a Q-enhancement circuit 290 to improve the quality factor (Q-factor) of the inductor L1. However, some Q-enhancement circuits may introduce noise or interference onto signals passed through the Input-BPF 205. - The resonant frequency of the Input-
BPF 205 can be tuned to (or near) the receive frequency of thereceiver 135 in order to pass interfering signals at that frequency that may be present on the sampled transmit signals and block or filter out aggressor signals, such as fundamental tones or carrier signals transmitted by thetransmitter 105 as well as other out-of-band blocker signals (signals having a frequency outside of the receiver's frequency band). If thereceiver 135 includes a mobile TV tuner or other frequency adjustable device, the resonant frequency of the Input-BPF 205 may be adjusted to match the frequency of a current channel to which the mobile TV tuner is set. For example,channel 50 of a mobile TV tuner may have a receive frequency within the frequency band of 686 MHz to 692 MHz. While the mobile TV is tuned to this frequency, the Input-BPF 205 also can be tuned to this frequency automatically. If the mobile TV is subsequently tuned to another channel having a different receive frequency, the resonant frequency of the Input-BPF 205 can be adjusted to match the receive frequency of the new channel. For example, thecontroller 235 may communicate with thereceiver 135 to obtain the current receive frequency for thereceiver 135. In response, thecontroller 235 may adjust the switchable capacitors C1 and C2 such that the resonant frequency of the Input-BPF 205 is close to or equal to the receive frequency. - The Input-
BPF 205 reduces the amplitude of signals having frequencies differing from the resonant frequency of the Input-BPF 205. For example, if thereceiver 135 and thetransmitter 105 are operating at different frequencies, the Input-BPF 205 can reduce the amplitude of the fundamental tones of the sampled transmit signal. In certain exemplary embodiments, the Input-BPF 205 may reduce the amplitude of the fundamental tones of the sampled transmit signal located at 824 MHz by approximately 13-18 dBc while its center frequency is tuned to 749 MHz (corresponding to channel 60 of a mobile TV tuner). The output of the Input-BPF 205 is electrically coupled to a low noise amplifier (LNA) 210. TheLNA 210 amplifies the signal output by the Input-BPF 205 and passes this amplified signal to a second band-pass filter, referred to herein as LNA-BPF 215. In certain exemplary embodiments, theLNA 210 is a cascode LNA. - In this exemplary embodiment, the LNA-
BPF 215 includes an inductor L2 and a switchable capacitor C3. In certain exemplary embodiments, the inductor L2 is a high-Q inductor. In certain exemplary embodiments, the inductor L2 is a low-Q inductor. In certain exemplary embodiments, the LNA-BPF 215 includes a Q-enhancement circuit 291 to improve the Q-factor of the inductor L2. In certain exemplary embodiments, the Q-factor of L2 is less than the Q-factor of L1. In certain exemplary embodiments, the Q-factor of L2 is greater than the Q-factor of L2. - Similar to the Input-
BPF 205, the resonant frequency of the LNA-BPF 215 can be set to the receive frequency of thereceiver 135 to pass signals at that frequency and to further filter the fundamental tones and out-of-band blockers from the sampled transmit signal. In certain exemplary embodiments, the LNA-BPF 215 may further reduce the amplitude of the fundamental tones located at 824 MHz by approximately 13-18 dBc while its center frequency is tuned to 749 MHz. - The output of the LNA-
BPF 215 is electrically coupled to a variable gain amplifier (VGA) 220 that adjusts the amplitude of signals output by the LNA-BPF 215. In certain exemplary embodiments, theVGA 220 includes multiple variable gain amplifiers for adjusting the amplitude of the signal received from the LNA-BPF 215. The amplitude adjusted signal output by theVGA 220 is then passed to a third band-pass filter (Q-Enhanced-BPF) 225. - The Q-Enhanced-
BPF 225 can include an inductor L3 and a switchable capacitor 615 (FIG. 6 ) for tuning the Q-Enhanced-BPF 225 to the receive frequency of thereceiver 135 to pass any signals at that frequency and to further filter the fundamental tones and out-of-band blockers of the sampled transmit signal. In certain exemplary embodiments, the inductor L3 can be a high-Q inductor (e.g., off-chip), or a low-Q on-chip spiral inductor. In certain exemplary embodiments, the Q-Enhanced-BPF 225 also includes a Q-enhancement circuit 292. In certain exemplary embodiments, the Q-Enhanced-BPF 225 includes current switching (FIG. 6 ) to adjust its Q-factor. In certain exemplary embodiments, the Q-Enhanced-BPF 225 can further reduce the amplitude of the fundamental tones remaining in the signal received from theVGA 220 located at 824 MHz by up to 26 dBc or more while its center frequency is tuned to 749 MHz. The output of the Q-Enhanced BPF 225 is electrically coupled to an I/Q modulator 230. - Although in the illustrated embodiment, a cascade of band-
pass filters receiver 135, other types of filters may be utilized in addition to or in place of one or more of the band-pass filters pass filters receiver 135 which would normally not interfere with the receiver's sensitivity. The signals within the receive frequency band of thereceiver 135 are passed through the band-pass filters Q modulator 230. These in-band signals are also amplified by theLNA 210 and theVGA 220. - The I/
Q modulator 230 adjusts at least one of the phase, amplitude, and delay of the signal received from the Q-Enhanced-BPF 225 to produce an interference compensation signal that, when applied to the receivepath 133 of thereceiver 135, reduces, suppresses, cancels, or otherwise compensates for the noise and/or interference present on the receivedpath 133 of thereceiver 135 imposed by signals transmitted by the transmittingantenna 115. In certain exemplary embodiments, this interference compensation signal has a 180 degree phase shift relative to that of the in-band noisy signal and an amplitude close to or the same as that of the in-band noisy signal. Thus, the interference compensation signal reduces or cancels the in-band noisy signal. - In certain exemplary embodiments, the aforementioned parameters of amplitude, phase, and delay are tuned based on a set of instructions (e.g., algorithms) stored in a memory device 760 (
FIG. 7 ) and executed by thecontroller 235 using feedback from the victim receiver's receive signal quality indicator, such as Bit-Error-Rate (BER), Packet-Error-Rate (PER), Receive Signal Strength Indicator (RSSI), noise floor, Signal-Noise-Ratio (SNR), Error Vector Magnitude (EVM), and Position Accuracy (for GPS) etc. Exemplary algorithms for determining settings for adjusting the amplitude, phase, and delay are described below with reference toFIGS. 17-31 . - As shown in
FIG. 7 , in certain exemplary embodiments, theHIPCF canceller 130 includes apower detector 745, such as a peak detector, coupled to the input of the I/Q modulator 230. Thepower detector 745 senses or measures the power level of the signal at the input of the I/Q modulator 230 and provides an indication of the power level to thecontroller 235. Thecontroller 235 uses this power level value to trim the currents and hence Qmax of the Q-Enhanced-BPF 225 for maintaining an acceptable suppression of the noise and/or interference imposed on thereceiver 135 by signals transmitted by the transmittingantenna 115. In certain exemplary embodiments, theHIPCF canceller 130 includes an analog-to-digital (A/D)converter 750 that receives the power level value from thepower detector 745 and provides a digital representation of the power level value to thecontroller 235. Thecontroller 235 executes a calibration routine to ensure an acceptable level of suppression of the noise and/or interference imposed on thereceiver 135 by signals transmitted by the transmittingantenna 115. Exemplary calibration routines are described below with reference toFIGS. 9-16 . - The
controller 235 can be implemented in the form of a microcontroller, microprocessor, computer, state machine, programmable device, control logic, analog and digital circuitry, or other appropriate technology. Thecontroller 235 can execute one or more processes or programs for adjusting the settings of each of the band-pass filters FIG. 6 ). In one example, thecontroller 235 automatically adjusts the resonant frequencies of one or more of the band-pass filters receiver 135. For example, if thereceiver 135 comprises a mobile TV tuner, thecontroller 235 adjusts the resonant frequency of the band-pass filters controller 235 adjusts the resonant frequencies of the band-pass filters SCA 615, respectively, as discussed below with reference toFIG. 3 . - The
controller 235 also can adjust or refine the settings of the I/Q modulator 230, the band-pass filters VGA 220 to account for environmental changes, such as changes to temperature, supply voltage, and antenna coupling. In certain exemplary embodiments, thecontroller 235 executes a calibration routine (FIG. 16 ) to identify acceptable settings based on these environmental changes and stores the identified optimal settings for subsequent use. The algorithm(s) can be embodied as software stored on thecontroller 235 or on amemory storage device 760. Alternatively, the algorithm(s) can be implemented in one or more hardware devices, such as discrete logic gates. - The
HIPCF canceller 130 also includesauxiliary circuits 240. As shown inFIG. 7 , theauxiliary circuits 240 include atemperature sensor 755, apower detector 745, one or more analog todigital converters 750, digital to analog converters, and other types of circuits for use by theHIPCF canceller 130. Theauxiliary circuits 240 can also include one or morememory storage devices 760, such as RAM, ROM, and/or flash memory. Settings for each band-pass filter memory storage device 760. Additionally, settings for the I/Q modulator 230 may be stored on thememory storage device 760. For example, settings for each channel of a mobile TV tuner may be stored on thememory storage device 760. - Certain elements or functions of the
HIPCF canceller 130 can be embodied in an integrated circuit, for example as depicted by thechip boundary 250 thatFIG. 2 illustrates. For example, the switchable capacitors C1-C3, theLNA 210, theVGA 220, the Q-Enhanced-BPF 225, the I/Q modulator 230, thecontroller 235 and one or more of theauxiliary circuits 240 can be embodied in a single integrated circuit or multiple integrated circuits. Although the inductors L1 and L2 are illustrated as off-chip inductors in the illustrated exemplary embodiment, other exemplary embodiments may employ on-chip inductors in the band-pass filters - Referring to
FIGS. 1 and 2 , theHIPCF canceller 130 suppresses, cancels, or otherwise compensates for in-band or nearby out-of-band (relative to the receive frequency of the receiver 135) interfering signals imposed on thereceiver 135 by signals transmitted by thetransmitter 105 via the transmittingantenna 115. That is, theHIPCF canceller 130 compensates for interfering signals transmitted by the transmittingantenna 115 that has a frequency within or near the frequency band of thereceiver 135. TheHIPCF canceller 130 obtains samples of signals transmitted by thetransmitter 105 from thesampling device 125 and process the samples to produce an interference compensation signal that, when applied to an input of thereceiver 135, compensates for the imposed interfering signals. - The
exemplary HIPCF 130 includes three band-pass filters sampling device 125 that are out-of-band with respect to the receive frequency of thereceiver 135. The components of the sampled transmit signals in-band with respect to thereceiver 135 are used to generate the interference compensation signal. At least one of phase, amplitude, and delay of these components of the sampled transmit signal are adjusted by the I/Q modulator 230 to generate the interference compensation signal. Thecontroller 235 can execute one or more calibration algorithms and/or one or more tuning algorithms to improve the level of interference compensation. Thecontroller 235 can obtain feedback from thepower detector 745 or from thereceiver 135 and use this feedback during execution of the algorithms. These algorithms are discussed in detail below with reference toFIGS. 9-31 . -
FIG. 3 is a block schematic diagram 300 of certain components of theHIPCF canceller 130 ofFIG. 2 , in accordance with certain exemplary embodiments. In particular,FIG. 3 is a transistor level diagram of an exemplary Input-BPF 205, an exemplary LNA-BPF 215, and anexemplary LNA 210. Referring toFIG. 3 , the Input-BPF 205 includes a first switched capacitor array (SCA) 305 and asecond SCA 310. Each of theSCAs SCAs BPF 205 can be adjusted by selecting one or more of the capacitors from theSCAs SCAs BPF 205, or have a different weighted value to cover the frequency band of thereceiver 135. In certain exemplary embodiments, thecontroller 235 can activate and deactivate the switches in theSCAs BPF 205. - The
SCAs receiver 135. In certain exemplary embodiments, theSCAs BPF 205 for high UHF (ultra high frequency) channels (such aschannel 50 for a mobile TV tuner) that have frequencies close to that of a GSM, CDMA, orLTE transmitter 105, switch M61 can be activated and switches M60 and M62 can be deactivated. This provides a voltage divider between a capacitor in thefirst SCA 305 and a capacitor in thesecond SCA 310. - To configure the Input-
BPF 205 for low UHF channels, such as channel 26 of a mobile TV tuner which may have a frequency between 542 MHz and 548 MHz, thesecond SCA 310 can be disconnected from the Input-BPF 205 circuit by activating switches M60 and M62 and deactivating switch M61. In certain exemplary embodiments, this configuration reduces the attenuation of the sampled signal by 15 dB. This compensates for frequency dependent gain variations of the three band-pass filters - In certain exemplary embodiments, the inductor L1 of the Input-
BPF 205 can be biased at half of Vdd for an integrated circuit that the inductor L1 is coupled to in order to maximize the input voltage swing without violating the integrated circuit's specification while capacitor C4 provides a return path to ground. In certain exemplary embodiments, the bias voltage for the inductor L1 may be higher if adequate precautions are taken regarding the maximum breakdown voltage of the circuit, for example by employing zener diodes for ESD, cascoded input stages, larger channel devices, LDD MOSFETs, etc. - The LNA-
BPF 215 also includes an SCA 315 having ‘n+1’ number of capacitors. In this exemplary embodiment, each capacitor in the SCA 315 includes a corresponding transistor switch (e.g., a MOS transistor) for activating the capacitor. Similar to the Input-BPF 205, the resonant frequency of the LNA-BPF 215 can be adjusted by selecting one or more of the capacitors of the SCA 315. - In this exemplary embodiment, the
LNA 210 is a cascode LNA having two transistors M4 and M5. Thecascode LNA 210 can use a frequency dependent degeneration that can be activated at high frequencies by deactivating switch M7. This serves the purpose of increasing input linearity at high frequencies as well as providing sufficient gain at low frequencies for maintaining low noise figure of theLNA 210 by activating switch M7. - The capacitors and the switches in each of the
SCAs FIG. 3 , this can be accomplished by inserting MOS switches M10 to M1 n between capacitor C10 to C1 n and the chip input, MOS switches M20 to M2 n between capacitor C20 to C2 n and the AC coupling capacitor C4, and MOS switches M30 to M3 n between capacitor C30 to C3 n and the output ofLNA 210. High-Q external inductors L1 and L2 may be used in theHIPCF canceller 130 instead of on-chip inductors to provide higher frequency selectivity. The use ofSCAs HIPCF canceller 130 can have an input pin, an AC ground pin, and an LNA pull-up pin, each having multiple ESD diodes arranged in series to allow for a larger signal swing. - In certain exemplary embodiments, one or more of the band-
pass filters pass filters pass filters pass filters transmitter 105 has a frequency greater than the frequency range for interference suppression. In certain exemplary embodiments, one or more of the band-pass filters pass filters transmitter 105 has a frequency less than the frequency range for interference suppression. In certain exemplary embodiments, a combination of low-pass, high-pass, and band-pass filters may be used in place of the band-pass filters -
FIG. 4 depicts a spectral diagram 400 of signals received at a victim receiver antenna, such asantenna 120 ofFIG. 1 , in accordance with certain exemplary embodiments. Referring toFIGS. 1 and 4 , the spectral diagram 400 shows theamplitude 403 of the signals received at theantenna 120 plotted againstsignal frequency 402. The spectral diagram 400 includes afirst peak 404 corresponding to the carrier frequency FT of theaggressor transmitter 105 and asecond peak 405 corresponding to the channel frequency FR of thevictim receiver 135. The spectral diagram 400 also includes anoise sideband 406 corresponding to the phase noise or other unwanted spectral components generated by theaggressor transmitter 105. In certain exemplary embodiments, the victim receiver's preferred signal-to-noise ratio (SNR) for proper reception is not met by the amplitude difference between thesecond peak 405 and thenoise sideband 406. -
FIG. 5 depicts a spectral diagram 500 of signals received at the input of a victim receiver, such asreceiver 135 ofFIG. 1 , after cancellation of in-band unwanted spectral components by an HIPCF canceller, such as theHIPCF canceller 130 ofFIG. 1 , in accordance with certain exemplary embodiments. Referring toFIGS. 1 and 5 , the spectral diagram, 500 shows theamplitude 403 of the signals received at thereceiver 135 plotted againstsignal frequency 402. The spectral diagram 500 includes anoise sideband 506 corresponding to the phase noise or other unwanted spectral components generated by theaggressor transmitter 105. Thisnoise sideband 506 differs from thenoise sideband 406 of spectral diagram 400 in that thenoise sideband 506 includes anotch 507 centered at the channel frequency FR of thevictim receiver 135. Thisnotch 507 results from the compensation provided by the interference compensation signal generated by theHIPCF canceller 130 and applied to the input of thereceiver 135. In certain exemplary embodiments, the SNR of the signal is improved by an amount corresponding to the depth of thenotch 507. Thus, thenotch 507 improves the signal SNR, thus increasing the sensitivity of thevictim receiver 135. For example, improved cancellation of phase noise or other unwanted spectral components by theHIPCF canceller 130 results in adeeper notch 507 and thus, better SNR for thevictim receiver 135. -
FIG. 6 is a block schematic diagram of the Q-enhancedBPF 225 ofFIG. 2 , in accordance with certain exemplary embodiments. In particular,FIG. 6 is a transistor level diagram of the Q-enhancedBPF 225. The exemplary Q-enhancedBPF 225 includes anLC tank 610 having an inductor L3, abypass switch 670, and anSCA 615. In certain exemplary embodiments, the inductor L3 is a low-Q on-chip spiral inductor. In certain exemplary embodiments, the inductor L3 is a high-Q off-chip inductor. Similar to the band-pass filters BPF 225 can be set (e.g., automatically by the controller 235) to the receive frequency of thereceiver 135 to pass in-band signal components and to further filter, block, or reduce the intensity of the fundamental tones and out-of-band blockers from the sampled transmit signal. - The
SCA 615 includes a number ‘n+1’ of capacitors C40-C4 n. In the illustrated embodiment, each capacitor C40-C4 n includes two corresponding transistor switches (e.g., a MOS transistor) for activating the capacitor. For example, the capacitor C40 includes transistor switches M40 and M50. In addition, the Q-Enhanced-BPF 225 also includes two series-connected voltage controlled capacitors VC1 and VC2 in parallel with theSCA 615. In certain exemplary embodiments, the voltage controlled capacitors VC1 and VC2 are varactors. Disposed between the two voltage controlled capacitors VC1 and VC2 is acenter tap 655 that electrically couples the voltage controlled capacitors VC1 and VC2 to a digital-to-analog (D/A)converter 650. The D/A converter 650 varies the voltage level of the voltage control capacitors VC1 and VC2 in response to a signal received from thecontroller 235. Thecontroller 235 can adjust the resonant frequency of the Q-Enhanced-BPF 225 by activating one or more of the capacitors C40-C4 n (via switches M40-M4 n and M50-M5 n) and by controlling the voltage level at thecenter tap 655 and thus, the capacitance of the voltage controlled capacitors VC1 and VC2. The voltage controlled capacitors VC1 and VC1 enable thecontroller 235 to finely tune the resonant frequency of the Q-Enhanced-BPF 225. - The exemplary Q-Enhanced-
BPF 225 also includes across-coupled pair 620 of transistor switches M8 and M9 in parallel with theSCA 615. Thecross-coupled pair 620 provides a negative resistance to reduce the resistance of an LC tank formed by inductor L3, theSCA 615, and voltage controlled capacitors VC1 and VC2. - The Q-Enhanced-
BPF 225 includes a number ‘n+1’ of current sources M60-M6 n (e.g., binary weighted), each having a gate terminal electrically coupled together and with a reference current (Ref_C). The Q-Enhanced-BPF 225 also includes a number ‘n+1’ of current switches M70-M7 n. By selecting one or more of the current sources M60-M6 n via activating and deactivating (e.g., by the controller 235) the corresponding current switch(es) M70-M7 n, the current in switches M8 and M9 can be adjusted which in turn adjusts the resistance of theLC tank 610. Thus, the Q-factor of the Q-Enhanced-BPF 225 can be adjusted. For example, the Q-factor of the Q-Enhanced-BPF 225 can be adjusted to a desired level such that filtering of out-of-band signals is improved or maximized without the Q-Enhanced-BPF 225 oscillating. - The Q-Enhanced-
BPF 225 also includes abypass switch 670 having a resistor R8 electrically coupled to and disposed between two transistor switches M80 and M81. As discussed in further detail with reference toFIGS. 11-15 , the switches M80 and M81 can be activated or turned on during calibration of the Input-BPF 205 and the LNA-BPF 215 while current sources M60-M6 n are deactivated. When the switches M80 and M81 are activated, the resistor R8 may detune the LC tank. During normal operation, the switches M80 and M81 are typically deactivated. -
FIG. 7 is another block schematic diagram of theHIPCF canceller 130 depicting additional components of theHIPCF 130, in accordance with certain exemplary embodiments. As shown inFIG. 7 , theexemplary HIPCF 130 also includes bypass switches 720 and 725 for use during calibration of theHIPCF 130. In particular, the Input-BPF 205 includes thebypass switch 720 and theLNA 215 includes thebypass switch 725. Thebypass switch 720 includes a transistor switch M82 and a resistor R2. Similarly, thebypass switch 725 includes a transistor switch M83 and a resistor R3. During the configuration of the HIPCF 130 (e.g., using automatic test equipment (ATE), bench measurement, or in-site calibration), each of the bypass switches 720, 725 and thebypass switch 670 of the Q-Enhanced-BPF 225 can be activated and deactivated to selectively tune the band-pass filters - The
HIPCF canceller 130 also includes abuffer 770 disposed between theVGA 220, Q-enhanced-BPF 225, and the I/Q Modulator 230. Theauxiliary circuits 240 include apower detector 745 electrically coupled to the output of thebuffer 770. Thepower detector 745 measures the power level of the sampled transmit signal at the output of thebuffer 770 and provides an indication of the measurement to an A/D converter 750. The A/D converter 750 converts the indication to a digital signal and provides the digital signal to thecontroller 235. - The
auxiliary circuits 240 also include atemperature sensor 755 having an output electrically coupled to thecontroller 235. Thetemperature sensor 755 is positioned on the chip (integrated circuit) that theHIPCF canceller 130 is mounted or fabricated on to measure the temperature of the chip. Thecontroller 235 can receive temperature measurements from thetemperature sensor 755 and use these measurements for monitoring, calibration, and for temperature compensation. In certain exemplary embodiments, the output of thetemperature sensor 755 is coupled to an A/D converter, such as A/D converter 750 or a second A/D converter. In exemplary embodiments having a shared A/D converter 750 for thepower detector 745 and thetemperature sensor 755, thecontroller 235 can provide a signal to the A/D converter requesting which of the two measurements (power or temperature) to obtain. -
FIG. 8 depicts a functional block diagram of acommunication system 800, in accordance with certain exemplary embodiments. Theexemplary communication system 800 includes twocommunication devices transmitter receiver communication system 800 includes afirst HIPCF canceller 880 for compensating for noise and/or interference imposed onto an input of thereceiver 865 from signals transmitted by thetransmitter 810 via afirst antenna 825. Thecommunication system 800 also includes asecond HIPCF canceller 885 for compensating for noise and/or interference imposed onto an input of thereceiver 820 from signals transmitted by thetransmitter 855 via asecond antenna 870. Thus, thecommunication system 800 includes interference compensation circuits for protecting bothcommunication devices communication device 805 may be a cellular radio and thecommunication device 850 may be a WiFi radio. In this example, the cellular radio would be protected from interference imposed on the cellular radio receiver caused by signals transmitted by the WiFi radio and, conversely, the WiFi radio would be protected from interference imposed on the WiFi receiver from signals transmitted by the cellular radio. - The
HIPCF canceller 880 receives samples of signals transmitted by thetransmitter 810 via asampling device 890 electrically coupled to the output of the transmitter'spower amplifier 815 and processes those samples to generate an interference compensation signal. TheHIPCF canceller 880 applies the generated interference compensation signal to the input of thereceiver 865 atcancellation point 833 and, in turn, the interference compensation signal cancels, suppresses, or otherwise compensates for noise and/or interference imposed on thereceiver 865. TheHIPCF canceller 880 can include a controller similar tocontroller 235 ofFIG. 2 that executes one or more calibration and one or more tuning algorithms to improve the noise and/or interference compensation. The controller can receive feedback, such as a “receive signal quality indicator,” and use the feedback during the execution of the algorithms to improve the noise and/or interference compensation. Similar to thecancellation point 134, thecancellation point 833 can be implemented as converging electrical conductors, a coupler, a summation node, an adder, or other suitable technology. - Similarly, the
HIPCF canceller 885 receives samples of signals transmitted by thetransmitter 855 via asampling device 895 electrically coupled to the output of the transmitter'spower amplifier 860 and processes those samples to generate an interference compensation signal. TheHIPCF canceller 885 applies the generated interference compensation signal to the input of thereceiver 820 atcancellation point 834 and, in turn, the interference compensation signal cancels, suppresses, or otherwise compensates for noise and/or interference imposed on thereceiver 820. TheHIPCF canceller 885 can include a controller similar tocontroller 235 ofFIG. 2 that executes one or more calibration and one or more tuning algorithms to improve the noise and/or interference compensation. The controller can receive feedback, such as a “receive signal quality indicator,” and use the feedback during the execution of the algorithms to improve the noise and/or interference compensation. Similar to thecancellation point 134, thecancellation point 834 can be implemented as converging electrical conductors, a coupler, a summation node, an adder, or other suitable technology. -
FIG. 9 depicts a lookup table 900, in accordance with certain exemplary embodiments. Referring toFIGS. 2 , 7, and 9, the lookup table 900 can be stored in thememory device 760 of theHIPCF canceller 130. The exemplary lookup table 900 includescenter frequency settings 910 for the Input-BPF 205,center frequency settings 920 for the LNA-BPF 215, andcenter frequency settings 930 for the Q-Enhanced-BPF 225. In this exemplary embodiment, the Input-BPFcenter frequency settings 910 include three frequency values (Freq1, Freq2, and Freq3) for which the band-pass filters pass filters mobile TV receiver 135 embodiment. The Input-BPFcenter frequency settings 910 also include switched capacitor array settings (SCA_Input_BPF1-SCA-Input_BPF3) for each of the three frequency values (Freq1-Freq3), respectively. The switched capacitor array settings (SCA_Input_BPF1-SCA-Input_BPF3) control how theSCA 305 and theSCA 310 are controlled for each of the frequencies (Freq1-Freq3) and thus, the resonant frequency of the Input-BPF 205 for those frequencies. The Input-BPFcenter frequency settings 910 also include temperature coefficient values (Tempco1-Tempco3) for each frequency value (Freq1-Freq3), respectively. The temperature coefficient values (Tempco1-Tempco3) are used by thecontroller 235 to adjust the settings of theSCA - Similarly, the LNA-BPF
center frequency settings 920 includes switched capacitor array settings (SCA_LNA_BPF1-SCA-LNA_BPF3) for each of the three frequency values (Freq1-Freq3), respectively. The switched capacitor array settings (SCA_LNA_BPF1-SCA-LNA_BPF3) control how the SCA 315 is controlled for each of the frequencies (Freq1-Freq3) and thus, the resonant frequency of the LNA-BPF 215 for those frequencies. The LNA-BPFcenter frequency settings 920 also include temperature coefficient values (Tempco1-Tempco3) for each frequency value (Freq1-Freq3), respectively. These temperature coefficient values (Tempco1-Tempco3) are used by thecontroller 235 to adjust the settings of the SCA 315 based on changes in temperature. - The Q-Enhanced-BPF
center frequency settings 930 include switched capacitor array settings (SCA_QE_BPF1-SCA-QE_BPF3) for each of the three frequency values (Freq1-Freq3), respectively. The switched capacitor array settings (SCA_QE_BPF1-SCA-QE_BPF3) control how theSCA 615 is controlled for each of the frequencies (Freq1-Freq3) and thus, the resonant frequency of the Input-BPF 205 for those frequencies. The Q-Enhanced-BPFcenter frequency settings 920 also include temperature coefficient values (Tempco1-Tempco3) for each frequency value (Freq1-Freq3), respectively. These temperature coefficient values (Tempco1-Tempco3) are used by thecontroller 235 to adjust the settings of theSCA 615 based on changes in temperature. The Q-Enhanced-BPFcenter frequency settings 930 also include DAC settings (DAC1-DAC3) for the voltage controlled capacitors VC1 and VC2 for each frequency (Freq1-Freq3), respectively. The Q-Enhanced-BPFcenter frequency settings 920 also include temperature coefficient values (CurrentTempco1-CurrentTempco3) for each frequency value (Freq1-Freq3), respectively. These temperature coefficient values (CurrentTempco1-CurrentTempco3) are used by thecontroller 235 to adjust the settings of the current switches M70-M7 n, and thus, the bias current in the Q-enhanced-BPF 225 based on changes in temperature. - The exemplary lookup table 900 also includes seed values 940 for the I/
Q modulator 230. The seed values 940 include in-phase and quadrature (I, Q) settings ((I1, Q1)−(I3, Q3)) for the I/Q modulator 230 at each frequency (Freq1-Freq3), respectively. The lookup table 900 also includesmiscellaneous settings 950. Themiscellaneous settings 950 include the temperature at which a calibration of theHIPCF canceller 130 was performed, the process parameters of the lot theHIPCF canceller 130 was fabricated in, the temperature coefficient of the settings of theDAC 650, the minimum current required to keep the transistor switches M8 and M9 of the Q-Enhanced-BPF 225 turned on, and the threshold of detecting oscillation for the on-chip power detector 745. - The lookup table 900 is stored on the
memory device 760 and accessed by thecontroller 235 to adjust the settings of certain components within theHIPCF canceller 130 during normal operation and during calibration and tuning processes discussed below. Many of the settings in the lookup table 900 are also populated during these calibration and tuning processes, as discussed in further detail below. -
FIG. 10 is a flow chart depicting amethod 1000 for calibrating certain components of theHIPCF canceller 130, in accordance with certain exemplary embodiments. After fabrication of theHIPCF canceller 130, for example in an integrated circuit, initial settings shown in the lookup table 900 ofFIG. 9 are populated during an ATE or bench characterization process inblock 1005. Inblock 1010, in the application stage when theHIPCF canceller 130 is powered on, the values for the settings in the lookup table 900 are loaded into an internal register of thecontroller 235. Thecontroller 235 can access the lookup table 900 and control the components of theHIPCF canceller 130 using a current temperature measurement from thetemperature sensor 755 and the channel frequency that thereceiver 135 is tuned to. An optional calibration routine may also be performed inblock 1010 to calibrate the band-pass filters Q modulator 230. - In
block 1015, if the channel of thereceiver 135 changes, the I/Q modulator 230 is recalibrated by thecontroller 235. This recalibration can improve the noise and/or interference cancellation based on the receiver's receive signal quality indicator and cancellation algorithms described below. Inblock 1020, thecontroller 235 triggers the calibration of the band-pass filters Q modulator 230 in response to a command from a user or in response to the temperature change exceeding a preset threshold, for example 10 degrees C. During the calibration process of themethod 1000, the values in the lookup table 900 are updated. -
FIG. 11 is a flow chart depicting amethod 1100 for configuring the filters of theHIPCF canceller 130 for a desired center frequency (e.g., 450 MHz, 600 MHz, or 770 MHz for a mobile TV embodiment), in accordance with certain exemplary embodiments. Inblock 1105, the Input-BPF 205 is calibrated. The LNA-BPF 215 and the Q-Enhanced-BPF 225 are bypassed by activatingbypass switches bypass switch 720. A pilot tone or tuner signal is applied to the input of theHIPCF canceller 130 and the power level of the pilot tone or tuner signal is measured at the output of theHIPCF canceller 130. The settings of theSCA 305 and theSCA 310 are adjusted based on the measured power level until the power level reaches an acceptable level. The settings of theSCA 305 and theSCA 310 corresponding to the acceptable power level are populated in the lookup table 900 for later use by thecontroller 235.Block 1105 is discussed in further detail below with reference toFIG. 12 . - In
block 1110, the LNA-BPF 215 is calibrated. The Input-BPF 205 and the Q-Enhanced-BPF 225 are bypassed by activatingbypass switches bypass switch 725. With the pilot tone or tuner signal still applied to the input of theHIPCF canceller 130, the settings of the SCA 315 are adjusted based on the measured power level until the measured power level reaches an acceptable level. The settings of the SCA 315 corresponding to the acceptable power level are populated in the lookup table 900 for later use by thecontroller 235.Block 1110 is discussed in further detail below with reference toFIG. 13 . Inblock 1115, the Q-Enhanced-BPF 225 is calibrated.Block 1115 is discussed in further detail below with reference toFIG. 14 . - In
block 1120, the temperature coefficients for the band-pass filters pass filters controller 235 can calculate the temperature coefficients by taking the difference between the settings for each band-pass-filter fields - In
block 1125, the I and Q seed values for the I/Q modulator 230 are calibrated. In certain exemplary embodiments, the ATE (or bench measurement equipment) can employ a setup similar to thecircuit 100 depicted inFIG. 1 . Thetransmitter 105 can be activated and one or more of the cancellation algorithms discussed below can be executed to identify a preferred or acceptable cancellation point for the desired center frequency. The (I, Q) settings corresponding to the identified cancellation point can be stored infield 940 of the lookup table 900. - After
block 1125, themethod 1100 ends. Of course, themethod 1100 could be executed more than one time. For example, themethod 1100 may be executed during ATE and then executed again after the chip or system being placed into operation. -
FIG. 12 is a flow chart depicting amethod 1105 for calibrating the Input-BPF 205 of theHIPCF canceller 130, in accordance with certain exemplary embodiments, as referenced inFIG. 11 . Inblock 1205, the bypass switches 725 and 670 are activated andbypass switch 720 is deactivated. This bypasses the LNA-BPF 215 and the Q-Enhanced-BPF 225 for the calibration of the Input-BPF 205. In certain exemplary embodiments, thecontroller 235 operates the bypass switches 670, 720, and 725 in response to a command to configure the band-pass filters - In
block 1210, a pilot tone or a tuner signal (e.g., a mobile TV signal) with the desired center frequency (e.g., 450 MHz, 600 MHz, or 770 MHz) is applied to the input of theHIPCF canceller 130. In certain exemplary embodiments, theHIPCF canceller 130 generates the pilot tone or tuner like signal using an on-chip phase locked loop. In certain exemplary embodiments, the pilot tone or tuner signal is generated by re-using the phase locked loop of thereceiver 135, for example via one of the receiver's output pins. - In
block 1215, the power level of the pilot tone or tuner signal is measured at the output of theHIPCF canceller 130. In certain exemplary embodiments, the output power level of the pilot tone or tuner signal is measured using ATE or bench characterization equipment. For example, the ATE or bench characterization equipment may include a spectrum analyzer. In certain exemplary embodiments, the output power level of the pilot tone or tuner signal is measured using a receive signal quality indicator obtained from thereceiver 135. In certain exemplary embodiments, the output power level of the pilot tone or tuner signal is measured using thepower detector 745. - In
block 1220, thecontroller 235 makes one or more adjustments to the settings of theSCA 305 and theSCA 310 and measures the output power level of the pilot tone or tuner signal resulting from each adjustment. Thecontroller 235 can continue to make adjustments until the output power level of the pilot tone or tuner signal reaches or exceeds an acceptable, preferred, or maximum level. In addition or in the alternative, thecontroller 235 can make a certain number of adjustments and record the output power level of the pilot tone or tuner signal (e.g., in memory device 760) and identify the recorded output power level having the best, preferred, or highest power level. In certain exemplary embodiments, thecontroller 235 sweeps the setting values for theSCA 305 and theSCA 310 in a monotonically increasing or decreasing process (e.g., one least significant bit (“LSB”) or multiple LSBs at a time for digital SCAs). In certain exemplary embodiments, a binary algorithm, such as the algorithm illustrated inFIG. 20 and discussed below, could be used to find a preferred setting for theSCA 305 andSCA 310. - In
block 1225, thecontroller 235 stores the desired center frequency and the settings for theSCA 305 andSCA 310 corresponding to the acceptable, preferred, or maximum level in the lookup table 900 in thememory device 760. For example, the desired center frequency may be stored in the field “Freq1” and the settings for theSCA 305 and theSCA 310 may be stored in field “SCA_Input_BPF1.” Afterblock 1225, themethod 1105 proceeds to block 1110, as referenced inFIG. 11 . -
FIG. 13 is a flow chart depicting amethod 1110 for calibrating the LNA-BPF 215 of theHIPCF canceller 130, in accordance with certain exemplary embodiments, as referenced inblock 1110 ofFIG. 11 . Inblock 1305, the bypass switches 720 and 670 are activated andbypass switch 725 is deactivated. This bypasses the Input-BPF 205 and the Q-Enhanced-BPF 225 for the calibration of the LNA-BPF 215. - In
block 1310, thecontroller 235 makes one or more adjustments to the settings of the SCA 315 and measures the output power level of the pilot tone or tuner signal resulting from each adjustment. Thecontroller 235 can continue to make adjustments until the output power level of the pilot tone or tuner signal reaches or exceeds an acceptable, preferred, or maximum level. In addition or in the alternative, thecontroller 235 can make a certain number of adjustments and record the output power level of the pilot tone or tuner signal (e.g., in memory device 760) and identify the recorded output power level having the best, preferred, or highest power level. In certain exemplary embodiments, thecontroller 235 sweeps the setting values for the SCA 315 in a monotonically increasing or decreasing process (e.g., one LSB or multiple LSBs at a time for digital SCAs). In certain exemplary embodiments, a binary algorithm, such as the algorithm illustrated inFIG. 20 and discussed below, could be used to find a preferred setting for the SCA 315. - In
block 1315, thecontroller 235 stores the settings for the SCA 315 corresponding to the acceptable, preferred, or maximum level in the lookup table 900 in thememory device 760. For example, the settings for the SCA 315 may be stored in field “SCA_LNA_BPF1.” Afterblock 1315, themethod 1110 proceeds to block 1115, as referenced inFIG. 11 . -
FIGS. 14A and 14B , collectivelyFIG. 14 , depict a flow chart of amethod 1115 for calibrating the Q-Enhanced-BPF 225 of theHIPCF canceller 130, in accordance with certain exemplary embodiments, as referenced inblock 1115 ofFIG. 11 . Inblock 1405, the bypass switches 720 and 725 are activated andbypass switch 670 is deactivated. This bypasses the Input-BPF 205 and the LNA-BPF 215 for the calibration of the Q-Enhanced-BPF 225. - In
block 1410, a bias current is applied (e.g., by the controller 235) to the current switches M70-M7 n for the purpose of keeping the transistor switches M8 and M9 in thecross-coupled pair 620 and the current sources M60-M6 n active or turned on, yet avoiding oscillation of the Q-Enhanced-BPF 225. The amount of current applied to the current switched M70-M7 n may correspond to the value of the “Minimum Current for QE” field of the lookup table 900. - In
block 1415, thecontroller 235 makes one or more adjustments to the settings of theSCA 615 and measures the output power level of the pilot tone or tuner signal resulting from each adjustment. Thecontroller 235 can continue to make adjustments until the output power level of the pilot tone or tuner signal reaches or exceeds an acceptable, preferred, or maximum level. In addition or in the alternative, thecontroller 235 can make a certain number of adjustments and record the output power level of the pilot tone or tuner signal (e.g., in memory device 760) and identify the recorded output power level having the best, preferred, or highest power level. In certain exemplary embodiments, thecontroller 235 sweeps the setting values for theSCA 615 in a monotonically increasing or decreasing process (e.g., one LSB or multiple LSBs at a time for digital SCAs). In certain exemplary embodiments, a binary algorithm, such as the algorithm illustrated inFIG. 20 and discussed below, could be used to find a preferred setting for theSCA 615. - In
block 1420, thecontroller 235 increases the amount of current applied to the cross-coupled transistor switches M8, M9 by increasing the settings of the current switches M70-M7 n. In certain exemplary embodiments, the amount of current is increased by a few (e.g., 4) LSB. Inblock 1425, the pilot tone or tuner signal is turned off. Inblock 1430, an inquiry is conducted by thecontroller 235 as to whether there is any oscillation generated by the Q-Enhanced-BPF 225. In certain exemplary embodiments, this inquiry includes comparing the measured output power level of theHIPCF canceller 130 with a predetermined threshold value at the ATE or a threshold value “Power Detector Output Threshold for Oscillation” stored in themiscellaneous values 950 of the lookup table 900. If the measured output power level is below the threshold, then thecontroller 235 determines that there is no oscillation. If thecontroller 235 determines that there is no or sufficiently low oscillation, themethod 1115 proceeds to block 1435, where thecontroller 235 turns on the pilot tone or tuner signal again and applies the pilot tone or tuner signal to the input of theHIPCF canceller 130. Afterblock 1435, themethod 1115 returns to block 1415. If thecontroller 235 determines that there is oscillation, themethod 1115 proceeds to block 1440. - In
block 1440, thecontroller 235 decreases the amount of current applied to the current switches M70-M7 n to the level prior to oscillation. Inblock 1445, thecontroller 235 stores the settings for theSCA 615 corresponding to current level prior to oscillation in the lookup table 900 in thememory device 760. For example, the settings for theSCA 615 may be stored in field “SCA_QE_BPF1.” - In
block 1450, the pilot tone or tuner signal is reactivated and applied to the input of theHIPCF canceller 130. Thecontroller 235 makes one or more adjustments to the settings for theDAC 650 for biasing the voltage controlled capacitors VC1 and VC2 and measures the output power level of the pilot tone or tuner signal resulting from each adjustment. The adjustments to theDAC 650 adjust the voltage level at the voltage controlled capacitors VC1 and VC2. Thecontroller 235 can continue to make adjustments until the output power level of the pilot tone or tuner signal reaches or exceeds an acceptable, preferred, or maximum level. In addition or in the alternative, thecontroller 235 can make a certain number of adjustments and record the output power level of the pilot tone or tuner signal (e.g., in memory device 760) and identify the recorded output power level having the best, preferred, or highest power level. In certain exemplary embodiments, thecontroller 235 sweeps the setting values for theDAC 650 in a monotonically increasing or decreasing process (e.g., one LSB or multiple LSBs at a time). In certain exemplary embodiments, a binary algorithm, such as the algorithm illustrated inFIG. 20 and discussed below could be used to find a preferred setting for theDAC 650. - In
block 1455, the pilot tone or tuner signal is turned off. Inblock 1460, the output power level of theHIPCF canceller 130 is measured. Inblock 1465, an inquiry is conducted by thecontroller 235 as to whether there is any oscillation generated by the Q-Enhanced-BPF 225, similar to block 1430. If thecontroller 235 determines that there is oscillation, themethod 1115 proceed to block 1470. If thecontroller 235 determines that there is no oscillation, themethod 1115 proceeds to block 1475. - In
block 1470, thecontroller 235 lowers the current level for biasing the cross-coupled transistors M8, M9 by decreasing the settings of current switches M70-7 n, for example by a few LSB. After the current level is lowered, themethod 1115 returns to block 1450. - In
block 1475, thecontroller 235 stores the settings for theDAC 650 and the current switches M70-M7 n in the lookup table 900 in thememory device 760. For example, the setting for the current switches M70-M7 n may be stored in the field “Currentl” and the setting for theDAC 650 may be stored in field “DAC1.” Afterblock 1475, themethod 1115 ends. Of course, themethod 1100 can be repeated any number of times for any number of frequencies. For example, the band-pass filters -
FIG. 15 is a flow chart depicting amethod 1500 for calibrating the Input-BPF 205 of theHIPCF canceller 130 ofFIG. 7 , in accordance with certain exemplary embodiments. Thismethod 1500 is an alternative method to that of themethod 1105 ofFIG. 12 . Inblock 1505, the bypass switches 720, 725, and 670 are deactivated. For example, thecontroller 235 may deactivate the bypass switches 720, 725, and 670. - In
block 1510, a pilot tone or tuner signal with the desired center frequency is applied to the input of theHIPCF canceller 130. Inblock 1515, a measurement of the reflected pilot tone or tuner signal (e.g., reflection coefficient or return loss) is made at the input of theHIPCF canceller 130. This measurement may be made by a power detector or spectral analyzer, for example. Inblock 1520, thecontroller 235 makes one or more adjustments to the settings of theSCA 305 and theSCA 310 and measures the reflected pilot tone or tuner signal. Thecontroller 235 can continue to make adjustments until the reflected pilot tone or tuner signal reaches or exceeds an acceptable, preferred, or minimum level. In addition or in the alternative, thecontroller 235 can make a certain number of adjustments and record the reflected pilot tone or tuner signal (e.g., in memory device 760) and identify the recorded reflected pilot tone or tuner signal having the best, preferred, or lowest level. In certain exemplary embodiments, thecontroller 235 sweeps the setting values for theSCA 305 and theSCA 310 in a monotonically increasing or decreasing process (e.g., one LSB or multiple LSBs at a time for digital SCAs). In certain exemplary embodiments, a binary algorithm, such as the algorithm illustrated inFIG. 20 and discussed below could be used to find a preferred setting for theSCA 305 andSCA 310. - In
block 1525, thecontroller 235 stores the desired center frequency and the settings for theSCA 305 andSCA 310 corresponding to the acceptable, preferred, or minimum level in the lookup table 900 in thememory device 760. For example, the desired center frequency may be stored in the field “Freq1” and the settings for theSCA 305 and theSCA 310 may be stored in field “SCA_Input_BPF1.” -
FIG. 16 is a flow chart depicting amethod 1600 for determining switch settings for a given frequency, in accordance with certain exemplary embodiments. For example, themethod 1600 may be performed in response to a user applying a channel change for a mobile TV. In certain exemplary embodiments, the lookup table 900 includes the settings for each band-pass filter receiver 135 may tune to. In certain exemplary embodiments, the lookup table 900 includes the settings for each band-pass filter method 1600 provides an exemplary process for computing the SCA switch settings for each band-pass filter exemplary method 1600 takes into account the calibration values in the lookup table 900 identified for the predetermined number of channel frequencies and the actual temperature measured by an on-chip temperature sensor, such astemperature sensor 755. - In
block 1605, thecontroller 235 conducts an inquiry to determine whether to start determining switch settings for each band-pass filter controller 235 communicates with thereceiver 135 to determine whether the receive frequency for thereceiver 135 has changed, for example as a result of a change in channel for thereceiver 135. If the receive frequency for the receiver has changed, thecontroller 235 determines to start determining the switch settings for each band-pass filter method 1600 remains inblock 1605. - In certain exemplary embodiments, the
controller 235 determines whether the temperature of the chip that theHIPCF canceller 130 resides has changed. Thecontroller 235 monitors the temperature measurement received to determine whether the temperature has changed by a certain threshold. If thecontroller 235 determines that the temperature has changed by an amount equal to or exceeding the threshold, thecontroller 235 determines to start determining the switch settings for each band-pass filter method 1600 remains inblock 1605. - In certain exemplary embodiments, the
controller 235 determines whether the lookup table 900 has changed or whether a setting or value in the lookup table 900 has been updated. If thecontroller 235 determines that the lookup table has changed, thecontroller 235 determines to start determining the switch settings for each band-pass filter method 1600 remains inblock 1605. - In
block 1610, thecontroller 235 receives the receive frequency for the receiver 135 (“target frequency”), the current calibration values for the band-pass filters temperature sensor 755, and the temperature value during calibration from the lookup table 900. - In
block 1615, thecontroller 235 conducts an inquiry to determine whether the target frequency is less than a frequency threshold. For example in certain mobile TV embodiments, this frequency threshold is set at 600 MHz which corresponds to the middle of the receive band of certain mobile TV tuners. If the target frequency is less than the frequency threshold, themethod 1615 proceeds to block 1620. Otherwise, themethod 1600 proceeds to block 1625. - In
block 1620,controller 235 computes a variable “DeltaF” indicating the difference between the target frequency and the frequency threshold. Thecontroller 235 performs an interpolation process, for example linear interpolation, using two or more of the calibration values in the lookup table 900 to determine the settings for theSCAs controller 235 uses the setting stored for a first frequency, such as Freq1, and the setting stored for a second frequency, such as Freq2, in a linear interpolation calculation with DeltaF to determine the setting for that component. - In
block 1625, the controller computes the variable DeltaF indicating the difference between the target frequency and a second frequency value. In certain exemplary embodiments, if the frequency threshold if 600 MHz, the second frequency value is 770 MHz. These frequency values are exemplary, rather than limiting, and other frequency values can be used without departing from the scope and spirit of the present invention. Similar to block 1620, the controller performs an interpolation process using two or more of the calibration values in the lookup table 900 to determine the settings for theSCAs controller 235 uses the setting stored for a first frequency, such as Freq2, and the setting stored for a second frequency, such as Freq3, in a linear interpolation calculation with DeltaF to determine the setting for that component. - As shown in
blocks method 1600 uses two different sets of calibrated settings to determine the settings for the components of theHIPCF canceller 130 depending upon the target frequency. This enables thecontroller 235 to use the calibrated settings nearest the target frequency to determine the appropriate settings for the components. - In
block 1630, thecontroller 235 determines a temperature compensation by computing a variable “DeltaTemp” which yields the difference between actual temperature and the temperature at which the last calibration was performed (stored infield 950 of the lookup table 900). Thecontroller 235 also computes offset values caused by the temperature difference for the settings for each component. Thecontroller 235 uses the offset values to determine final setting for the components at the target frequency. Thecontroller 235 stores the final settings in internal registers for use in operating the components. Note that the I and Q settings for the I/Q modulator 230 may not be temperature compensated in themethod 1600 as the I and Q settings may be calibrated using one of the cancellation algorithms discussed below with reference toFIGS. 17-31 . -
FIG. 17 depictsimplementation layers 1700 of noise and/or interference cancellation algorithms, in accordance with certain exemplary embodiments. These algorithms can use a feedback signal from thevictim receiver 135 to determine appropriate I and Q settings for theHIPCF canceller 130. This feedback signal includes a quality indicator (e.g., BER, PER, RSSI, noise floor, SNR, EVM, and Position Accuracy, etc.) for thecommunication system 100. This exemplary implementation of algorithms includes four layers, alink control layer 1710, asignal processing layer 1720, analgorithm control layer 1730, and analgorithm execution layer 1740. In certain exemplary embodiments, each of the layers 1710-1740 may reside in any of the following three components: 1) a baseband integrated circuit of thevictim receiver 135, 2) a stand alone microcontroller, or 3) the on-chip controller 235 (or another control device) of theHIPCF 130. For ease of discussion, the layers 1710-1740 will be discussed hereinafter in terms of thecontroller 235 performing the respective functions. - In the
link control layer 1710, the feedback signal is analyzed and tested for quality to determine whether cancellation should be activated to improve the sensitivity of thevictim receiver 135. Due to the nature of the active noise and/or interference cancellation which theHIPCF canceller 130 provides, theHIPCF canceller 130 may also output its own noise floor while canceling the noise and/or interference generated by the power amplifier 110 (or another component) at the input of thevictim receiver 135. As a result, the overall noise floor seen by thevictim receiver 135 is the summation of the output noise floor of theHIPCF canceller 130, the receivingantenna 120, the power amplifier noise and/or interference received by the receivingantenna 120, and the phase and gain adjusted noise floor of the power amplifier 110 (via the HIPCF canceller 130), which in turn can affect the sensitivity of thevictim receiver 135. Thus, the determination as to whether or not to activate theHIPCF canceller 130 to improve the sensitivity of thevictim receiver 135 may be decided based on the actual noise and/or interference of thepower amplifier 110 received by thevictim receiver 135. -
FIG. 18 depicts a diagram 1800 of receiver sensitivity plotted versus coupled power amplifier noise for a mobile TV tuner tuned at 746 MHz with a channel bandwidth of 8 MHz and a CDMA800 power amplifier, in accordance with certain exemplary embodiments. Referring toFIG. 18 , the diagram 1800 includes afirst curve 1805 depicting the mobile TV tuner sensitivity with theHIPCF canceller 130 inactive and asecond curve 1810 depicting the mobile TV tuner sensitivity with theHIPCF canceller 130 canceling or suppressing the power amplifier noise. As illustrated in the exemplary implementation, there is no advantage to activating theHIPCF noise canceller 130 for power amplifier noise below −174 dBm/Hz as the output noise floor of theHIPCF canceller 130 would exceed the benefit of canceling the coupled power amplifier noise. For power amplifier noise above approximately −160 dBm/Hz, maximum cancellation/sensitivity improvement (e.g., approximately 10 dB in this exemplary implementation) would be achieved with theHIPCF canceller 130 active as the received power amplifier noise is typically much higher than the output noise floor of theHIPCF canceller 130. Additionally, thelink control layer 1710 detects in multi-channel systems whether a particular channel has been optimized in the past and passes the setting corresponding to prior optimization from the memory to theHIPCF 130. An indication of whether the channel has been optimized previously can be stored in memory by thecontroller 235 at the conclusion of or during an optimization. - The desired victim receive signal quality may be assessed with respect to feedback (e.g., BER, PER, RSSI, noise floor, SNR, EVM, and Position Accuracy, etc.) received from the receiver to determine whether to activate the
HIPCF canceller 130. For example, theHIPCF 130 may be activated if the feedback indicates that the receive signal is above the combined noise floor (i.e., the summation of the output noise floor of theHIPCF canceller 130, the receivingantenna 120, the power amplifier noise and/or interference received by the receivingantenna 120, and the phase and gain adjusted noise floor of the power amplifier 110 (via the HIPCF canceller 130). This feature is illustrated inFIG. 19 , which depicts a diagram 1900 of an output SNR of a mobile TV tuner tuned at 746 MHz with a channel bandwidth of 8 MHz versus received mobile TV signal strength with a coupled CDMA800 power amplifier phase noise at −161 dBm/Hz at the input ofvictim receiver 135, in accordance with certain exemplary embodiments. Referring toFIG. 19 , the diagram 1900 includes afirst curve 1905 depicting the mobile TV tuner output SNR with theHIPCF canceller 130 inactive, and asecond curve 1910 depicting the mobile TV tuner output SNR with theHIPCF canceller 130 canceling or suppressing the power amplifier phase noise. Athird curve 1915 depicts a desired minimum SNR output for the mobile TV tuner. As illustrated in the exemplary implementation, there may not be an advantage in activating theHIPCF noise canceller 130 when the received mobile TV signal is below −90 dBm. - The exemplary
link control layer 1710 includes several modes of operation. Thecontroller 235 can determine which mode of operation to be active based on the signal quality of the signal received by thereceiver 135. In certain exemplary embodiments, thelink control layer 1710 includes four modes of operation, a maximum cancellation mode, a limited cancellation mode, a wait for acceptable signal mode, and a no signal mode. In this exemplary embodiment, if the received signal strength is acceptable (e.g., above an acceptable threshold level which is −81 dBm inFIG. 19 ), thecontroller 235 commences the maximum cancellation mode whereby the noise and/or interference cancellation of theHIPCF canceller 130 is at a high level. If the received signal strength is low (e.g., between an acceptable level threshold and a very low level threshold, which is between −81 dBm and −90 dBm inFIG. 19 ), thecontroller 235 commences the limited cancellation mode of theHIPCF canceller 135 whereby the noise and/or interference cancellation level is bounded by the noise floor. If the received signal is lower than a threshold indicating, for example, a very low signal (e.g. around −90 dBm inFIG. 19 ), thecontroller 235 will commence the wait for acceptable signal mode whereby thecontroller 235 delays entering the subsequent layers 1720-1740 until the received signal meets or exceeds the threshold. If the received signal meets or exceeds the threshold while thecontroller 235 is in the wait for acceptable signal mode, thecontroller 235 can enter the subsequent layers 1720-1740. If there is no signal received at thereceiver 135, then thecontroller 235 commences the no signal mode whereby theHIPCF canceller 130 is inactive. - The
link control layer 1710 also may deduct information on the time dependent passing of the thresholds, for example when both threshold levels are passed in less than one second. A mobile device may be transported into a tunnel or under a bridge and all settings can be held constant until the passing of the acceptable level threshold indicates that the mobile device has returned from the tunnel or bridge. The operation of thelink control layer 1710 can then resume using the held settings. - The
signal processing layer 1720 includes several processes that ensure stability and robustness of the feedback signal. A first process includes averaging a predetermined number of feedback values of the feedback signal before executing a noise cancellation algorithm. - A second process includes correction for errors in feedback signals during the execution of a noise cancellation algorithm. One exemplary error correction process includes obtaining two feedback values from the
receiver 135 and computing the difference between the two feedback values. If this difference is less than a tolerance level, then the two feedback values are averaged. Otherwise, a third feedback value is obtained from thereceiver 135 and the difference between the third feedback value and the second feedback value is determined. If this difference is less than the tolerance level, then the second and third feedback values are averaged. Otherwise, a fourth feedback value is obtained and a similar process is performed for a predetermined number of iterations. If no two feedback values are found that have a difference less than the tolerance level, then an error may be indicated and theHIPCF canceller 130 may be deactivated. A second exemplary error correction process includes ranking a certain number of feedback values and selecting a certain number of the feedback values while a noise cancellation algorithm is running. For example, thecontroller 235 may rank ten feedback values and select the five feedback values ranked in the middle. The average of the selected feedback signals are calculated and used in the noise cancellation algorithms. - A third process of the
signal processing layer 1720 includes SNR averaging. This SNR averaging process includes computing the average value of the SNRs for different satellites (SVs), for example GPS systems, DARS (Digital Audio Radio Service), or Iridium. The SNR averaging may be performed for satellites that have a certain elevation level above an elevation threshold only, to avoid an incorrect decision in thealgorithm execution layer 1740. - In the
algorithm control layer 1730, several user controls can be implemented to control the algorithms described in thealgorithm execution layer 1740. One such user control is the polarity of the logic used to compare two feedback values, for example before and after a change of I and/or Q settings of theHIPCF canceller 130. The polarity can either be positive (e.g., higher feedback value is better) or negative (e.g., lower feedback value is better). Some exemplary feedback signals where a positive polarity may be used are SNR, Carrier to Noise Ratio (C/N), and Repeater Amplifier Gain. Some exemplary feedback signals where a negative polarity may be used are PER, BER, Error Vector Magnitude, Noise Floor Level, Adjacent Channel Power Ratio, and Adjacent Channel Leakage Ratio. - The
algorithm execution layer 1740 includes the execution of one of several noise cancellation algorithms. These algorithms include acts to adjust the I and Q values of theHIPCF canceller 130 and evaluate the feedback signal resulting from the adjustment to find acceptable I and Q values for operating theHIPCF canceller 130. The algorithms include two types of binary algorithms (a fast binary algorithm (FBA) and a binary correction algorithm (BCA)), a minstep algorithm (MSA), a blind shot algorithm (BSA), a dual slope algorithm (DSA), and a track and search algorithm (TSA). -
FIG. 20 is a flow chart depicting afast binary algorithm 2000 for canceling noise and/or interference, in accordance with certain exemplary embodiments. In thisexemplary FBA 2000, each bit in I and Q values for theHIPCF canceller 130 are sequentially reversed and tested for a better feedback value as determined by the polarity defined in thealgorithm control layer 1730. TheFBA 2000 may start with a start bit and progress sequentially through each of the bits of the I-value and Q-value until reaching a pre-defined stop bit. In certain exemplary embodiments, the start bit and stop bit may be user selected. - In
block 2005, thecontroller 235 selects a first I-value and a first Q-value for operating theHIPCF canceller 130. These first values may be start values from the lookup table 900, seed values, or middle of range values. Inblock 2010, theHIPCF canceller 130 applies the first I-value and first Q-value to the I/Q modulator 230. - In
block 2015, thereceiver 135 provides a feedback signal having a feedback value to thecontroller 235. The feedback value may be an SNR, an RSSI, a Carrier to Noise Ratio (C/N), RSSI, a Repeater Amplifier Gain, a PER, a BER, an Error Vector Magnitude, a Noise Floor Level, an Adjacent Channel Power Ratio, or an Adjacent Channel Leakage Ratio. After obtaining the feedback value from thereceiver 135, thecontroller 235 stores the feedback value in memory. - In
block 2020, thecontroller 235 inverts a bit of the I-value and transmits the updated I-value to theHIPCF canceller 130. In response, theHIPCF canceller 130 applies the updated I-value to the I/Q modulator 230. For example,bit 2075 of I-value 2071 may be inverted from a value of “1” to a value of “0.” In the first iteration of thisblock 2020, thecontroller 235 may invert the start bit of the I-value. In each subsequent iteration, the next bit may be inverted until the stop bit is completed. - In
block 2025, thecontroller 235 obtains an updated feedback value from thereceiver 135. Inblock 2030, thecontroller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in thealgorithm control layer 1730. For example, if the polarity is positive and the updated feedback value is greater than the stored feedback value, then thecontroller 235 will determine that the updated feedback value is better. Likewise, if the polarity is negative and the updated feedback value is greater than the stored feedback value, then thecontroller 235 will determine that the stored feedback value is better. Thecontroller 235 stores the better feedback value and sets the I-value to the I-value that resulted in the better feedback value. Thecontroller 235 also applies the I-value that resulted in the better feedback value to theHIPCF canceller 130. - In
block 2035, thecontroller 235 inverts a bit of the Q-value and transmits the updated Q-value to theHIPCF canceller 130. In response, theHIPCF canceller 130 applies the updated Q-value to the I/Q modulator 230. For example,bit 2085 of Q-value 2081 may be inverted from a value of “1” to a value of “0.” In the first iteration of thisblock 2035, thecontroller 235 may invert the start bit of the Q-value. In each subsequent iteration, the next bit may be inverted until the stop bit is completed. - In
block 2040, thecontroller 235 obtains an updated feedback value from thereceiver 135. Inblock 2045, thecontroller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in thealgorithm control layer 1730. Thecontroller 235 stores the better feedback value and sets the Q-value to the Q-value that resulted in the better feedback value. Thecontroller 235 also applies the Q-value that resulted in the better feedback value to theHIPCF canceller 130. - In
block 2050, thecontroller 235 conducts an inquiry to determine whether there are more bits in the I-value and Q-value to test. For example, thecontroller 235 may determine whether the previous iteration of blocks 2020-2050 evaluated the stop bit. If there are more bits to test, the “Yes” branch is followed back to block 2020 where another bit is inverted and evaluated for better feedback. Otherwise, the “No” branch is followed to block 2055. Inblock 2055, thecontroller 235 operates theHIPCF canceller 130 using the final stored I-value and Q-value. - In certain exemplary embodiments, the
FBA 2000 illustrated inFIG. 20 may not cover every condition and thus, it may be improved by assigning a one or two bit start value (e.g., most significant bit (MSB)) for both the I-value and the Q-value. Another improvement to theFBA 2000 includes executing the BSA described below prior to executing theFBA 2000 to obtain a start value for the I-value and for the Q-value. - The BCA is a modification to the fast binary algorithm illustrated in
FIG. 20 and described above. In the BCA, each bit of both the I and Q values are sequentially reversed as in the fast binary algorithm, and either increased by a value of “1” if the original value of the bit is “1” (and thus, causing a carry to its immediate neighboring more significant bit) or decreased by “1” if the original value is “0” (and thus, causing a borrow from its immediate neighboring more significant bit). In both cases, thecontroller 235 would evaluate the feedback value to determine which value (I-value or Q-value depending on the block) resulted in better feedback. Similar to the FBA, the I-value and Q-value that resulted in the better feedback value is stored and used at the completion of the algorithm to control theHIPCF canceller 130. The BCA may begin with a start bit and proceed through each bit until the stop bit is completed. In certain exemplary embodiment, the start and stop bits may be user selected. In certain exemplary embodiments, if no BSA is performed prior to the execution of the binary correction algorithm, the binary correction algorithm may start with the MSB and with bit reversal only for the MSB of the I-value and the Q-value as there is not a more significant bit to carry to or borrow from for the MSB. After the MSB for the I-value and Q-value have been completed, the feature of increasing or decreasing an evaluated bit by a value of “1” may begin with the second MSB. - The motivation for implementing the BCA in place of the
FBA 2000 can be discussed with reference toFIG. 21 , which depicts agraph 2100 of I and Q values adjusted using the binary algorithms. Referring toFIG. 21 , point X1 represents a plot of an initial I-value and Q-value for theHIPCF canceller 130. As part of the binary algorithms, the MSB of the I-value is inverted to proceed from point X1 point X2. In this graph, the feedback value at point X2 is determined to be better than the feedback value at point X1. Thus, the binary algorithms would keep the I-value for point X2 and invert the MSB of the Q-value to proceed to point X3. - At point X3, assuming the feedback value is determined to be better at point X3 than at point X2, in the
FBA 2000, the second MSB of the I-value would be inverted. This bit inversion would cause the algorithm to proceed from point X3 to point A, which is further away from optimal point C and thus, would have an inferior feedback value to that of point C. In the binary correction algorithm, the feedback value would be tested at both points A and B by increasing or decreasing the second MSB of the I-value by a value of “1” and thus, affecting the MSB. Because point B is closer to the optimal point C, point B would result in a better feedback value than point A and the BCA would continue from point B rather than point X3. Thus, the BCA can be more accurate than the fast binary algorithm. However, the BCA may require more iterations and more hardware in certain implementations. -
FIG. 22 is a flow chart depicting aminstep algorithm 2200 for canceling noise and/or interference, in accordance with certain exemplary embodiments. Theexemplary MSA 2200 can provide fine tuning to the noise cancellation, for example after one of the binary algorithms has been executed. TheMSA 2200 can follow changes in the coupling channel between an interferer/noise source (e.g., power amplifier 110) and thevictim receiver 135. For a given step size (e.g., 1 LSB to 7 LSB resolution), noise and/or interference cancellation can be achieved by incrementing (plus step size) or decrementing (minus step size) I-values and Q-values sequentially. In certain exemplary embodiments, the incrementing or decrementing stops at maximum or minimum values for the appropriate I-value or Q-value (e.g., range criteria). In certain exemplary embodiments, theMSA 2200 runs for a given number of iterations or time period and can be interrupted by a user. I-values and Q-values each oscillate around an desirable value or follow changes in coupling channel. - Referring to
FIGS. 1 , 2, and 22, inblock 2205, thecontroller 235 selects a first I-value and a first Q-value for operating theHIPCF canceller 130. Inblock 2210, theHIPCF canceller 130 applies the first I-value and the first Q-value to the I/Q modulator 230. - In
block 2215, thereceiver 135 provides a feedback signal having a feedback value to thecontroller 235. The feedback value may be a SNR, a RSSI, a Carrier to Noise Ratio (C/N), a Repeater Amplifier Gain, a PER, a BER, an Error Vector Magnitude, a Noise Floor Level, an Adjacent Channel Power Ratio, or an Adjacent Channel Leakage Ratio, etc. After obtaining the feedback value from thereceiver 135, thecontroller 235 stores the feedback value in memory. - In
block 2220, thecontroller 235 increments the I-value by a given step size (e.g., 1 LSB) and transmits the updated I-value to theHIPCF canceller 130. In response, theHIPCF canceller 130 applies the updated I-value to the I/Q modulator 230. Thecontroller 235 also obtains an updated feedback value from thereceiver 135. - In
block 2225, thecontroller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in thealgorithm control layer 1730. Thecontroller 235 stores the better feedback value and sets the I-value to the I-value that resulted in the better feedback value. Thecontroller 235 also applies the I-value that resulted in the better feedback value to theHIPCF canceller 130. - In
block 2230, thecontroller 235 increments the Q-value by a given step size (e.g., one LSB) and transmits the updated Q-value to theHIPCF canceller 130. In response, theHIPCF canceller 130 applies the updated Q-value to the I/Q modulator 230. Thecontroller 235 also obtains an updated feedback value from thereceiver 135. - In
block 2235, thecontroller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in thealgorithm control layer 1730. Thecontroller 235 stores the better feedback value and sets the Q-value to the Q-value that resulted in the better feedback value. Thecontroller 235 also applies the Q-value that resulted in the better feedback value to theHIPCF canceller 130. - In
block 2240, thecontroller 235 decrements the I-value by a given step size (e.g., one LSB) and transmits the updated I-value to theHIPCF canceller 130. In response, theHIPCF canceller 130 applies the updated I-value to the I/Q modulator 230. Thecontroller 235 also obtains an updated feedback value from thereceiver 135. - In
block 2245, thecontroller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in thealgorithm control layer 1730. Thecontroller 235 stores the better feedback value and sets the I-value to the I-value that resulted in the better feedback value. Thecontroller 235 also applies the I-value that resulted in the better feedback value to theHIPCF canceller 130. - In
block 2250, thecontroller 235 decrements the Q-value by a given step size (e.g., one LSB) and transmits the updated Q-value to theHIPCF canceller 130. In response, theHIPCF canceller 130 applies the updated Q-value to the I/Q modulator 230. Thecontroller 235 also obtains an updated feedback value from thereceiver 135. - In
block 2255, thecontroller 235 compares the updated feedback value to the stored feedback value to determine which of the two feedback values is better based on the polarity defined in thealgorithm control layer 1730. Thecontroller 235 stores the better feedback value and sets the Q-value to the Q-value that resulted in the better feedback value. Thecontroller 235 also applies the Q-value that resulted in the better feedback value to theHIPCF canceller 130. - In
block 2260, thecontroller 235 conducts an inquiry to determine whether to continue repeatingblocks 2220 through 2255. In certain exemplary embodiments, the determination is based on a time period. If the time period has expired, then thecontroller 235 determines not to continue. In certain exemplary embodiments, the determination is based on the sensitivity of thereceiver 135 or based on the feedback value obtained inblock 2255. In certain exemplary embodiments, the determination is based on the number of iterations executed. If thecontroller 235 determines to continue repeating blocks 2220-2255, then the “Yes” branch is followed back toblock 2220. Otherwise, the “No” branch is followed to block 2265. Inblock 2265, thecontroller 235 operates theHIPCF canceller 130 using the final selected I-value and Q-value. - In certain exemplary embodiments, the decision to change from an increment in I-value or Q-value to a decrement is based upon whether the previous iteration rejected the new feedback value, i.e. the new feedback value was not preferred over the previous feedback value.
- Although the
FBA 2000, theBCA 2100, and theMSA 2200 have been discussed above in terms of a changing sequence of IQIQIQ, theFBA 2000, theBCA 2100, and theMSA 2200 could also be implemented using other sequences, including IIQQIIQQ, and IIIQQQIIIQQQ, for example. - The BSA can be executed when signal conditions are poor (e.g., acceptable start I- and Q-values are not available), or the victim receiver baseband ICs have limited accuracy for BER or SNR as the feedback value. In such an implementation, the BSA can be executed to determine a start I-value and a start Q-value for the algorithms discussed above (i.e., the
FBA 2000, theBCA 2100, or theMSA 2200.FIG. 23 depicts anI-Q plane 2300 having 16 sub-regions with feedback values that are pseudorandom. The BSA can evaluate the feedback for multiple different I and Q pre-samples (e.g., from a lookup table) and select the pre-sample having the best feedback value. After the best pre-sample is determined, the BSA can transition to either theFBA 2000, theBCA 2100, or theMSA 2200 and use the I-value and Q-value for the pre-sample as a starting point for the algorithm. - There are several methods for implementing the BSA. In one method, the I and Q values associated with the best feedback value are selected from a number of samples (e.g., 4 or 16) with preset I and Q values. In the case of 10-bit I and Q values, four samples of feedback values may be taken from the following locations in the I and Q plane:
- I=(0xFF, 0x2FF, 0xFF, 0x2FF)
- Q=(0x2FF, 0x2FF, 0xFF, 0xFF)
- In the case of 10-bit I and Q values, sixteen samples of feedback values may be taken from the following locations in the I and Q plane:
- I=(0x80, 0x80, 0x80, 0x80, 0x180, 0x180, 0x180, 0x180, 0x280, 0x280, 0x280, 0x280, 0x380, 0x380, 0x380, 0x380)
- Q=0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380, 0x80, 0x180, 0x280, 0x380)
- The above locations are exemplary rather than limiting and many other locations are feasible without departing from the scope and spirit of the present invention.
- A second method for implementing the BSA includes obtaining feedback values at each of four (or other number) preset I and Q points. The maximum and minimum feedback values of the obtained feedback values can be identified. A feedback threshold is determined by either a) averaging the minimum and maximum feedback values, or b) adding a user selected offset value to the minimum feedback value. After determining the feedback threshold, the BSA can evaluate the feedback values for I and Q points proximal the best field out of the four I and Q points. For example, the BSA can use a user specified step size to explore I and Q points proximal the best of the four I and Q points. The BSA can terminate when one sample feedback meets or exceeds the feedback threshold. The BSA can then transition to the
MSA 2200. - The DSA uses an isosceles triangle approximation with two equal and opposite slopes for approximating a noise funnel curve.
FIG. 24 is agraph 2400 depicting a receivesignal quality indicator 2405 plotted versus I or Q values resulting from an implementation of a DSA, in accordance with certain exemplary embodiments. The DSA can select four points (X1-X4) along a noise funnel curve formed by the receivesignal quality indicator 2405 and compute a vertex, which is close to cancellation point C. The vertex can be computed using point-slope form of a linear equation. Once the vertex is found, the DSA can transition to theMSA 2200, using the vertex as starting I and Q values. -
FIG. 25 is a flow chart depicting aDSA 2500 for canceling noise and/or interference, in accordance with certain exemplary embodiments.FIG. 26 is agraph 2600 depicting acurve 2605 of receive signal quality indicator plotted versus either an I or Q axis (against I and Q axes ifFIG. 26 is plotted in three dimensions) resulting from an implementation of theDSA 2500 ofFIG. 25 , in accordance with certain exemplary embodiments. Theexemplary DSA 2500 uses an isosceles triangle approximation with two equal and opposite slopes for a noise funnel curve formed by the receiversignal quality indicator 2605. Referring toFIGS. 25 and 26 , inblock 2505, thecontroller 235 selects a number of samples of I-values and/or Q-values along the I or Q axis. For example, thecontroller 235 may select four samples. In certain exemplary embodiments, thecontroller 235 uses a BSA to select the location for the samples of I-values and/or Q-values. - In
block 2510, thecontroller 235 communicates the samples to the I/Q modulator 230 and the I/Q modulator applies each of the samples one at a time. Inblock 2520, thecontroller 235 obtains a feedback value, such as a “receive signal quality indicator,” for each of the applied samples and stores each feedback value and the corresponding sample I and Q values in thememory device 760. In certain exemplary embodiments, thecontroller 235 receives a “receive signal quality indicator” for each sample from thereceiver 135. - In
block 2520, thecontroller 235 compares the stored feedback values and identifies the better feedback value. For example, inFIG. 26 , thecontroller 235 identifies point X1 as resulting in the better feedback value. Let point X1 have an I and Q value of (I1, Q1) and a feedback value of Y1. - In
block 2525, with a preset step size, “STEP,” (e.g., STEP=most significant bit (MSB) of the I-value or Q-value or the MSB/2 or the MSB/4) thecontroller 235 selects another two points around point X1 by varying the I-value. For example, thecontroller 235 may select points X2 (e.g., I1+STEP, Q1) and X3 (e.g., I1−STEP, Q1). Thecontroller 235 communicates the samples X2 and X3 to the I/Q modulator 230 and the I/Q modulator 230 applies the settings for the samples X2 and X3 one at a time. For each sample, thecontroller 235 receives a feedback value, for example from thereceiver 135. Let the feedback value for X2 be Y+ and the feedback value for X3 be Y−. - In
block 2530, thecontroller 235 computes another sample point based on dual slope. In certain exemplary embodiments, thecontroller 235 computes another sample using SLOPE=(Y+−Y1)/STEP. This equation represents the slope of astraight line 2610 connecting points X2 and X1. Anotherstraight line 2615 is illustrated inFIG. 26 extending from point X3 and having a slope opposite theline 2610. Thelines point 2620. - In
block 2530, the controller computes the next I-value forpoint 2620 using: I2=I1−STEP*(Y+−Y−)/(Y+−Y1). Inblock 2535, the controller communicates I and Q values of (I2, Q1) to the I/Q modulator 230 and the I/Q modulator 230 applies the I and Q values. Inblock 2540, thecontroller 235 receives a feedback value for (I2, Q1) and stores the feedback value in thememory device 760. Let the feedback value for (I2, Q1) be Y2. - In block 2545, with the preset step size, “STEP,” the
controller 235 selects another two points around point X1 by varying the Q-value from point (I2, Q1). For example, thecontroller 235 may select points (I2, Q1+STEP) and (I2, Q1−STEP). Thecontroller 235 communicates the samples to the I/Q modulator 230 and the I/Q modulator 230 applies the settings for the samples one at a time. For each sample, thecontroller 235 receives a feedback value. Let the feedback value for (I2, Q1+STEP) be Y+ and the feedback value for (I2, Q1−STEP) be Y−. - In
block 2550, thecontroller 235 computes: - Q2=Q1−STEP*(Y+−Y−)/(Y+−Y2). In
block 2555, thecontroller 235 communicates I and Q values of (I2, Q2) to the I/Q modulator 230 and the I/Q modulator 230 applies the I and Q values. Inblock 2560, thecontroller 235 receives a feedback value for (I2, Q2) and stores the feedback value in thememory device 760. Let the feedback value for (I2, Q2) be Y3. - In
block 2565, thecontroller 235 reduces the size of STEP. In this exemplary embodiment, the size of STEP is halved. However, other (e.g., less conservative) reduction sizes are also feasible. Inblock 2570, thecontroller 235 conducts an inquiry to determine whether the size of STEP is less than a threshold, “STEPEND.” If the size of STEP is less than STEPEND, then theDSA 2500 proceeds to block 2580, where thecontroller 235 initiates an MSA (e.g., MSA 2200) using (I2, Q2) as a starting point. If the size of STEP is not less than STEPEND, then themethod 2500 proceeds to block 2575. Inblock 2575, thecontroller 235 assigns the I2, Q2, and Y2 values to I1, Q1, and Y1, respectively. Afterblock 2575, theDSA 2500 returns to block 2525. - The
exemplary DSA 2500 can be particularly useful when there are local preferred cancellation points with one global preferred cancellation point. The local preferred cancellation points refer to I and Q values where their feedback values are “locally” preferred. For example, an MSA, such asMSA 2200, would not jump outside the area proximal to the local preferred cancellation point. Implementing theDSA 2500 for upper bits, thecontroller 235 could avoid getting stuck with those local preferred cancellation points, while theMSA 2200 could finely tune to find the globally preferred cancellation point. -
FIG. 27 is a flow chart depicting aTSA 2700 for canceling noise and/or interference, in accordance with certain exemplary embodiments.FIG. 28 is agraph 2800 depicting cancellation points along anI-Q plane 2801 evaluated in an implementation of the TSA ofFIG. 27 , in accordance with certain exemplary embodiments. Referring toFIGS. 27 and 28 , inblock 2705, thecontroller 235 selects a number (e.g., 4) of samples in theI-Q plane 2801. In certain exemplary embodiments, thecontroller 235 uses the BSA to select the location in theI-Q plane 2801 for the samples. - In
block 2710, thecontroller 235 communicates the settings for the selected samples to the I/Q modulator 230 and the I/Q modulator 230 applies the settings for each sample one at a time. Inblock 2715, thecontroller 235 receives, for each sample, a feedback value (e.g., from the receiver 135) and stores the feedback value and its corresponding setting in thememory device 760. Inblock 2720, thecontroller 235 compares the feedback value and identifies the better or preferred feedback value. Let X1 inFIG. 28 be the sample resulting in the preferred feedback value. - In
block 2725, with a predetermined step size, “STEP,” (e.g., STEP=MSB/2 or MSB/4) thecontroller 235 selects another four samples proximal to X1. For example, thecontroller 235 may select (I1+STEP, Q1), (I1−STEP, Q1), (I1, Q1+STEP), and (I1, Q1−STEP). Thecontroller 235 communicates the four settings to the I/Q modulator 230 and the I/Q modulator 230 applies the settings for each sample one at a time. Thecontroller 235 receives a feedback value for each sample and stores the feedback value for each sample and the settings for each sample in thememory device 760. Thecontroller 235 compares the feedback values for the four samples and identifies the preferred feedback value. Let X2 inFIG. 28 be the sample resulting in the preferred feedback value. - In
block 2730, thecontroller 235 reduces the size of STEP. In this exemplary embodiment, the size of STEP is halved. Other size reductions are also feasible. Inblock 2735, thecontroller 235 conducts an inquiry to determine whether the size of STEP is less than a threshold, “STEPEND.” If the size of STEP is less than STEPEND, then theTSA 2700 proceeds to block 2755, where thecontroller 235 uses the setting (In+1, Qn+1) to control the I/Q modulator 230. If only one iteration of the TSA is performed, then thecontroller 235 uses the settings for the sample corresponding to the preferred feedback value inblock 2725 to control the I/Q modulator 230. If the size of STEP is not less than STEPEND, then theTSA 2700 proceeds to block 2740. - In
block 2740, thecontroller 235 selects another four samples proximal to the sample having the best stored feedback value. If it is the first iteration, the sample is X2 with (I2, Q2). This sample is designated as Xn asblock 2740 may be executed multiple times. For example, thecontroller 235 selects samples (In+STEP, Qn), (In−STEP, Qn), (In, Qn+STEP), (In, Qn−STEP). Thecontroller 235 communicates the four settings to the I/Q modulator 230 and the I/Q modulator 230 applies the settings for each sample one at a time. Thecontroller 235 receives a feedback value for each sample and stores the feedback value for each sample and the settings for each sample in thememory device 760. Inblock 2745, thecontroller 235 compares the feedback values for the four samples and identifies the preferred feedback value. Inblock 2750, thecontroller 235 reduces the size of STEP and the TSA returns to block 2735.FIG. 28 illustrates theTSA 2700 employing four iterations, represented by points X1-X4, that identifies a preferred cancellation point 2815 in theI-Q plane 2801. - The
exemplary TSA 2700 can be particularly useful when searching for an improved cancellation point and corresponding I/Q setting based on a previously preferred cancellation point, for example in response to a change in temperature. In such scenarios, theblock 2705 can be adapted to use the previous preferred I/Q setting rather than selecting four samples. TheTSA 2700 can narrow the field of search to the area in theI-Q plane 2801 near the previously preferred cancellation point. - Individual algorithms (e.g., BSA, FBA, BCA, MSA, DSA, and TSA) discussed above may be implemented as a standalone algorithm to decide acceptable I and Q values. Or, multiple ones of the algorithms can be employed together to increase the speed of the evaluation and attain a desired accuracy. For example, the BSA can be executed to determine either the first MSB or first MSB and second MSB of both I and Q values. Following the BSA, the FBA or BCA can be executed to determine the middle few bits of both I and Q values. Finally, the MSA can be executed to finely tune both I and Q values to achieve a better feedback value and thus, better noise or interference cancellation.
- Multiple iterations of the algorithms can be executed and/or the algorithms can be executed for longer periods of time to achieve better results. In certain exemplary embodiments, algorithms used for fine tuning (e.g., MSA and TSA) are employed in an always “on” mode where the
controller 150 continues to execute the algorithms while the noise canceller is in normal operation. This enables thecontroller 150 to adjust the settings of the noise canceller to account for environmental changes, such as changes in temperature or operating conditions. In addition, noise cancellers operating in parallel can each execute one or more of the algorithms simultaneously or sequentially. -
FIG. 29 is a flow chart depicting amethod 2900 for finding a preferred noise cancellation point for two noise cancellers disposed in a communication system, such as thecommunication system 100, in accordance with certain exemplary embodiments. For example, thecommunication system 100 may include, in alternative embodiments, twoHIPCF cancellers 130 in parallel. - In
block 2905, a control device, such as thecontroller 235 of one of twoHIPCF canceller 130, arranges the (I, Q) settings for the two cancellers in a sequence. For example, this sequence may be arranged as: (IninQnqn . . . I0i0Q0q0) with In . . . I0 and Qn . . . Q0 designating (I,Q) settings for a first canceller and in . . . i0 and qn . . . q0 designating IQ settings for a second canceller. The control device can then treat the two cancellers as a single canceller having the arranged sequence. - In
block 2910, the control device executes one or more of the cancellation algorithms discussed above (e.g., BSA, FBA, BCA, MSA, DSA, or TSA) using the sequence to determine a preferred cancellation setting for the cancellers. Inblock 2915, the control device stores the preferred cancellation settings in memory. -
FIG. 30 is a flow chart depicting analternative method 3000 for finding a preferred noise cancellation point for two noise cancellers disposed in a communication system, in accordance with certain exemplary embodiments. Inblock 3005, a control device, such as thecontroller 235 of one of the two noise cancellers, finds a preferred cancellation point for one of the two noise cancellers while the settings for the second of the two noise cancellers remain unchanged. One of the cancellation algorithms discussed above (e.g., BSA, FBA, BCA, MSA, DSA, or TSA) can be used to find the preferred noise cancellation point for the first noise canceller. - In
block 3010, the control device finds a preferred cancellation point for the second noise canceller using one or more of the cancellation algorithms (e.g., BSA, FBA, BCA, MSA, DSA, or TSA) while the settings for the first noise canceller remain unchanged at the preferred cancellation point found during execution ofblock 3005. Inblock 3015, with both cancellers operating using their respective preferred cancellation points, the control device obtains a feedback value resulting from the two noise cancellers. Inblock 3020, the control device compares the obtained feedback value to a preset threshold value. If the feedback value is better than the threshold or themethod 3000 has ran for more than a preset number of iterations, themethod 3000 proceeds to block 3025. Otherwise, the method returns to block 3005 with the current (I, Q) settings for both cancellers as starting values for the algorithm(s). Inblock 3025, the control device stores the settings for the two noise cancellers and controls the noise cancellers using the settings. -
FIG. 31 is a flow chart depicting analternative method 3100 for finding a preferred noise cancellation point for two noise cancellers disposed in a communication system, in accordance with certain exemplary embodiments. Thismethod 3100 addresses implantations where two noise cancellers are used to increase cancellation bandwidth. - In
block 3105, a control device, such as thecontroller 235 of one of the two noise cancellers, finds a preferred cancellation setting (e.g., (I, Q) settings) for the first of the two noise cancellers based on a feedback value for a lower portion of bandwidth while the second noise canceller is turned off. Inblock 3110, the control device stores the preferred noise cancellation setting for the first noise canceller. - In
block 3115, the control device finds a preferred cancellation setting (e.g., (I, Q) settings) for the second of the noise cancellers based on a feedback value for an upper portion of bandwidth while the first noise canceller is turned off. Inblock 3120, the control device stores the preferred noise cancellation setting for the second noise canceller. - In
block 3125, the control device turns both noise cancellers on and applies the respective preferred cancellation setting to each of the two noise cancellers. Inblock 3130, the control device executes an MSA for one step on the first noise canceller for the lower portion of the bandwidth. Inblock 3135, the control device executes an MSA for one step on the second noise canceller for the upper portion of the bandwidth. - In
block 3140, the control device obtains a feedback value for the noise cancellers and compares the feedback value to a preset value. If the feedback value is greater than the preset value or ifblocks method 3100 proceeds to block 3145. Otherwise, themethod 3100 returns to block 3130. Inblock 3145, the control device stores the final settings in memory and controls the noise cancellers using the final settings. Although themethods method methods methods method - In summary, a communication system in accordance with certain exemplary embodiments of the present invention can comprise a transmitter that communicates information at a first frequency, a receiver that receives communication signals at a second frequency that may be the same or near the first frequency, and an interference suppression device that cancels, corrects, addresses, or compensates for interference, EMI, noise, spurs, or other unwanted spectral components imposed onto the receiver by signals transmitted by the transmitter. The interference suppression device can be coupled to a transmit path of the transmitter (e.g., at the output of the transmitter's power amplifier) to obtain a sample of the transmitted signals. The interference compensation circuit can include a plurality of filters, such as band-pass filters, that block or suppress signals outside the frequency band of the receiver while passing noise or other interference signals within the frequency band of the receiver. The interference compensation circuit also can include an I/Q modulator that generates an interference compensation signal using the signal output by the filters. This interference compensation signal can have an amplitude the same as or close to the amplitude of the noise and a phase shift of 180 degrees relative to interference. These parameters are tuned in using a “receive signal quality indicator” feedback from the victim receiver. The interference compensation signal generated by the I/Q modulator is applied to a receive path of the receiver to cancel or suppress the interference imposed on the receiver by the transmitted signals.
- The communication systems described herein can be embodied in various communication devices, including cellular telephones, mobile computers, PDAs, personal navigation devices (e.g., GPS devices), or any other communication device comprising two or more communication elements. For example, the communication system can be embodied in a smartphone having a LTE/CDMA/GSM transceiver and a mobile TV tuner. Another example is a smartphone having a GSM/PCS/DCS/W-CDMA transceiver and a GPS receiver. Yet another example includes a notebook computer having a WLAN transceiver and a WiMAX or Bluetooth transceiver.
- In a mobile device embodiment, the two or more communication elements may communicate via two or more antennas with little spatial separation. Thus, signals transmitted by the two or more communication elements may impose interference on each other. To suppress or cancel this interference, a HIPCF canceller as described above can be employed in each communication direction. That is, a first HIPCF canceller can cancel or suppress interference imposed on a first of the two or more communication elements by a second of the two or more communication elements, while a second HIPCF canceller cancels or suppresses interference imposed on the second of the two or more communication elements by the first of the two or more communication elements. Certain components of both HIPCF cancellers can be fabricated on a single integrated circuit or on multiple integrated circuits, such as one or more CMOS circuits.
- Embodiments of the invention can be used with computer hardware and software that perform the methods and processing functions described above. As will be appreciated by those skilled in the art, the systems, methods, and procedures described herein can be embodied in a programmable computer, computer executable software, or digital circuitry. The software can be stored on computer readable media. For example, computer readable media can include a floppy disk, RAM, ROM, hard disk, removable media, flash memory, memory stick, optical media, magneto-optical media, CD-ROM, etc. Digital circuitry can include integrated circuits, gate arrays, building block logic, field programmable gate arrays (“FPGA”), etc.
- Although specific embodiments of the invention have been described above in detail, the description is merely for purposes of illustration. It should be appreciated, therefore, that many aspects of the invention were described above by way of example only and are not intended as required or essential elements of the invention unless explicitly stated otherwise. Various modifications of, and equivalent acts corresponding to, the disclosed aspects of the exemplary embodiments, in addition to those described above, can be made by a person of ordinary skill in the art, having the benefit of the present disclosure, without departing from the spirit and scope of the invention defined in the following claim(s), the scope of which is to be accorded the broadest interpretation so as to encompass such modifications and equivalent structures.
Claims (36)
1. A system for canceling interference on a first communication path associated with a transmission on a second communication path, comprising:
an input operable to couple to the second communication path to obtain a sample of the transmission;
an output operable to couple an interference compensation signal to the first communication path; and
a circuit disposed between the input and the output, comprising:
a plurality of filters arranged in a cascaded manner for receiving the sample and outputting a filtered signal; and
a modulator coupled to an output of the cascaded filters and operable to produce the interference compensation signal in response to adjusting at least one of amplitude, phase, and delay of the filtered signal.
2. The system of claim 1 , wherein the system is an integrated circuit.
3. The system of claim 1 , wherein the first communication path and the second communication path are disposed within a cellular telephone system comprising the system.
4. The system of claim 1 , wherein the each of the plurality of filters comprises a band-pass filter.
5. The system of claim 1 , wherein the plurality of filters substantially reduce amplitude of components of the sample outside of a frequency band associated with a receiver coupled to the first communication path.
6. The system of claim 1 , wherein the transmission is generated by a transmitter comprising a power amplifier and wherein the input is operable to couple to an output of the power amplifier.
7. The system of claim 1 , wherein the input comprises at least one of (a) a fixed capacitor and (b) a voltage-controlled variable capacitor.
8. The system of claim 1 , wherein the plurality of filters comprises a first band-pass filter that receives the sample from the input, the first band-pass filter comprising a high-Q inductor and at least one switchable capacitor.
9. The system of claim 8 , wherein the plurality of filters further comprises a second band-pass filter coupled to an output of the first band-pass filter, the second band-pass filter comprising a high-Q inductor and at least one switchable capacitor, and wherein the second band-pass filter couples to the output of the first band-pass filter via a low noise amplifier.
10. The system of claim 9 , wherein the plurality of filters further comprises a third band-pass filter coupled to an output of the second band-pass filter, the third band-pass filter comprising a high-Q inductor and at least one switchable capacitor, and wherein the third band-pass filter couples to the output of the second band-pass filter via a variable gain amplifier.
11. The system of claim 1 , wherein each of the plurality of filters comprise a band-pass filter, and wherein the circuit further comprises a controller logically coupled to each band-pass filter, the controller being operable to adjust a resonant frequency of each band-pass filter.
12. The system of claim 11 , wherein each band-pass filter comprises at least one switchable capacitor comprising a plurality of selectable capacitors, and wherein the controller adjusts the resonant frequency of one of the band-pass filters by selecting at least one of the selectable capacitors of the one band-pass filter.
13. The system of claim 11 , wherein the controller adjusts the resonant frequency of each band-pass filter in response to a change in frequency associated with a receiver coupled to the first communication path.
14. The system of claim 13 , wherein the controller adjusts the resonant frequency of each band-pass filter to match the frequency associated with the receiver.
15. The system of claim 1 , wherein the transmission is generated by a transmitter and the first communication path is coupled to a receiver.
16. The system of claim 15 , wherein the transmitter comprises a mobile telephone transmitter and the receiver comprises one of a mobile TV tuner and a GPS receiver.
17. The system of claim 1 , wherein the modulator comprises an I/Q modulator and at least a portion of the components of each of the plurality of band-pass filters and the I/Q modulator are fabricated on an integrated circuit.
18. A method for suppressing interference on a first communication path associated with a transmission on a second communication path, the method comprising:
obtaining a sample of the transmission;
filtering the sample by a plurality of cascaded filters;
producing an interference compensation signal by adjusting at least one of amplitude, phase, and delay of the filtered signal;
applying the interference compensation signal to the first communication path; and
suppressing interference on the first communication path in response to applying the interference compensation signal to the first communication path.
19. The method of claim 18 , wherein the interference compensation signal comprises an amplitude substantially the same as amplitude of the interference and a phase shifted approximately 180 degrees with respect to the interference.
20. The method of claim 18 , wherein the plurality of cascaded filters reduce amplitude of a portion of the sample having a frequency outside a frequency band for a receiver electrically connected to the first communication path.
21. The method of claim 18 , wherein the filtering comprises:
filtering the sample by a first band-pass filter having a first quality factor; and
further filtering the filtered sample by a second band-pass filter having a second quality factor; and
further filtering the filtered sample by a third band pass filter having a third quality factor.
22. The method of claim 18 , further comprising:
obtaining an indication of an amount of interference imposed on a victim receiver by the transmission; and
adjusting the interference compensation signal based on the amount of interference.
23. The method of claim 22 , wherein the indication comprises a receive signal quality indicator for the victim receiver.
24. The method of claim 18 , further comprising:
receiving a receive signal quality indicator for a victim receiver; and
adjusting one or more settings of one or more of the plurality of filters and one or more settings of an I/Q modulator that adjusts at least one of amplitude, phase, and delay of the filtered signal based on the receive quality indicator.
25. The method of claim 18 , further comprising:
detecting a change in frequency for a victim receiver electrically connected to the first communication path; and
adjusting a parameter of each of the cascaded filters based on the change in frequency.
26. The method of claim 25 , wherein each of the cascaded filters comprises a band-pass filter and wherein the parameter comprises a resonant frequency.
27. The method of claim 18 , further comprising adjusting at least one of (a) a resonant frequency, (b) a resonance gain, and (c) a quality factor of each of the plurality of filters in response to a change in one or more environmental conditions.
28. An interference compensation device for suppressing interfering signals introduced onto a receive path of a victim receiver by a transmission from a transmitter, comprising:
a first input for receiving an indication of interference imposed on the victim receiver;
a first band pass filter comprising a first quality factor for receiving and filtering a sample of the transmission;
a second band pass filter comprising a second quality factor and disposed along an output signal path of the first band pass filter for further filtering the sample; and
an I/Q modulator disposed along an output of the second band pass filter and operable to produce an interference compensation signal in response to adjusting at least one of amplitude, phase, and delay of the filtered sample based on the indication, the interference compensation signal operable to suppress the interfering signals.
29. The interference compensation device of claim 28 , wherein at least a portion of the first band pass filter, at least a portion of the second band pass filter, and the I/Q modulator are fabricated in an integrated circuit.
30. The interference compensation device of claim 28 , wherein the first band pass filter and the second band pass filter each comprise adjustable resonant frequencies and wherein the resonant frequencies are adjusted based on a frequency of the victim receiver.
31. The interference compensation device of claim 28 , further comprising a controller operable to adjust a resonant frequency of the first band pass filter and a resonant frequency of the second band pass filter.
32. The interference compensation device of claim 31 , wherein the controller is further operable to receive the indication and to adjust in-phase and quadrature settings of the I/Q modulator based on the indication.
33. The interference compensation device of claim 31 , wherein the controller is operable to control the first quality factor and the second quality factor based upon an out of band signal and blocker levels measured by a power detector connected to the signal path.
34. The interference compensation device of claim 31 , wherein the controller is operable to control the first quality factor and the second quality factor based upon a signal quality indicator feedback from the victim receiver.
35. The interference compensation device of claim 28 , further comprising a third band pass filter disposed along a signal path between the first band pass filter and the second band pass filter, the third band pass filter comprising a third quality factor.
36. The interference compensation device of claim 35 , further comprising at least one amplifier disposed along the output signal path of the first band pass filter, at least one variable gain amplifier disposed along the output signal path of the second band pass filter, and at least one buffer or amplifier disposed along the output signal path of the third band pass filter.
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Cited By (32)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20100048156A1 (en) * | 2008-08-21 | 2010-02-25 | Intersil Americas Inc. | Noise cancellation for antenna module |
US20110095812A1 (en) * | 2006-08-03 | 2011-04-28 | Broadcom Corporation | Transmission line coupled to circuits and q-enhancement cell |
US20110310936A1 (en) * | 2010-06-18 | 2011-12-22 | Teranetics, Inc. | Reducing Electromagnetic Interference in a Receive Signal with an Analog Correction Signal |
US20130176912A1 (en) * | 2012-01-10 | 2013-07-11 | Rf Micro Devices, Inc. | Rf duplexing device |
US20130196643A1 (en) * | 2011-12-15 | 2013-08-01 | Broadcom Corporation | Dynamic space, frequency and time domain coexistence |
US20130272312A1 (en) * | 2012-04-16 | 2013-10-17 | Broadcom Corporation | Method and apparatus for mixed-mode spectrum communication |
US8625704B1 (en) | 2008-09-25 | 2014-01-07 | Aquantia Corporation | Rejecting RF interference in communication systems |
US8724678B2 (en) | 2010-05-28 | 2014-05-13 | Aquantia Corporation | Electromagnetic interference reduction in wireline applications using differential signal compensation |
US20140253249A1 (en) * | 2013-03-11 | 2014-09-11 | Mediatek Singapore Pte. Ltd. | Circuit for providing a flat gain response over a selected frequency range and method of use |
US20140273898A1 (en) * | 2013-03-15 | 2014-09-18 | Dockon Ag | Frequency Selective Logarithmic Amplifier With Intrinsic Frequency Demodulation Capability |
US8861663B1 (en) | 2011-12-01 | 2014-10-14 | Aquantia Corporation | Correlated noise canceller for high-speed ethernet receivers |
US8891595B1 (en) | 2010-05-28 | 2014-11-18 | Aquantia Corp. | Electromagnetic interference reduction in wireline applications using differential signal compensation |
US8929468B1 (en) | 2012-06-14 | 2015-01-06 | Aquantia Corp. | Common-mode detection with magnetic bypass |
US8928425B1 (en) | 2008-09-25 | 2015-01-06 | Aquantia Corp. | Common mode detector for a communication system |
US20150044979A1 (en) * | 2013-08-09 | 2015-02-12 | Broadcom Corporation | Transmitter With Reduced Counter-Intermodulation |
US20150056940A1 (en) * | 2013-08-23 | 2015-02-26 | Qualcomm Incorporated | Harmonic trap for common gate amplifier |
US9118469B2 (en) | 2010-05-28 | 2015-08-25 | Aquantia Corp. | Reducing electromagnetic interference in a received signal |
US20150303981A1 (en) * | 2012-11-15 | 2015-10-22 | Telefonaktiebolaget L M Ericsson (Publ) | Transceiver front-end |
US9236892B2 (en) | 2013-03-15 | 2016-01-12 | Dockon Ag | Combination of steering antennas, CPL antenna(s), and one or more receive logarithmic detector amplifiers for SISO and MIMO applications |
US9263787B2 (en) | 2013-03-15 | 2016-02-16 | Dockon Ag | Power combiner and fixed/adjustable CPL antennas |
US9356643B2 (en) | 2011-12-29 | 2016-05-31 | Rf Micro Devices, Inc. | RF duplexing device |
US9374043B2 (en) | 2014-05-30 | 2016-06-21 | Qualcomm Incorporated | Dual stage carrier-aggregation (CA) low noise amplifier (LNA) having harmonic rejection and high linearity |
US9391717B2 (en) | 2012-11-13 | 2016-07-12 | Stephane Laurent-Michel | Method and system for signal dynamic range improvement for frequency-division duplex communication systems |
US20160218683A1 (en) * | 2013-09-12 | 2016-07-28 | Dockon Ag | Logarithmic detector amplifier system in open-loop configuration for use as high sensitivity selective receiver without frequency conversion |
US9503133B2 (en) | 2012-12-03 | 2016-11-22 | Dockon Ag | Low noise detection system using log detector amplifier |
US9548776B2 (en) | 2014-01-16 | 2017-01-17 | Qualcomm Incorporated | Interference cancelation using cooperative sensing |
US9577690B2 (en) | 2007-05-23 | 2017-02-21 | Hypres, Inc. | Wideband digital spectrometer |
US9590572B2 (en) | 2013-09-12 | 2017-03-07 | Dockon Ag | Logarithmic detector amplifier system for use as high sensitivity selective receiver without frequency conversion |
US20170187337A1 (en) * | 2013-09-12 | 2017-06-29 | Dockon Ag | Advanced amplifier system for ultra-wide band rf communication |
US9755691B2 (en) | 2012-11-14 | 2017-09-05 | Andrew Joo Kim | Method and system for mitigating the effects of a transmitted blocker and distortions therefrom in a radio receiver |
US10263568B2 (en) * | 2017-02-23 | 2019-04-16 | Avaga Technologies International Sales PTE. Limited | Radio frequency feedback power amplifiers |
US11057126B2 (en) * | 2018-12-19 | 2021-07-06 | Rohde & Schwarz Gmbh & Co. Kg | Test system and method of testing follower jammer robustness of a radio |
Families Citing this family (69)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US10284356B2 (en) | 2011-02-03 | 2019-05-07 | The Board Of Trustees Of The Leland Stanford Junior University | Self-interference cancellation |
US10230419B2 (en) | 2011-02-03 | 2019-03-12 | The Board Of Trustees Of The Leland Stanford Junior University | Adaptive techniques for full duplex communications |
US9331737B2 (en) | 2012-02-08 | 2016-05-03 | The Board Of Trustees Of The Leland Stanford Junior University | Systems and methods for cancelling interference using multiple attenuation delays |
US8532566B2 (en) * | 2011-06-08 | 2013-09-10 | Andrew Llc | System and method for reducing desensitization of a base station transceiver for mobile wireless repeater systems |
GB201113130D0 (en) * | 2011-07-29 | 2011-09-14 | Bae Systems Plc | Radio frequency communication |
US10243719B2 (en) | 2011-11-09 | 2019-03-26 | The Board Of Trustees Of The Leland Stanford Junior University | Self-interference cancellation for MIMO radios |
US9325432B2 (en) | 2012-02-08 | 2016-04-26 | The Board Of Trustees Of The Leland Stanford Junior University | Systems and methods for full-duplex signal shaping |
US9113481B2 (en) | 2012-09-27 | 2015-08-18 | Qualcomm Incorporated | Adaptive non-linear interference cancellation using side-band information |
US20140119244A1 (en) * | 2012-11-01 | 2014-05-01 | Research In Motion Limited | Cognitive radio rf front end |
US20140170991A1 (en) * | 2012-12-18 | 2014-06-19 | Qualcomm Incorporated | Agile active interference cancellation (aaic) for multi-radio mobile devices |
US20140271270A1 (en) * | 2013-03-12 | 2014-09-18 | Geotek Energy, Llc | Magnetically coupled expander pump with axial flow path |
US11163050B2 (en) | 2013-08-09 | 2021-11-02 | The Board Of Trustees Of The Leland Stanford Junior University | Backscatter estimation using progressive self interference cancellation |
US9036749B2 (en) | 2013-08-09 | 2015-05-19 | Kumu Networks, Inc. | Systems and methods for frequency independent analog self-interference cancellation |
US8976641B2 (en) | 2013-08-09 | 2015-03-10 | Kumu Networks, Inc. | Systems and methods for non-linear digital self-interference cancellation |
US9698860B2 (en) | 2013-08-09 | 2017-07-04 | Kumu Networks, Inc. | Systems and methods for self-interference canceller tuning |
US9054795B2 (en) | 2013-08-14 | 2015-06-09 | Kumu Networks, Inc. | Systems and methods for phase noise mitigation |
CN105493416A (en) | 2013-08-29 | 2016-04-13 | 库姆网络公司 | Full-duplex relays |
US10673519B2 (en) | 2013-08-29 | 2020-06-02 | Kuma Networks, Inc. | Optically enhanced self-interference cancellation |
US9520983B2 (en) | 2013-09-11 | 2016-12-13 | Kumu Networks, Inc. | Systems for delay-matched analog self-interference cancellation |
US10230422B2 (en) | 2013-12-12 | 2019-03-12 | Kumu Networks, Inc. | Systems and methods for modified frequency-isolation self-interference cancellation |
WO2015089460A1 (en) | 2013-12-12 | 2015-06-18 | Kumu Networks, Inc. | Systems and methods for hybrid self-interference cancellation |
US9774405B2 (en) | 2013-12-12 | 2017-09-26 | Kumu Networks, Inc. | Systems and methods for frequency-isolated self-interference cancellation |
US9712312B2 (en) | 2014-03-26 | 2017-07-18 | Kumu Networks, Inc. | Systems and methods for near band interference cancellation |
US10128879B2 (en) * | 2014-03-31 | 2018-11-13 | Intel IP Corporation | Enhanced receive sensitivity for concurrent communications |
US11209536B2 (en) | 2014-05-02 | 2021-12-28 | The Board Of Trustees Of The Leland Stanford Junior University | Method and apparatus for tracking motion using radio frequency signals |
WO2015179874A1 (en) | 2014-05-23 | 2015-11-26 | Kumu Networks, Inc. | Systems and methods for multi-rate digital self-interference cancellation |
US9184974B1 (en) * | 2014-06-26 | 2015-11-10 | The Boeing Company | In-phase and quadrature radio frequency digital-to-analog converter |
US9521023B2 (en) | 2014-10-17 | 2016-12-13 | Kumu Networks, Inc. | Systems for analog phase shifting |
US9712313B2 (en) | 2014-11-03 | 2017-07-18 | Kumu Networks, Inc. | Systems for multi-peak-filter-based analog self-interference cancellation |
CN104617965B (en) * | 2014-12-25 | 2017-06-27 | 大唐半导体设计有限公司 | Improve the method and device of receiver sensitivity |
US9590673B2 (en) * | 2015-01-20 | 2017-03-07 | Qualcomm Incorporated | Switched, simultaneous and cascaded interference cancellation |
US9673854B2 (en) | 2015-01-29 | 2017-06-06 | Kumu Networks, Inc. | Method for pilot signal based self-inteference cancellation tuning |
CN104678407B (en) * | 2015-02-16 | 2017-06-20 | 络达科技股份有限公司 | It is used to reduce the device and method that harmonic wave interference gps signal is received |
US9800287B2 (en) * | 2015-05-22 | 2017-10-24 | Qualcomm Incorporated | Pilot-based analog active interference canceller |
US9866336B2 (en) * | 2015-06-17 | 2018-01-09 | Google Llc | Phased array antenna self-calibration |
US20170019240A1 (en) | 2015-07-16 | 2017-01-19 | LGS Innovations LLC | Tone based in-phase and quadrature-phase (iq) compensation |
US9634823B1 (en) | 2015-10-13 | 2017-04-25 | Kumu Networks, Inc. | Systems for integrated self-interference cancellation |
JP6770083B2 (en) | 2015-11-23 | 2020-10-14 | アンロテック リミテッド | Variable filter |
US10666305B2 (en) | 2015-12-16 | 2020-05-26 | Kumu Networks, Inc. | Systems and methods for linearized-mixer out-of-band interference mitigation |
US9819325B2 (en) | 2015-12-16 | 2017-11-14 | Kumu Networks, Inc. | Time delay filters |
US9800275B2 (en) | 2015-12-16 | 2017-10-24 | Kumu Networks, Inc. | Systems and methods for out-of band-interference mitigation |
US9742593B2 (en) | 2015-12-16 | 2017-08-22 | Kumu Networks, Inc. | Systems and methods for adaptively-tuned digital self-interference cancellation |
US20170188264A1 (en) * | 2015-12-23 | 2017-06-29 | Qualcomm Incorporated | Analog interference cancellation using digital computation of cancellation coefficients |
US9979374B2 (en) | 2016-04-25 | 2018-05-22 | Kumu Networks, Inc. | Integrated delay modules |
US10454444B2 (en) | 2016-04-25 | 2019-10-22 | Kumu Networks, Inc. | Integrated delay modules |
US10338205B2 (en) | 2016-08-12 | 2019-07-02 | The Board Of Trustees Of The Leland Stanford Junior University | Backscatter communication among commodity WiFi radios |
US10230413B2 (en) * | 2016-08-29 | 2019-03-12 | Skyworks Solutions, Inc. | Filtering architectures and methods for wireless applications |
EP3532981A4 (en) | 2016-10-25 | 2020-06-24 | The Board of Trustees of the Leland Stanford Junior University | Backscattering ambient ism band signals |
JP6571064B2 (en) * | 2016-11-21 | 2019-09-04 | 株式会社東芝 | Detection device and sensor device |
CN106849877A (en) * | 2016-12-20 | 2017-06-13 | 中国电子科技集团公司第四十三研究所 | One kind miniaturization low-noise amplifier |
CN108322237B (en) * | 2017-01-14 | 2020-09-29 | 鸿富锦精密工业(深圳)有限公司 | Interference suppression system and method |
US10361736B2 (en) * | 2017-03-24 | 2019-07-23 | Lg Electronics Inc. | Method for transmitting and receiving signal by aggregating two uplink carriers |
WO2018183384A1 (en) | 2017-03-27 | 2018-10-04 | Kumu Networks, Inc. | Systems and methods for intelligently-tunded digital self-interference cancellation |
KR102234970B1 (en) | 2017-03-27 | 2021-04-02 | 쿠무 네트웍스, 아이엔씨. | System and method for mitigating interference outside of tunable band |
EP3602776B1 (en) | 2017-03-27 | 2021-04-21 | Kumu Networks, Inc. | Enhanced linearity mixer |
TWI776901B (en) | 2017-05-24 | 2022-09-11 | 英商安諾特克有限公司 | Apparatus and method for controlling a resonator |
US10200076B1 (en) | 2017-08-01 | 2019-02-05 | Kumu Networks, Inc. | Analog self-interference cancellation systems for CMTS |
WO2019169047A1 (en) | 2018-02-27 | 2019-09-06 | Kumu Networks, Inc. | Systems and methods for configurable hybrid self-interference cancellation |
CN109164700B (en) * | 2018-08-07 | 2021-06-18 | 中国科学院光电技术研究所 | Control method for inhibiting full-frequency disturbance |
FR3091432B1 (en) * | 2018-12-28 | 2022-03-18 | Thales Sa | PROCEDURE FOR VERIFYING THE OPERATION OF A RADIO CHANNEL |
US10673606B1 (en) * | 2019-01-22 | 2020-06-02 | Realtek Semiconductor Corp. | High-speed full-duplex transceiver and method thereof |
US10868661B2 (en) | 2019-03-14 | 2020-12-15 | Kumu Networks, Inc. | Systems and methods for efficiently-transformed digital self-interference cancellation |
US11277110B2 (en) | 2019-09-03 | 2022-03-15 | Anlotek Limited | Fast frequency switching in a resonant high-Q analog filter |
EP4070171A1 (en) | 2019-12-05 | 2022-10-12 | Anlotek Limited | Use of stable tunable active feedback analog filters in frequency synthesis |
CN111030079B (en) * | 2020-03-06 | 2020-07-10 | 锐石创芯(深圳)科技有限公司 | Power supply network capable of switching loop gain and signal processing system |
US11876499B2 (en) | 2020-06-15 | 2024-01-16 | Anlotek Limited | Tunable bandpass filter with high stability and orthogonal tuning |
CN114070420B (en) * | 2020-07-31 | 2023-04-07 | 华为技术有限公司 | Anti-interference electronic equipment and anti-interference method |
US11955942B2 (en) | 2021-02-27 | 2024-04-09 | Anlotek Limited | Active multi-pole filter |
CN115276681B (en) * | 2022-07-27 | 2023-11-24 | 东集技术股份有限公司 | RFID reader-writer system, transmitting power closed-loop control method and main controller |
Citations (15)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5737035A (en) * | 1995-04-21 | 1998-04-07 | Microtune, Inc. | Highly integrated television tuner on a single microcircuit |
US6131013A (en) * | 1998-01-30 | 2000-10-10 | Motorola, Inc. | Method and apparatus for performing targeted interference suppression |
US6441843B1 (en) * | 1998-08-24 | 2002-08-27 | Samsung Electronics Co., Ltd. | Suppression of effects of co-channel NTSC interference artifacts upon digital TV receiver adaptive equalizer |
US20050153677A1 (en) * | 1998-11-12 | 2005-07-14 | Broadcom Corporation | System and method for on-chip filter tuning |
US6968171B2 (en) * | 2002-06-04 | 2005-11-22 | Sierra Wireless, Inc. | Adaptive noise reduction system for a wireless receiver |
US20050270094A1 (en) * | 2004-06-04 | 2005-12-08 | Toshifumi Nakatani | Multistage amplifying devices, and reception device and transmission device using the same |
US20060040628A1 (en) * | 2004-08-20 | 2006-02-23 | Alain-Serge Porret | Television receiver including an integrated band selection filter |
US20070218850A1 (en) * | 2004-10-08 | 2007-09-20 | Jianping Pan | Integrated Tuner for Terrestrial and Cable Television |
US7336745B2 (en) * | 2002-09-03 | 2008-02-26 | Honeywell International Inc. | Methods and apparatus to provide communication protection technology for satellite earthstations |
US20080242245A1 (en) * | 2007-03-27 | 2008-10-02 | Vladimir Aparin | Rejection of transmit signal leakage in wireless communication device |
US7538621B2 (en) * | 1997-08-01 | 2009-05-26 | Microtune (Texas), L.P. | Broadband integrated tuner |
US7619282B2 (en) * | 1997-01-18 | 2009-11-17 | Semiconductor Energy Laboratory Co., Ltd. | Hybrid circuit and electronic device using same |
US20100159837A1 (en) * | 2008-12-19 | 2010-06-24 | Paul Wilkinson Dent | Own Transmitter Interference Tolerant Transceiver and Receiving Methods |
US20110294443A1 (en) * | 2010-05-26 | 2011-12-01 | George Nohra | High isolation switch with notch filter |
US20120147928A1 (en) * | 2009-11-17 | 2012-06-14 | Hisao Nakano | Radio communication apparatus |
Family Cites Families (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
CA2022050C (en) * | 1989-07-27 | 1993-03-23 | Toru Matsuura | Cross-polarization interference cancellation system capable of stably carrying out operation |
US6915112B1 (en) * | 2000-11-12 | 2005-07-05 | Intel Corporation | Active cancellation tuning to reduce a wireless coupled transmit signal |
WO2004008647A2 (en) * | 2002-07-16 | 2004-01-22 | In Kwan Hwang | Multistage adaptive parallel interference canceller |
US7616700B2 (en) * | 2003-12-22 | 2009-11-10 | Quellan, Inc. | Method and system for slicing a communication signal |
US8098779B2 (en) * | 2006-08-08 | 2012-01-17 | Qualcomm Incorporated | Interference detection and mitigation |
WO2009156510A2 (en) * | 2008-06-27 | 2009-12-30 | Telefonaktiebolaget L M Ericsson (Publ) | Own transmitter interference tolerant transceiver and receiving methods |
-
2011
- 2011-01-26 US US13/014,657 patent/US20110212692A1/en not_active Abandoned
- 2011-01-26 US US13/014,681 patent/US8724731B2/en active Active
- 2011-02-23 CN CN2011100517141A patent/CN102195660A/en active Pending
- 2011-02-23 TW TW100105897A patent/TW201201530A/en unknown
-
2014
- 2014-03-25 US US14/224,595 patent/US9231712B2/en active Active
Patent Citations (17)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5737035A (en) * | 1995-04-21 | 1998-04-07 | Microtune, Inc. | Highly integrated television tuner on a single microcircuit |
US7619282B2 (en) * | 1997-01-18 | 2009-11-17 | Semiconductor Energy Laboratory Co., Ltd. | Hybrid circuit and electronic device using same |
US7538621B2 (en) * | 1997-08-01 | 2009-05-26 | Microtune (Texas), L.P. | Broadband integrated tuner |
US6131013A (en) * | 1998-01-30 | 2000-10-10 | Motorola, Inc. | Method and apparatus for performing targeted interference suppression |
US6441843B1 (en) * | 1998-08-24 | 2002-08-27 | Samsung Electronics Co., Ltd. | Suppression of effects of co-channel NTSC interference artifacts upon digital TV receiver adaptive equalizer |
US7423699B2 (en) * | 1998-11-12 | 2008-09-09 | Broadcom Corporation | Fully integrated tuner architecture |
US20050153677A1 (en) * | 1998-11-12 | 2005-07-14 | Broadcom Corporation | System and method for on-chip filter tuning |
US7515895B2 (en) * | 1998-11-12 | 2009-04-07 | Broadcom Corporation | System and method for on-chip filter tuning |
US6968171B2 (en) * | 2002-06-04 | 2005-11-22 | Sierra Wireless, Inc. | Adaptive noise reduction system for a wireless receiver |
US7336745B2 (en) * | 2002-09-03 | 2008-02-26 | Honeywell International Inc. | Methods and apparatus to provide communication protection technology for satellite earthstations |
US20050270094A1 (en) * | 2004-06-04 | 2005-12-08 | Toshifumi Nakatani | Multistage amplifying devices, and reception device and transmission device using the same |
US20060040628A1 (en) * | 2004-08-20 | 2006-02-23 | Alain-Serge Porret | Television receiver including an integrated band selection filter |
US20070218850A1 (en) * | 2004-10-08 | 2007-09-20 | Jianping Pan | Integrated Tuner for Terrestrial and Cable Television |
US20080242245A1 (en) * | 2007-03-27 | 2008-10-02 | Vladimir Aparin | Rejection of transmit signal leakage in wireless communication device |
US20100159837A1 (en) * | 2008-12-19 | 2010-06-24 | Paul Wilkinson Dent | Own Transmitter Interference Tolerant Transceiver and Receiving Methods |
US20120147928A1 (en) * | 2009-11-17 | 2012-06-14 | Hisao Nakano | Radio communication apparatus |
US20110294443A1 (en) * | 2010-05-26 | 2011-12-01 | George Nohra | High isolation switch with notch filter |
Cited By (69)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20110095812A1 (en) * | 2006-08-03 | 2011-04-28 | Broadcom Corporation | Transmission line coupled to circuits and q-enhancement cell |
US8190115B2 (en) * | 2006-08-03 | 2012-05-29 | Broadcom Corporation | Transmission line coupled to circuits and Q-enhancement cell |
US9906248B2 (en) | 2007-05-23 | 2018-02-27 | Hypres, Inc. | Wideband digital spectrometer |
US9577690B2 (en) | 2007-05-23 | 2017-02-21 | Hypres, Inc. | Wideband digital spectrometer |
US20100048156A1 (en) * | 2008-08-21 | 2010-02-25 | Intersil Americas Inc. | Noise cancellation for antenna module |
US8422974B2 (en) * | 2008-08-21 | 2013-04-16 | Intersil Americas Inc. | Noise cancellation for antenna module |
US9912375B1 (en) | 2008-09-25 | 2018-03-06 | Aquantia Corp. | Cancellation of alien interference in communication systems |
US9590695B1 (en) | 2008-09-25 | 2017-03-07 | Aquantia Corp. | Rejecting RF interference in communication systems |
US8625704B1 (en) | 2008-09-25 | 2014-01-07 | Aquantia Corporation | Rejecting RF interference in communication systems |
US8928425B1 (en) | 2008-09-25 | 2015-01-06 | Aquantia Corp. | Common mode detector for a communication system |
US9118469B2 (en) | 2010-05-28 | 2015-08-25 | Aquantia Corp. | Reducing electromagnetic interference in a received signal |
US8724678B2 (en) | 2010-05-28 | 2014-05-13 | Aquantia Corporation | Electromagnetic interference reduction in wireline applications using differential signal compensation |
US8891595B1 (en) | 2010-05-28 | 2014-11-18 | Aquantia Corp. | Electromagnetic interference reduction in wireline applications using differential signal compensation |
US8792597B2 (en) * | 2010-06-18 | 2014-07-29 | Aquantia Corporation | Reducing electromagnetic interference in a receive signal with an analog correction signal |
US20110310936A1 (en) * | 2010-06-18 | 2011-12-22 | Teranetics, Inc. | Reducing Electromagnetic Interference in a Receive Signal with an Analog Correction Signal |
US8861663B1 (en) | 2011-12-01 | 2014-10-14 | Aquantia Corporation | Correlated noise canceller for high-speed ethernet receivers |
US20130196643A1 (en) * | 2011-12-15 | 2013-08-01 | Broadcom Corporation | Dynamic space, frequency and time domain coexistence |
US10880884B2 (en) | 2011-12-15 | 2020-12-29 | Avago Technologies International Sales Pte. Limited | Dynamic space, frequency and time domain coexistence |
US9826524B2 (en) * | 2011-12-15 | 2017-11-21 | Avago Technologies General Ip (Singapore) Pte. Ltd. | Dynamic space, frequency and time domain coexistence |
US9667304B2 (en) | 2011-12-29 | 2017-05-30 | Qorvo Us, Inc. | RF duplexing device |
US9356643B2 (en) | 2011-12-29 | 2016-05-31 | Rf Micro Devices, Inc. | RF duplexing device |
US20130176912A1 (en) * | 2012-01-10 | 2013-07-11 | Rf Micro Devices, Inc. | Rf duplexing device |
US9319208B2 (en) * | 2012-01-10 | 2016-04-19 | Rf Micro Devices, Inc. | RF duplexing device |
US9083390B2 (en) * | 2012-04-16 | 2015-07-14 | Broadcom Corporation | Method and apparatus for mixed-mode spectrum communication |
US20130272312A1 (en) * | 2012-04-16 | 2013-10-17 | Broadcom Corporation | Method and apparatus for mixed-mode spectrum communication |
US8929468B1 (en) | 2012-06-14 | 2015-01-06 | Aquantia Corp. | Common-mode detection with magnetic bypass |
US9391717B2 (en) | 2012-11-13 | 2016-07-12 | Stephane Laurent-Michel | Method and system for signal dynamic range improvement for frequency-division duplex communication systems |
US9960804B2 (en) | 2012-11-14 | 2018-05-01 | Spectra7 Microsystems Ltd | Method and system for mitigating the effects of a transmitted blocker and distortions therefrom in a radio receiver |
US9755691B2 (en) | 2012-11-14 | 2017-09-05 | Andrew Joo Kim | Method and system for mitigating the effects of a transmitted blocker and distortions therefrom in a radio receiver |
US10084506B2 (en) * | 2012-11-15 | 2018-09-25 | Telefonaktiebolaget Lm Ericsson (Publ) | Transceiver front-end |
US20150303981A1 (en) * | 2012-11-15 | 2015-10-22 | Telefonaktiebolaget L M Ericsson (Publ) | Transceiver front-end |
US9503133B2 (en) | 2012-12-03 | 2016-11-22 | Dockon Ag | Low noise detection system using log detector amplifier |
US9621203B2 (en) | 2012-12-03 | 2017-04-11 | Dockon Ag | Medium communication system using log detector amplifier |
US20140253249A1 (en) * | 2013-03-11 | 2014-09-11 | Mediatek Singapore Pte. Ltd. | Circuit for providing a flat gain response over a selected frequency range and method of use |
CN104052408A (en) * | 2013-03-11 | 2014-09-17 | 联发科技(新加坡)私人有限公司 | Integrated circuit |
US9166545B2 (en) * | 2013-03-11 | 2015-10-20 | Mediatek Singapore Ptd. Ltd. | Circuit for providing a flat gain response over a selected frequency range and method of use |
KR20200142102A (en) * | 2013-03-15 | 2020-12-21 | 도콘 아게 | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
US11012953B2 (en) * | 2013-03-15 | 2021-05-18 | Dockon Ag | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
US9236892B2 (en) | 2013-03-15 | 2016-01-12 | Dockon Ag | Combination of steering antennas, CPL antenna(s), and one or more receive logarithmic detector amplifiers for SISO and MIMO applications |
KR102332682B1 (en) | 2013-03-15 | 2021-12-02 | 도콘 아게 | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
EP2974000A1 (en) * | 2013-03-15 | 2016-01-20 | Dockon AG | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
EP2974000A4 (en) * | 2013-03-15 | 2017-04-05 | Dockon AG | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
KR102268740B1 (en) | 2013-03-15 | 2021-06-24 | 도콘 아게 | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
US9397382B2 (en) | 2013-03-15 | 2016-07-19 | Dockon Ag | Logarithmic amplifier with universal demodulation capabilities |
US9684807B2 (en) * | 2013-03-15 | 2017-06-20 | Dockon Ag | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
KR20200095582A (en) * | 2013-03-15 | 2020-08-10 | 도콘 아게 | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
US20170222603A1 (en) * | 2013-03-15 | 2017-08-03 | Dockon Ag | Frequency Selective Logarithmic Amplifier With Intrinsic Frequency Demodulation Capability |
KR20160005690A (en) * | 2013-03-15 | 2016-01-15 | 도콘 아게 | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
US9263787B2 (en) | 2013-03-15 | 2016-02-16 | Dockon Ag | Power combiner and fixed/adjustable CPL antennas |
US9356561B2 (en) | 2013-03-15 | 2016-05-31 | Dockon Ag | Logarithmic amplifier with universal demodulation capabilities |
US20140273898A1 (en) * | 2013-03-15 | 2014-09-18 | Dockon Ag | Frequency Selective Logarithmic Amplifier With Intrinsic Frequency Demodulation Capability |
KR102226415B1 (en) | 2013-03-15 | 2021-03-11 | 도콘 아게 | Frequency selective logarithmic amplifier with intrinsic frequency demodulation capability |
US20150044979A1 (en) * | 2013-08-09 | 2015-02-12 | Broadcom Corporation | Transmitter With Reduced Counter-Intermodulation |
US9425835B2 (en) * | 2013-08-09 | 2016-08-23 | Broadcom Corporation | Transmitter with reduced counter-intermodulation |
US20150056940A1 (en) * | 2013-08-23 | 2015-02-26 | Qualcomm Incorporated | Harmonic trap for common gate amplifier |
US20170187337A1 (en) * | 2013-09-12 | 2017-06-29 | Dockon Ag | Advanced amplifier system for ultra-wide band rf communication |
US20160218683A1 (en) * | 2013-09-12 | 2016-07-28 | Dockon Ag | Logarithmic detector amplifier system in open-loop configuration for use as high sensitivity selective receiver without frequency conversion |
US9590572B2 (en) | 2013-09-12 | 2017-03-07 | Dockon Ag | Logarithmic detector amplifier system for use as high sensitivity selective receiver without frequency conversion |
US20180205350A1 (en) * | 2013-09-12 | 2018-07-19 | Dockon Ag | Amplifier System for Use as High Sensitivity Selective Receiver Without Frequency Conversion |
US20180205351A1 (en) * | 2013-09-12 | 2018-07-19 | Dockon Ag | Amplifier System for Use as High Sensitivity Selective Receiver Without Frequency Conversion |
US11183974B2 (en) * | 2013-09-12 | 2021-11-23 | Dockon Ag | Logarithmic detector amplifier system in open-loop configuration for use as high sensitivity selective receiver without frequency conversion |
US11095255B2 (en) * | 2013-09-12 | 2021-08-17 | Dockon Ag | Amplifier system for use as high sensitivity selective receiver without frequency conversion |
US10333475B2 (en) * | 2013-09-12 | 2019-06-25 | QuantalRF AG | Logarithmic detector amplifier system for use as high sensitivity selective receiver without frequency conversion |
US11050393B2 (en) * | 2013-09-12 | 2021-06-29 | Dockon Ag | Amplifier system for use as high sensitivity selective receiver without frequency conversion |
US11082014B2 (en) * | 2013-09-12 | 2021-08-03 | Dockon Ag | Advanced amplifier system for ultra-wide band RF communication |
US9548776B2 (en) | 2014-01-16 | 2017-01-17 | Qualcomm Incorporated | Interference cancelation using cooperative sensing |
US9374043B2 (en) | 2014-05-30 | 2016-06-21 | Qualcomm Incorporated | Dual stage carrier-aggregation (CA) low noise amplifier (LNA) having harmonic rejection and high linearity |
US10263568B2 (en) * | 2017-02-23 | 2019-04-16 | Avaga Technologies International Sales PTE. Limited | Radio frequency feedback power amplifiers |
US11057126B2 (en) * | 2018-12-19 | 2021-07-06 | Rohde & Schwarz Gmbh & Co. Kg | Test system and method of testing follower jammer robustness of a radio |
Also Published As
Publication number | Publication date |
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US20110211649A1 (en) | 2011-09-01 |
TW201201530A (en) | 2012-01-01 |
CN102195660A (en) | 2011-09-21 |
US9231712B2 (en) | 2016-01-05 |
US20140206300A1 (en) | 2014-07-24 |
US8724731B2 (en) | 2014-05-13 |
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