WO1998032490A1

WO1998032490A1 - Current and temperature compensated voltage reference

Info

Publication number: WO1998032490A1
Application number: PCT/US1998/000954
Authority: WO
Inventors: Terrence L. Marshall; Suneel Arora; Michael W. Dooley
Original assignee: Cardiac Pacemakers, Inc.
Priority date: 1997-01-29
Filing date: 1998-01-16
Publication date: 1998-07-30
Also published as: CA2278498A1; EP1011801A1

Abstract

A current and temperature compensated voltage reference circuit operates from a power supply voltage as low as approximately 1.3 Volts. A current reference circuit uses a folded-cascode amplifier to measure the temperature dependent voltages provided by first and second bias circuits, each of which includes two series-coupled diodes or diode-connected bipolar junction transistors (BJTs), and provides a proportional to absolute temperature (PTAT) current in response thereto. A voltage reference circuit uses a PTAT current through a resistor to create a PTAT voltage in series with a diode-voltage, and the resistor value is adjusted until the sum of these voltages is approximately equal to the bandgap voltage of silicon, thereby providing a temperature compensated voltage reference. The reference circuit is suitable for use with an implantable cardiac rythm management system having a battery that provides a power supply voltage approximately between 1.3 Volts and 3.25 Volts.

Description

CURRENT AND TEMPERATURE COMPENSATED VOLTAGE REFERENCE

Field of the Invention

This invention relates generally to precision voltage and current references.

Background Many integrated circuit functions require precise voltage and current references. For example, an analog-to-digital converter typically requires a precise voltage reference to establish and quantize an analog input voltage range. In another example, many analog filters, such as transconductance-capacitance (g_mC) filters, have filter gain and rolloff frequency characteristics that depend upon their bias currents. A precise current reference is useful for generating accurate bias currents in such filters and other circuits.

Many battery powered electronics applications, including implantable cardiac rhythm management systems, need current and temperature compensated voltage reference circuits that operate at low power supply voltages and power consumption. There is a critical need for circuits, including reference circuits, that operate at low power supply voltages and draw less current from the low voltage power supply in order to increase battery longevity. There is also a need for such reference circuits in which the value of the reference voltage and reference current are capable of fine adjustment. There is a further need for such reference circuits in which the resulting reference voltage is temperature compensated, i.e. the sensitivity of the reference voltage to temperature variations is reduced.

Summary The present invention provides a system that includes current and temperature compensated voltage reference that is capable of operating from a power supply voltage as low as approximately 1.3 Volts. For battery powered electronics applications, operation at lower power supply voltages offers significant advantages; it may eliminate the need for electrically coupling more than one battery in series in order to obtain a higher power supply voltage. Also, in some batteries, an internal battery impedance increases over the course of the battery life, thereby reducing the voltage available at the battery terminals. Thus, the useful life of the battery may be extended by using a reference circuit that is capable of operating from this reduced battery terminal voltage near the end of the battery's life. This is particularly true for implantable medical devices, including cardiac rhythm management systems, such as pacemakers and defibrillators, in which circuits that are capable of operating at reduced power supply voltages can extend battery life for several years, thereby avoiding surgical explantation and replacement of the implanted device. In elderly patients, such traumatic surgical intervention may be both uncomfortable and risky.

The system, by operating from a power supply voltage as low as approximately 1.3 Volts, may significantly increase the longevity of the implanted device. The system also operates with less power consumption, further increasing the longevity of the implanted device. The increased device longevity provides the implanting physician with more options in selecting the most appropriate therapy for the patient. Discomfort and patient mortality due to explantation and replacement of the implanted medical device may be decreased. The system also provides a reference circuit in which the value of the reference voltage and reference current are capable of fine adjustment.

Furthermore, the resulting reference voltage is temperature compensated, i.e. the sensitivity of the reference voltage to temperature variations is reduced. The system also provides a method of generating a temperature compensated reference voltage. A voltage difference is measured between first and second bias circuits, each having different temperature dependencies and each having a voltage drop of less than that of three series-coupled diodes. A temperature dependent reference current is generated from the voltage difference. A current proportional to the reference current is applied to a third bias circuit to establish a first voltage. The first voltage has a temperature dependence that substantially offsets the temperature dependence of a second voltage in series with the first voltage to provide the temperature compensated reference voltage.

The system provides a current reference circuit comprising a first and second bias circuits and a first amplifier. In one embodiment, the first bias circuit has a first temperature dependence, and includes less than three series- coupled diodes. The second bias circuit has a second temperature dependence, different from the first temperature dependence, and also includes less than three series-coupled diodes. The first amplifier has first and second inputs electrically coupled to the respective first and second bias circuits. The first amplifier also has an output providing a temperature dependent reference current in response thereto.

In one embodiment, the system also provides a voltage reference circuit, including a third bias circuit and a second amplifier. The third bias circuit is electrically coupled for receiving a current proportional to the reference current and providing a temperature compensated reference voltage in response thereto. The received current, which is proportional to the reference current, establishes a first voltage having a temperature dependence that substantially offsets the temperature dependence of a second voltage that is in series with the first voltage, thereby providing the temperature compensated reference voltage. A second amplifier is electrically coupled to the first and third bias circuits for stabilizing the reference voltage and providing a load current.

Brief Description of the Drawings In the drawings, like numerals describe substantially similar components throughout the several views. Figure 1 is a schematic illustration of a reference circuit including a current reference circuit and a voltage reference circuit.

Figure 2 is a more detailed schematic illustration of the current reference circuit of Figure 1.

Figure 3 is a more detailed schematic illustration of the voltage reference circuit of Figure 1. Figure 4 is a schematic/block diagram illustrating one embodiment of the system in which the reference circuit is used in a cardiac rhythm management system.

Detailed Description In the following detailed description, reference is made to the accompanying drawings which form a part hereof, and in which is shown by way of illustration specific embodiments in which the invention may be practiced. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention, and it is to be understood that the embodiments may be combined, or that other embodiments may be utilized and that structural, logical and electrical changes may be made without departing from the spirit and scope of the present invention. The following detailed description is, therefore, not to be taken in a limiting sense, and the scope of the present invention is defined by the appended claims and their equivalents. Figure 1 is a schematic illustration of one embodiment of the system providing a silicon integrated circuit (IC) reference circuit 300 including current reference circuit 305 and voltage reference circuit 310, each receiving a power supply voltage V_DD at power supply node 315 and a ground voltage V_ss at ground voltage node 320. Current reference circuit 305 provides an internal reference current, as discussed below, and provides a voltage at node 325 from which a current proportional to the reference current can be generated in voltage reference circuit 310. Current reference circuit 305 is capable of providing a current proportional to the reference current through node 330. Current reference circuit 305 also provides through node 335 a first bias voltage to voltage reference circuit 310 for biasing an amplifier input therein. Current reference circuit 305 also provides through node 336 a fourth bias voltage to voltage reference circuit 310 for biasing current mirrors therein. Voltage reference circuit 310 provides a reference voltage at node 340.

Figure 2 is a more detailed schematic illustration of one embodiment of current reference circuit 305. In Figure 2, current reference circuit 305 comprises first amplifier 350, first bias network 355, second bias network 360, and startup circuit 365. First bias network 355 includes two series-coupled diodes 366 and 367, implemented as base-emitter (BE) junctions of bipolar junction transistors (BJTs), each BJT receiving an independent collector-emitter current, I_CE, as described below. Second bias network 360 includes two series- coupled diodes 368 and 369, implemented as BE junctions of BJTs, each BJT receiving an independent I_CE, as described below. Second bias network 360 also includes first resistor 370 in series with the two series-coupled diodes, 368 and 369, therein.

The BE junction areas of the diodes 368 and 369 in the second bias network 360 are different from the BE junction areas of the diodes 366 and 367 in the first bias network 355. In one embodiment, the BE junction areas of the diodes 368 and 369 in the second bias network 360 are made four times larger than the BE junction areas of the diodes 366 and 367 in the first bias network 355. Each of the diodes 368 and 369 is implemented as four parallel-connected BJT's of the same area as the single BJT's that implement the diodes 366 and 367. Other BE junction area ratios and connection configurations could also be used without departing from the scope and spirit of the present invention.

Substantially identical I_CE currents are provided to each of diodes 366, 367, 368, and 369. Since diodes 368 and 369 in second bias circuit 360 each have four times larger BE junction area than each of diodes 366 and 367 in first bias circuit 355, diodes 368 and 369 in second bias circuit 360 each have approximately one fourth the current density of diodes 366 and 367 in first bias circuit 355, as a result of the substantially identical I_CE currents.

First bias network 355 provides at node 335 to a first input of first amplifier 350 the first bias voltage having a first temperature dependence resulting from the two series-coupled diodes, 366 and 367, therein. Second bias network 360 provides across its two series-coupled diodes, 368 and 369, a second bias voltage, having a second temperature dependence, at node 380, which is coupled through first resistor 370 to the output and the second input at node 385 of first amplifier 350. First amplifier 350 measures the voltage difference, at nodes 335 and 385, between the first and second bias networks 355 and 360, providing in response thereto a reference current, I_PTAT, through first resistor 370, that is proportional to absolute temperature (PTAT). The value of I_PTAT is illustrated by Equation (1).

NV_τ*\n(M)

PTAT p O

In Equation (1), the value of reference current I_PTAT is determined from: the number, N, of diodes connected in series in each of first and second bias circuits, 355 and 360, in this case N=2; the thermal voltage, V_τ, which is approximately 0.0267 Volts at 37 degrees Celsius and is proportional to absolute temperature; the resistance value R₃₇₀ of first resistor 370, and the argument M of the natural logarithm, which is equal to the ratio of current densities in the diodes 366 and 367 in first bias network 355 to the current densities in the diodes 368 and 369 in second bias network 360, in this case M=4. In one embodiment, first resistor 370 is a variable or trimmable resistor having an adjustable resistance value R₃₇₀ near approximately 1.482 megaohms for adjusting reference current _AT to 50 nA at 37 degrees Celsius, or other fine adjustment to a desired value at a particular temperature.

The first bias voltage at node 335, the fourth bias voltage at node 336, and a voltage at node 325 from which reference current I_PTAT is generated by output transconductor 390 of first amplifier 350, are each provided to voltage reference circuit 310. The voltage at node 325, from which reference current I_PTAT is generated, is also provided to startup circuit 365 and to each of current mirror p-channel field-effect transistors PFETs 391-395. Current mirror PFETs 391-393 provide a I_CE currents, proportional to reference current, I_PTAT, to respective diodes 366-368. Output transconductor PFET 390 of first amplifier 350 provides the reference current, I_PTAT, as the I_CE current of diode 369. Current mirror PFET 394 provides a current, proportional to reference current I_PTAT, to diode-connected transistor 395, which generates the fourth bias voltage at node 336 in response thereto. The fourth bias voltage at node 336 is also provided to first amplifier 350 for biasing the gate terminals of current mirror n-channel field-effect transistors (NFETs) 415-417 contained therein. Current mirror PFET 395 is optionally included for providing a bias current at node 330, such as to another circuit on the integrated circuit. Other current mirror PFETs may similarly be used to provide other bias current sources to other circuits. Also, current mirror NFETs, having a gate terminal electrically coupled to node 336 and a source terminal electrically coupled to ground node 320, may be used to provide bias current sinks to other circuits. In one embodiment, first amplifier 350 is a folded cascode amplifier that is carefully designed to operate from a reduced power supply voltage V_DD at power supply node 315, such as a power supply voltage V_DD as low as approximately 1.3 Volts, when biased by first and second bias circuits, 355 and 360. This allows reference circuit 300 to operate at a reduced power supply voltage V_DD, such as, for example, a voltage approximately between 1.3 Volts and 3.25 Volts provided by a single lithium-silver vanadium pentoxide battery cell in an implantable defibrillator. In another example, reference circuit 300 is operated from a power supply voltage V_DD approximately between 1.65 Volts and 3.25 Volts. Thus, operation at a low power supply voltage, V_DD, is particularly advantageous for battery powered electronics devices, such as portable electronics or implantable electronics, but operation at higher values of power supply voltage V_DD is also possible.

Folded cascode first amplifier 350 comprises source-coupled input NFETs 400 and 401, having respective gate terminals electrically coupled to the first and second bias circuits 355 and 360. The drain terminals of input NFETs 400 and 401 are respectively coupled to the drain terminals of load PFETs 405 and 406, the source terminals of which are electrically coupled to power supply node 315. Cascode PFETs 410 and 411 are each electrically coupled at their respective source terminals to the drain terminals of respective load PFETs 405 and 406. The drain terminals of cascode PFETs 410 and 411 are respectively electrically coupled to drain terminals of current mirror NFETs 415 and 416 at respective nodes 325 and 420.

Current mirror NFET 417 is electrically coupled at its drain terminal to the source terminals of each of input NFETs 400 and 401. The source terminals of each of current mirror NFETs 415 -417 are coupled to ground node 320. The gate terminals of each of load PFETs 405 and 406 and cascode PFETs 410 and

411 are coupled to node 420, with cascode PFET 411 being diode-connected, i.e. the gate and drain terminals of cascode PFET 411 are both connected to node

420. Output transconductor 390 is a PFET having a gate terminal electrically coupled to node 325, a source terminal electrically coupled to power supply node

315, and a drain terminal electrically coupled to the second bias circuit 360 at node 385.

In order to obtain the above-described operation at reduced power supply voltages V_DD, first amplifier 350 must be carefully designed and biased. In one embodiment, input NFETs 400 and 401 are sized for operation in the weak inversion region. Current mirror NFETs 415-417 and load PFETs 405 and 406 are sized for operation in the moderate inversion region, i.e. between the weak inversion and strong inversion regions. Cascode PFETs 410 and 411 are sized for operation in the weak inversion region., and more particularly, cascode PFETs 410 and 411 are sized for operation at much lower current densities than load PFETs 405 and 406. Current density, J, through a field-effect transistor

(FET) is illustrated in Equation (2)

L J=/^χ — w (2)

In Equation (2), current density J is determined from: the current I through the FET, L is the length of the FET between its drain and source regions, and W is the width of the FET in a direction transverse to the current flow between the drain and source regions. In one embodiment, the dimensions, L and W, and bias currents, I, in the cascode PFETs 410 and 411 and load PFETs 405 and 406 are designed such that the load PFETs 405 and 406 have approximately between six and seven hundred times the current density J as in the cascode PFETs 410 and 411. At the bias point of first amplifier 350, i.e. nodes 335 and 385 at approximately equal voltages, the above-described sizing of load PFETs 405 and 406 and cascode PFETs 410 and 411 keeps the load PFETs 405 and 406 out of the nonsaturation/linear/triode region of operation. By diode-connecting cascode PFET 411, as described above, a separate bias voltage generation circuit for cascode devices 410 and 411 is not needed. In one embodiment, current mirror PFETs 391-394 and output transconductor PFET 390 are operated in the strong inversion region, and sized similarly to load PFETs 405 and 406.

As described below, the folded cascode topology of first amplifier 350 advantageously operates at a lower power supply voltage V_DD than, for example, a conventional differential amplifier having a current mirror load. The minimum power supply voltage, V_{DD mιn} needed to operate first amplifier 350 is illustrated in Equation (3).

In Equation (3), the minimum supply voltage, V_{DD mιn}, is determined from: the BE voltage of diode 366, V₃₆₆, of approximately 0.5 Volts; the BE voltage of diode 367, V₃₆₇, of approximately 0.5 Volts; the process minimum threshold voltage of input NFET 400 of approximately 0.5 Volts; the minimum drain-source voltage, ΔV₄₀₀, of approximately 0.1 Volts to keep input NFET 400 out of the nonsaturation/linear/triode region of operation; and, the minimum drain-source voltage, ΔV₄₀₅, required to keep load FET 405 out of the nonsaturation/linear/triode region of operation, or approximately 0.2 Volts.

Thus, according to Equation (3), first amplifier 350 is capable of operating from a minimum power supply voltage, V_{DD mιn}, as low as approximately 0.8 Volts, for this biasing arrangement. In this case, the minimum power supply voltage, V_{DD min}, needed for operation of the current reference circuit 305 is not limited by the minimum power supply voltage needed for operation of first amplifier 305. Instead, the minimum power supply voltage needed for operation of current reference circuit 305 is either limited by the voltage requirements of first bias circuit 355 and its associated current mirror PFET 392, or similarly limited by the voltage requirement of second bias circuit 360 and output transconductor PFET 390. For example, the voltage requirement of first bias circuit 355 and associated current mirror PFET 392 is illustrated in Equation (4).

DD. m 366 367 39₂ V*)

In Equation (4), the minimum power supply voltage, V_{DD min} is determined from: the BE voltage of diode 366, V₃₆₆, of approximately 0.5 Volts; the BE voltage of diode 367, V₃₆₇, of approximately 0.5 Volts; and, the minimum drain-source voltage, ΔV₃₉₂, required to keep current mirror PFET 392 out of the nonsaturation/linear/triode region of operation, or approximately 0.25 Volts. Thus, according to Equation (4), current reference circuit 305 is capable of operating from a minimum power supply voltage, V_{DD min}, as low as approximately 1.3 Volts for this biasing arrangement. In one embodiment, current reference circuit 305 is operated from a power supply voltage V_DD that is approximately equal to 1.3 Volts. In another embodiment, current reference circuit 305 is operated at a higher power supply voltage V_DD, such as at approximately between 1.65 Volts and 3.25 Volts. It is also understood that current reference circuit 305 may be operated at even higher power supply voltages V_DD .

The value of the reference current, I_PTAT, through R₃₇₀, as described in Equation (1) represents the desired one of two possible stable operating points of current reference circuit 305. More particularly, current reference circuit 305 also has a stable operating point when I_PTAT = 0. A bootstrapping function performed by startup circuit 365 ensures that current reference circuit 305 operates at the desired stable operating point described in Equation (1), and not at the undesired stable operating point of I_PTAT = 0.

Startup circuit 365 is designed to establish a voltage at node 325 that turns on output transconductor PFET 390 and current mirror PFETs 391-394 such that these devices provide bias current to first and second bias circuits 355 and 360 and first amplifier 350. Startup circuit 365 is also designed to release control over the voltage at node 325 when current reference circuit 350 operates as described in Equation (1).

Startup circuit 365 comprises PFET 500, NFET 505, and pnp BJT 510. PFET 500 has a source terminal that is electrically coupled to power supply node 315, a gate terminal that is electrically coupled to node 325, and a drain terminal that is electrically coupled to the drain terminal of NFET 505 at node 515. NFET 505 has a gate terminal electrically coupled to power supply node 315 and a source terminal that is electrically coupled to ground node 320. BJT 510 has an emitter terminal that is electrically coupled to node 325, a base terminal that is electrically coupled to node 515, and a collector terminal that is electrically coupled to ground node 320.

When power is initially applied to power supply node 315, there is initially no charge on capacitor 520, hence the voltage at node 325 is approximately equal to the voltage at power supply node 315. In this first state, PFET 500 is turned off due to insufficient voltage between its gate and source terminals at nodes 325 and 315, respectively. NFET 505 is turned on, since the voltage between its gate terminal, at power supply node 315, and its source terminal, at ground node 320, is approximately equal to the power supply voltage V_DD. The voltage at node 515 is a diode voltage drop below the voltage at node 325, which is approximately equal to the voltage at power supply node 315. As a result, NFET 505 operates in a saturation region, and provides base current to BJT 510.

The collector-emitter current, I_CE, of BJT 510 charges capacitor 520 to decrease the voltage at node 325 from its initial value at approximately V_DD until PFET 500, output transconductor PFET 390, and current mirror PFETs 391-395 turn on. In this second state, current reference circuit 305 establishes operation at the stable operating point described in Equation (1). PFET 500 is designed to source more current than NFET 505 can sink under all process and supply voltage variations, hence the voltage of node 515 is increased to V_DD, and BJT 510 turns off. NFET 505 remains on, but is sized to draw an acceptably low quiescent current in this second state.

Figure 3 is a more detailed schematic illustration of the voltage reference circuit 310 of Figure 1. In Figure 3, voltage reference circuit 310 comprises a second amplifier 600 and a third bias circuit 605. Second amplifier 600 receives at a first input at node 335 the first bias voltage, which is two series-coupled diode voltage drops above the voltage at ground node 320. Second amplifier 600 is connected in a negative feedback configuration, i.e. the output of second amplifier 600 is coupled to the inverting second input, at node 610, of second amplifier 600, through third bias circuit 605. Because second amplifier 600 is connected in a negative feedback configuration, a "virtual ground" exists between the first input, at node 335, and the second input, at node 610, of second amplifier 600, thereby providing a voltage approximately equal to the first bias voltage at its second input at node 610. Thus, the voltage at node 610 is approximately equal to two series-coupled diode voltage drops above the voltage at ground node 320. Third bias circuit 605 comprises second resistor 620, in series with diode 611, which is implemented as a BJT biased with a collector- emitter current, I_CE, received from current mirror PFET 613 having a gate terminal electrically coupled to node 325.

In third bias circuit 605, the BE voltage of diode 611 offsets the voltage at node 610 from a second voltage at node 615, such that the resulting second voltage at node 615 is approximately one diode voltage drop above the voltage at ground node 320. The temperature dependence of the second voltage at node 615 is substantially offset by the temperature dependence of a first voltage across resistor 620, which is created by the temperature dependent current provided by current mirror NFET 617. When the resistance of second resistor 620 is such that the resulting voltage at node 340 is approximately equal to the bandgap voltage of silicon, i.e. approximately 1.25 Volts, the resulting voltage at node 340 is approximately temperature compensated. In one embodiment, second resistor 620 is a variable or trimmable resistor having an adjustable resistance value

near approximately 7.0 megaohms for adjusting the reference voltage at node 340 to 1.25 Volts with the desired degree of accuracy.

Second amplifier 600 is a folded-cascode amplifier designed and operating similarly to first amplifier 350, which was discussed in detail above. Second amplifier 600 comprises: input NFETs 700 and 701; load PFETs 705 and 706 ; output transconductor NFET 720 ; cascode PFETs 710 and 711; and, current mirror NFETS 730-733. Unlike first amplifier 350, second amplifier 600 is output compensated, i.e. its low frequency pole is determined, in part, by capacitor 800. Gain degeneration resistor 810, electrically coupling the common source terminals of load PFET 706 and output transconductor PFET 720 to the power supply node 315, and capacitor 820 provide improved frequency response characteristics of second amplifier 600. Second amplifier 600 stabilizes the reference voltage at node 340 and provides a load current to any other circuits that are electrically coupled to thereto.

The minimum power supply voltage, V_{DD mιn}, needed for operation of the voltage reference circuit 310 is illustrated by Equation (5).

V DD,mm y 340 +AV 7_0 +V 810

In Equation (5), the minimum power supply voltage, V_{DD min}, needed for operation of the voltage reference circuit 310 is approximately limited by: the value of the reference voltage at node 340, V₃₄₀, of approximately 1.25 Volts; the minimum drain-source voltage needed for operation of output transconductor NFET 720 in the saturation region, ΔV₇₂₀, of approximately 0.175 Volts; and, the voltage V_g]0, across resistor 810, which is negligible. According to Equation (5), the resulting minimum power supply voltage, V_{DDι min}, needed is approximately 1.45 Volts, however, as will be shown, simulation data indicates that voltage reference circuit 310 is capable of operating at an even lower power supply voltage V_{DD> min}. Table 1 illustrates the performance of reference circuit 300, as obtained from a circuit simulation performed using the HSPICE Version H96.1 circuit simulator from Metasoftware, Inc., of Campbell, CA.

Table 1 : Operation of Reference Circuit 300 At Various Power Supply Voltages

The results illustrated in Table 1 indicate that current reference circuit 305 and voltage reference circuit 310 are capable of operating together from a power supply voltage V_DD that is even lower than the 1.45 Volts yielded by Equation (5), such as a minimum power supply voltage V_{DD min} that is approximately equal to 1.3 Volts. In one embodiment, current reference circuit 305 and voltage reference circuit 310 are operated from a power supply voltage V_DD of approximately 1.3 Volts. In another embodiment, a power supply voltage V_DD approximately between 1.65 Volts and 3.25 Volts is used to ensure proper operation of current reference circuit 305 and voltage reference circuit 310. However, it is understood that an even higher power supply voltage V_DD could also be used.

Figure 4 is a schematic/block diagram illustrating one embodiment of the system in which reference circuit 300 is used in a cardiac rhythm management system such as implantable cardioverter-defibrillator (ICD) 900. In one embodiment, ICD 100 includes a single lithium-silver vanadium pentoxide battery 905, providing the power supply voltage V_DD at node 315 to reference circuit 300, and providing the ground voltage V_ss at node 320 to reference circuit 300 and other electronics 910. Reference circuit 300 provides the reference voltage at node 340 and a current proportional to the reference current I_PTAT, through node 330, to electronics 910.

In one embodiment, the power supply voltage V_DD provided by battery 905 is approximately equal to 3.25 Volts during time periods in which ICD 900 is monitoring a patient's heart. However, the power supply voltage V_DD provided by the battery may drop to between approximately 1.65 Volts and 3.25 Volts while charging capacitors for providing an electrical defibrillation countershock to restore the patient's heart to a normal rhythm.

For battery powered electronics applications, reference circuit 300 offers significant advantages; it may eliminate the need for electrically coupling more than one battery in series in order to obtain a higher power supply voltage. Also, in some batteries, an internal battery impedance increases over the course of the battery life, thereby reducing the voltage available at the battery terminals when appreciable currents are drawn therefrom. Thus, the useful life of the battery may be extended by using a reference circuit that is capable of operating from this reduced battery terminal voltage near the end of the battery's life. This is particularly true for implantable medical devices such as pacemakers and defibrillators. For example, a pacemaker may draw appreciable currents from the battery during the charging of a capacitor for the subsequent delivery of an electrical pacing pulse. In another example, a defibrillator may draw appreciable currents from the battery during the charging of a capacitor for the subsequent delivery of an electrical defibrillation countershock. Circuits that are capable of operating at reduced power supply voltages can extend battery life for several years, thereby avoiding surgical explantation and replacement of the implanted device. In elderly patients, such traumatic surgical intervention may be both uncomfortable and risky. The system, by operating from a power supply voltage as low as approximately 1.3 Volts, may significantly increase the longevity of the implanted device in which it is used. The system also operates with low power consumption, further enhancing the longevity of the implanted device. The increased device longevity provides the implanting physician with more options in selecting the most appropriate therapy for the patient. Discomfort and patient mortality due to explantation and replacement of the implanted medical device may be decreased. The system also provides a reference circuit in which the value of the reference voltage and reference current are capable of fine adjustment. Furthermore, the resulting reference voltage is temperature compensated, i.e. the sensitivity of the reference voltage to temperature variations is reduced.

It is to be understood that the above description is intended to be illustrative, and not restrictive. Many other embodiments will be apparent to those of skill in the art upon reviewing the above description. The scope of the invention should, therefore, be determined with reference to the appended claims, along with the full scope of equivalents to which such claims are entitled.

Claims

WHAT IS CLAIMED IS:

1. A reference circuit comprising: a first bias circuit, having a first temperature dependence and including less than three series-coupled diodes; a second bias circuit having a second temperature dependence, different from the first temperature dependence, and including less than three series- coupled diodes; and a first amplifier having first and second inputs electrically coupled to the respective first and second bias circuits, and having an output providing a temperature dependent reference current in response thereto.

2. The reference circuit of claim 1 , wherein the first amplifier is a folded- cascode amplifier.

3. The reference circuit of claim 1 , wherein a power supply voltage provided to the first amplifier is greater than approximately 1.3 Volts.

4. The reference circuit of claim 1, wherein a power supply voltage provided to the first amplifier is approximately between 1.65 Volts and 3.25 Volts.

5. The reference circuit of claim 1, wherein the first amplifier is capable of operating from a power supply voltage greater than approximately 1.3 Volts.

6. The reference circuit of claim 1, wherein the first amplifier is capable of operating from a power supply voltage approximately between 1.65 Volts and 3.25 Volts.

7. The reference circuit of claim 1, wherein the first and second bias circuits bias the first amplifier for operation from a power supply voltage greater than approximately 1.3 Volts.

8. The reference circuit of claim 1, wherein the first and second bias circuits bias the first amplifier for operation from a power supply voltage approximately between 1.65 Volts and 3.25 Volts.

9. The reference circuit of claim 1 , wherein the first bias circuit comprises first and second series-coupled diodes, and the second bias circuit comprises third and fourth diodes, and at least one of the first and second diodes has a junction area that is different from at least one of the third and fourth diodes.

10. The reference circuit of claim 9, wherein each of the first and second diodes has a junction area that is different from each of the third and fourth diodes.

11. The reference circuit of claim 1 , further comprising a third bias circuit, electrically coupled for receiving a current proportional to the reference current and providing a reference voltage in response thereto.

12. The reference circuit of claim 11 , wherein the reference voltage is substantially temperature compensated.

13. The reference circuit of claim 12, wherein the current proportional to the reference current establishes a first voltage having a temperature dependence that substantially offsets the temperature dependence of a second voltage in series with the first voltage to provide the reference voltage.

14. The reference circuit of claim 11 , further comprising a second amplifier electrically coupled to the first and third bias circuits for stabilizing the reference voltage and providing a load current.

15. The reference circuit of claim 14, wherein a power supply voltage provided to the second amplifier is greater than approximately 1.3 Volts.

16. The reference circuit of claim 14, wherein a power supply voltage provided to the second amplifier is approximately between 1.65 Volts and 3.25 Volts.

17. The reference circuit of claim 14, wherein the second amplifier is capable of operating from a power supply voltage that is greater than approximately 1.3 Volts.

18. The reference circuit of claim 14, wherein the second amplifier is capable of operating from a power supply voltage that is approximately between 1.65 Volts and 3.25 Volts.

19. A cardiac rhythm management system, comprising: a battery; an electronics circuit, for controlling delivery of cardiac therapy to a patient; and a reference circuit, electrically coupled to the battery for receiving a power supply voltage therefrom, and providing a reference voltage to the electronics circuit, wherein the reference circuit is capable of operating from the power supply voltage as low as approximately 1.3 Volts.

20. A cardiac rhythm management system, comprising: a battery; an electronics circuit, for controlling delivery of cardiac therapy to a patient; and a reference circuit, electrically coupled to the battery for receiving a power supply voltage therefrom, and providing a reference voltage to the electronics circuit, wherein the reference circuit includes: a first bias circuit, having a first temperature dependence and including less than three series-coupled diodes; a second bias circuit having a second temperature dependence, different from the first temperature dependence, and including less than three series-coupled diodes; and a first amplifier having first and second inputs electrically coupled to the respective first and second bias circuits, and having an output providing a temperature dependent reference current in response thereto.

21. A reference circuit, comprising: a first bias circuit, having a first temperature dependence and including less than three series-coupled diodes; a second bias circuit, having a second temperature dependence, different from the first temperature dependence, and including less than three series- coupled diodes; a folded cascode first amplifier, receiving a power supply voltage approximately between 1.3 Volts and 3.25 Volts, and having first and second inputs electrically coupled to the respective first and second bias circuits, and having an output providing a temperature dependent reference current in response thereto; and a voltage reference circuit, receiving a current proportional to the reference current, and providing a first voltage having a temperature dependence that substantially offsets the temperature dependence of a second voltage in series with the first voltage to provide a resulting temperature compensated reference voltage.

22. A method comprising the steps of: measuring a voltage difference between a first bias circuit and a second bias circuit, the first and second bias circuits having different temperature dependencies and each having a voltage drop less than that of three series- coupled diodes; and generating a temperature dependent reference current from the voltage difference.

23. The method of claim 22, wherein the first bias circuit comprises less than three series-coupled diodes.

24. The method of claim 22, wherein the second bias circuit comprises a first resistor in series with less than three series-coupled diodes.

25. The method of claim 24, further comprising the step of setting the resistance of the first resistor to establish the value of the reference current at a particular temperature.

26. The method of claim 22, wherein the step of measuring the voltage difference between the first bias circuit and the second bias circuit includes using a first amplifier.

27. The method of claim 26, wherein using the first amplifier comprises using a folded-cascode amplifier.

28. The method of claim 26, wherein a power supply voltage provided to the first amplifier is greater than approximately 1.3 Volts.

29. The method of claim 26, wherein a power supply voltage provided to the first amplifier is approximately between 1.65 Volts and 3.25 Volts.

30. The method of claim 22, further comprising the step of providing the reference current to a third bias circuit to generate a temperature compensated reference voltage.

31. The method of claim 22, further comprising the step of applying a current proportional to the reference current to a third bias circuit to establish a first voltage having a temperature dependence that substantially offsets the temperature dependence of a second voltage in series with the first voltage to provide the reference voltage.

32. The method of claim 31 , wherein the third bias circuit comprises a second resistor, across which the first voltage is established, in series with the second voltage.

33. The method of claim 32, further comprising the step of establishing the resistance of the second resistor to establish the value of the first voltage such that is substantially offsets the temperature dependence of the second voltage.