WO2006012245A1

WO2006012245A1 - Receiver for use in wireless communications and method and terminal using it

Info

Publication number: WO2006012245A1
Application number: PCT/US2005/022335
Authority: WO
Inventors: Moshe Ben-Ayun; Nir Corse; Ovadia Grossman; Mark Rozental
Original assignee: Motorola, Inc.
Priority date: 2004-06-29
Filing date: 2005-06-23
Publication date: 2006-02-02
Also published as: CN100588123C; CA2572236A1; GB2415846A; JP2008509577A; GB0414459D0; CN1981437A; GB2415846B; DE112005001456T5; AU2005267308A1; CA2572236C; AU2005267308B2

Abstract

A wireless receiver comprising an input signal path (101), a circuit (113) for producing in-phase and quadrature components, and an estimator (204) for periodically estimating an imbalance in amplitude between the in-phase and quadrature components and for applying a relative adjustment in amplitude to compensate for the detected imbalance, wherein the estimator (204) is operable (i) to divide samples Ii of the in-phase component I(t) and corresponding samples Qi of the quadrature component Q(t) into blocks; (ii) to calculate for each block, a block power value In corresponding to a summation of values of squares of the samples Ii; (iii) to calculate from the block power values In and Qn a block amplitude imbalance value, Formule (I); and (iv) to calculate for a set of the block amplitude imbalance values an average value.

Description

RECEIVER FOR USE IN WIRELESS COMMUNICATIONS AND METHOD AND TERMINAL USING IT

Field of the Invention

This invention relates to a receiver for use in wireless communications and a method and terminal using it. In particular, the invention relates to a direct conversion receiver capable of demodulating a frequency modulated (FM) RF (radio frequency) signal by resolution and use of in-phase (I) and quadrature (Q) components of the modulated signal.

Background of the Invention

Conventional FM wireless receivers built using direct conversion architectures to detect I and Q components of a received signal have an underlying problem. As illustrated later, such receivers can develop an error in relative phase and amplitude between the I and Q components. This error, sometimes referred to as ^quadrature imbalance' , can cause a distortion in the resulting output audio signal. The distortion may be unacceptable to users particularly under conditions when the received signal is subject to Rayleigh fading (herein ^fading') and/or has a low signal to noise ratio. The prior art does not provide a satisfactory solution to the problem of quadrature imbalance.

The present invention is concerned in particular with the amplitude imbalance component of quadrature imbalance. US5705949 proposes a procedure for removing amplitude or gain error between I and Q components. The procedure requires complex processing capacity and is unlikely to be satisfactory in a fading environment.

Summary of Invention

In accordance with a first aspect of the present invention there is provided a wireless receiver in accordance with claim 1 of the accompanying claims.

In accordance with a second aspect of the present invention there is provided a wireless communication method in accordance with claim 12 of the accompanying claims.

In accordance with a third aspect of the present invention, there is provided a wireless communication terminal in accordance with claim 13 of the accompanying claims.

Embodiments of the present invention will now be described by way of example with reference to the accompanying drawings, in which:

Brief Description of the Drawings

FIG. 1 is a schematic block circuit diagram of a known direct conversion RF receiver.

FIG. 2 is a schematic block circuit diagram of a direct conversion RF receiver embodying the invention Description of embodiments of the invention

FIG. 1 shows a known RF direct conversion FM receiver 100 illustrating the problem to be addressed by the present invention. An incoming FM signal x(t) is delivered via an input path 101 having branched connections 103, 105 respectively to two mixers 107, 109. A local oscillator 111 generates a reference signal having the same frequency as the carrier frequency of the incoming signal x(t). A first component of the reference signal is applied directly to the mixer 107 where it is multiplied with the input signal x(t) . A second component of the reference signal is applied to a phase shifter 113 and a phase shifted output of the phase shifter 113 is applied to the mixer 109 where it is multiplied with the input signal x(t). Although the phase shifter 113 in combination with the mixers 107 and 109 is intended to introduce a phase shift of 90 degrees with unity gain between the components of the reference signal applied to the mixers 107 and 109, in practice a phase shift slightly different from 90 degrees and a gain slightly different from unity are introduced. An output signal from the mixer 107 is passed through a low pass filter (LPF) 115 to produce an output in-phase component signal I(t) and an output signal from the mixer 109 is passed through a low pass filter (LPF) 117 to produce an output quadrature component signal Q(t). The imbalance in amplitude introduced into the output of the mixer 109 is shown in block 119 as an imbalance gain A. A mathematical analysis of the arrangement shown in FIG. 1 is as follows:

The input signal may be represented as :

where ω is RF carrier frequency of the input RF signal x(t), /is oscillator arbitrary phase and φ(t) is the frequency modulation of x(t) to be detected.

In addition,x(t) =I(t)+j*Q(t), whereI(t) and Q(t) are in-phase and quadrature components ofx(t).

I(t) = 2 cos(wt + φ(t) + γ) cos(wt) = = cos{2wt + φ(t) + γ) + cos(φ(t) + γ) = cos(φ(t) + γ) after LPF

Q(t) = 2 A cos(wt + φ(t) + γ) sin(M + a) =

= A sin(2wt + φ(t) + γ + a) + A sva(φ(t) + γ + a) = A sin(φ(t) + γ + a) . after LPF

where A represents the amplitude imbalance and arepresents the phase imbalance angle between the phase angles of I(t) and Q(t).

In accordance with an embodiment of the present invention to be described the components I(t) and Q(t) are processed in a manner to be described to estimate and apply an adjustment to eliminate the amplitude imbalance A. The phase imbalance is also estimated and eliminated, e.g. as described in Applicant's copending UK patent application number 0411888.1 The resulting adjusted components are combined to construct the modulation signal φ(t) to provide an audio signal output.

FIG. 2 is a block schematic diagram of a circuit 200 embodying the invention for use in a direct conversion FM receiver. Components having the same reference numerals as components in FIG. 1 have the same function as such components and will not be further described.

The output signal Iff) passed by the low pass filter (LPF) 115 is sampled by a connection 201 and the output signal Q(t) passed by the low pass filter (LPF) 117 is sampled by a connection 203. The respective sampled signals obtained by the connections 201 and 203 are provided as respective inputs to a processor 204 which operates an amplitude imbalance algorithm to be described in detail later. An output signal from the processor 204 is an amplitude imbalance correction signal indicating a value of I/A. This correction signal is applied via a connection 202 to an amplitude modifier 205 which modifies the amplitude of Q(t) by a factor of I/A to eliminate the detected amplitude imbalance A.

A phase adjustment processing circuit (not shown) using samples of I(t) and Q(t) estimates a phase imbalance between I(t) and Q(t), e.g. in the manner described in Applicant's copending UK patent application number 0411888.1, and generates a phase shift control signal corresponding to an equal and opposite value of this estimated phase imbalance. The phase adjustment signal estimated in this way is applied by a phase shifter 207. A signal corresponding to the quadrature component Q(t) is applied from the low pass filter 117 via a connection 226 to the phase shifter 207. The phase shifter 207 thereby applies a phase angle adjustment which compensates for the detected phase imbalance angle CC., An output from the phase shifter 207 corresponding to a phase adjusted value of Q(t) is applied to a processor 209. A signal corresponding to the in-phase component I(t) is also applied as an input to the processor 209 via a connection 224. The processor 209 calculates a value of the quotient Q(t)/I(t) from its respective inputs and supplies a signal representing the result to a processor 211. The processor 211 calculates the value of the arctangent (arctg) of the quotient parameter represented by the input signal from the processor 209. An output signal from the processor 211 is applied to a further processor 213 which calculates the differential with respect to time t of the input signal to the processor 213. Finally, an output signal from the processor 213 is applied to an audio output 215. The audio output 215 includes a transducer (not shown specifically) such as an audio speaker which converts an electronic signal output from the processor 213 into an audio signal, e.g. speech information.

The amplitude imbalance algorithm operated by the processor 204 is as follows. Samples of the components I(t) and Q(t) are taken at a frequency of 20 ksamples/sec. So the length of time of each sample is 1/2Ok = 50μsec. The samples are taken over a sampling period of 500 msec. So the total number of samples is 500msec/50μsec = 10000 samples. The samples are divided into blocks. The block size is selected according to the operating conditions, e.g. the received signal strength or S/N (signal to noise ratio) . For example, for received

signal strength or received S/N equal to or greater than a threshold value, the block size may be set to a first value and for received signal strength or S/N less than the threshold value, the block size may be set for a second, higher value. For example, for S/N equal to or greater than the threshold value of 15dB, there may be 15 samples per block. So there are 10000/15 = 666 blocks in the sampling period. For a S/N lower than the threshold value of 15dB, there may be 100 samples /block. So there are

10000/100 = 100 blocks. The algorithm performs . better in a fading environment with a small block size .

For each block of samples, a value of the power block _s he of I [I_n= ∑ -C ) ^anc* ^{a va}l^ue °f ^tne power of Q block _size

[Q_n = ∑ Qi ) ^is calculated. /=i

For each block, an amplitude imbalance value is calculated from the power of I and the power of Q

using the following calculation: A_n- I— . Thus, the block value of A_n is the square root value of the power of Q divided by the power of

I.

The values of amplitude imbalance found for each of the blocks in a given set of blocks, say 1000 blocks, are sorted in order from lowest to highest.

A sub-set of 45% of the highest block amplitude imbalance values in the sorted set and a sub-set of 45% of the lowest block amplitude imbalance values in the sorted set are rejected leaving only a sub-set of the 10% block amplitude imbalance values between the rejected sub-sets. So for example where there are 1000 blocks in the set, the 450 highest and the 450 lowest block amplitude imbalance values results are rejected leaving 100 block amplitude imbalance values which are further processed.

From the remaining K imbalance results, where K is the number of blocks in the remaining subset, e.g. 10 per cent of the set of blocks evaluated in the above example, a geometric average value is obtained using the following calculation:

where A_00n is the amplitude imbalance to be corrected for . Thus , A_corr is equal to the kth root of the product of the k sample results for A multiplied by each other . A signal corresponding to 1/A_∞n is issued by the processor 204 to be applied by the amplitude modifier 205. The algorithm is performed continuously and adaptively on the received FM modulated signal during periods when a signal is received.

The processor 204 may be operated when any received speech signal plus associated sub-audio signalling is received by the receiver 200. However, if desired, the algorithm may be operated selectively only when a specific input signal is received by the receiver 200. For example, the receiver may operate on a known analogue FM signal received from a RF transmitter. This may for example be a standard FM modulated signal in accordance with the industry standard TIA 603.

Division into blocks of the samples processed by the processor 204 in the manner described earlier is beneficial for processing a signal received in a fading environment. If division into blocks is not applied there is no possibility to reject results that are not correct due to fading. In a fading environment there are fast variations in signal envelope. When a signal is in a deep fade the result for the quotient Q/I (for the block processed when this applies) can be very large (I is close to zero) or very small (Q is close to zero) . Division into blocks, sorting and rejecting high and low results allows incorrect results caused by fading not to be included in the amplitude imbalance estimation. In practice, a wireless terminal is always likely to work in fading environment. Block size is also significant: for fast fading a small block size is optimal, for low S/N a larger block size would be optimal. The receiver 200 may constantly measure the received signal power using a known RSSI (Received

Signal Strength Indicator) . The result may be provided to the estimator 204 which may be operable to adjust the block size automatically using the result provided. It may be assumed that there is a relationship between received signal power and received S/N so that for high received power the S/N is also high. A threshold received power value is used to determine whether a small block size is to be used when the received power value is equal to or above the threshold or.a greater block size is to be used when the received power value is below the threshold.

Using only a selected subset of block amplitude imbalance results, e.g. only 10% of the results as in the example described above, also has the benefit of significantly reducing the complexity of the algorithm operated by the processor 204 and therefore the amount of signal processing required. This results in reduced consumption of power from the battery of the terminal in which the receiver 200 is incorporated.

Finding the geometric average has been found to be better than finding the arithmetic average, because the latter was found to introduce a bias to the results and gave an incorrect amplitude imbalance estimation. An example to illustrate this is as follows: Say for block 1 Ql=3 and 11=2, and for block 2 Ql=2 and 11=3 and A is 1. Al=3/2 and A2=2/3. Using an arithmetic average for A will give 0.5(3/2+2/3) = 1.0833 - an incorrect result. However, using a geometric average for A: will give sqrt ( (3/2) * (2/3) ) =1 - a correct result (where ^λsqrt' is the square root) .

Various processors are shown in FIG. 2. These processors may be separate processors as shown or the functions of -two or more of the processors may be combined into a single processor, e.g. digital signal processor programmed with computational software, as will be apparent to those skilled in the art.

Results

The algorithm operated by processor 204 was tested on simulated and actual analogue FM recorded signals. The actual signals were recorded using a Direct Conversion receiver operating in the manner described with reference to FIG. 2. We measured amplitude imbalance (Amp IM) in %. Amp IM [%] = lOOe where e is given by

Q{t)=Asin(φ(t)+γ+a)=(1+e)sin(φ(t)+γ+a)

Target performance for the error in applying the algorithm is a maximum error in Amp IM = 0.5%.

For a variety of recorded real βOdbm signals in a fading environment at 450MHz, the error in Amp IM was measured and the results ranged from 0.08% to a maximum of 0.45% with an average error of 0.2%. Similarly, the error in Amp IM was estimated for various simulated signals with a signal to noise ratio (SNR) ranging from 15dB to 35dB and the error ranged from 0.08% (35dB SNR) to 0.2% (15dB SNR) .

In contrast, we also estimated the amplitude imbalance

error using the known calculation A = . In a fading

signal environment we obtained an average error value in Amp IM of 4% using the known procedure.

Where the invention is used in a radio receiver, a memory of the radio may be programmed following manufacture to store a table of initial imbalance values versus RF frequency. During operation of the radio the imbalance values (amplitude and phase) will change with time. Thus, updated imbalance information may be gathered in use as described in the above embodiments and used to provide suitable compensation to maintain a suitable quality of audio output signal . The updated imbalance information may also be stored in the memory of the radio to replace the originally stored information.

Summary

In summary, an improved method for adaptive amplitude imbalance compensation in a direct conversion receiver has been provided together with a receiver operating using the method. The method is gives a substantial improvement to estimating the required imbalance compensation in conditions where the received signal is subject to noise and/or fading.

A look up table of initial amplitude imbalance values vs. RF frequencies may be programmed in a memory associated with the receiver, e.g. in a memory of a mobile station in which the receiver is used. This may be for example the so called codeplug which stores the operating programs and data of the mobile station.

During use of the receiver, the amplitude imbalance as a function of frequency will change gradually with time. Information gathered by the processor 204 may be used to update the stored information in the memory.

The invention gives improved audio performance in a wireless terminal having a receiver operating on an FM analogue signal in a direct conversion mode.

Claims

1. A wireless receiver for receiving and demodulating a frequency modulated RF signal by a direct conversion procedure, including an input signal path for delivering an RF input received signal, a circuit for producing in- phase and quadrature components of the received signal, and an estimator for periodically estimating an imbalance in amplitude between the in-phase and quadrature components and for applying an relative adjustment in amplitude to compensate for the detected imbalance, wherein the estimator is operable to

(i) divide samples Ij of the in-phase component I(t) and corresponding samples Q₂- of the quadrature component Q(t) into blocks;

(ii) calculate for each block a block power value I_n corresponding to a summation of values of squares of the samples Ij and a block power value Q_n corresponding to a summation of values of squares of the samples Qi; . (iii) calculate from the block power values I_n and

Q_n a block amplitude imbalance value A_n =—- ; and n

(iv) calculate for a set of the block amplitude imbalance values an average value.

2. A receiver according to any claim 1 wherein the set of block amplitude imbalance values for which an average value is calculated by the estimator is a selected subset of a larger set of block amplitude imbalance values.

3. A receiver according to claim 2 wherein the estimator is operable to reject at least one other subset of block amplitude imbalance values.

4. A receiver according to claim 3 wherein the estimator is operable to sort the block amplitude imbalance values in terms of their sizes and to reject (i) a first subset of block amplitude imbalance values greater in size than a first threshold and (ii) a second subset of block amplitude imbalance values less in size than a second threshold, wherein each of the first and second thresholds corresponds to a subset having a predetermined number of block amplitude imbalance values.

5. A receiver according to claim 4 wherein the estimator is further operable to select from the sorted set a third sub-set of block amplitude imbalance values having block amplitude imbalance values which are less than those of the first sub-set and greater than those of the second sub-set.

6. A receiver according to claim 1 wherein the estimator is operable to calculate a geometric average of the block amplitude imbalance values.

7. A receiver according to claim 1 wherein the receiver is operable to select a received signal of predetermined form to estimate the current amplitude imbalance between the in phase and quadrature components.

8. A receiver according to claim 1 wherein the estimator is operable to select a size for the blocks of samples according to at least one of a detected property of the received signal.

9. A receiver according to claim 8 wherein estimator is operable to select a size for the blocks of samples according to whether the signal to noise ratio or the received signal strength of the received signal is detected to be at least one of a) above or below a given threshold and b) equal to or above a given threshold and to select a second greater size for the blocks of samples according to whether the signal to noise ratio or signal strength is below a given threshold.

10. A receiver according to claim 1 which includes means for periodically detecting an imbalance in phase between the in-phase and quadrature components and for applying an adjustment in relative phase to compensate for the detected imbalance.